LINEAR TECHNOLOGY LTC1966, LTC1967, LTC1968 Technical data

February 2007

Instrumentation Circuitry Using RMS-to-DC Converters

RMS Converters Rectify Average Results

Jim Williams

INTRODUCTION

It is widely acknowledged that RMS (Root of the Mean of the Square) measurement of waveforms furnishes the most accurate amplitude information.1 Rectify-and- average schemes, usually calibrated to a sine wave, are only accurate for one waveshape. Departures from this waveshaperesultinpronouncederrors.Althoughaccurate, RMS conversion often entails limited bandwidth, restricted range, complexity and difficult to characterize dynamic and static errors. Recent developments address these issues while simultaneously improving accuracy. Figure 1 showstheLTC®1966/LTC1967/LTC1968devicefamily.Low frequency accuracy, including linearity and gain error, is inside 0.5% with 1% error at bandwidths extending to 500kHz. These converters employ a sigma-delta based computational scheme to achieve their performance.2

Figure 2’s pinout descriptions and basic circuits reveal an easily applied device. An output filter capacitor is all that is required to form a functional RMS-to-DC converter. Split and single supply powered variants are shown. Such ease of implementation invites a broad range of application; examples begin with Figure 3.

Isolated Power Line Monitor

BEFORE PROCEEDING ANY FURTHER, THE READER IS WARNED THAT CAUTION MUST BE USED IN THE CONSTRUCTION, TESTING AND USE OF THIS CIRCUIT. HIGH VOLTAGE, LETHAL POTENTIALS ARE PRESENT IN THIS CIRCUIT. EXTREME CAUTION MUST BE USED IN WORKINGWITH,ANDMAKINGCONNECTIONSTO,THIS CIRCUIT. REPEAT: THIS CIRCUIT CONTAINS DANGEROUS, HIGH VOLTAGE POTENTIALS. USE CAUTION.

Figure 3’s AC power line monitor has 0.5% accuracy over a sensed 90VAC to 130VAC input and provides a safe, fully isolated output. RMS conversion provides accurate reporting of AC line voltage regardless of waveform distortion, which is common.

, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners.

1See Appendix A, “RMS-to-DC Conversion” for complete discussion of RMS measurement.

2Appendix A details sigma-delta based RMS-to-DC converter operation.

	LINEARITY	CONVERSION	1% ERROR	3dB ERROR
PART NUMBER	ERROR	GAIN ERROR	BANDWIDTH	BANDWIDTH	SUPPLY VOLTAGE		ISUPPLY
	TYP/MAX (%)	TYP/MAX (%)	(kHz)	(kHz)	MIN(V)	MAX(V)	MAX (µA)
LTC1966	0.02/0.15	0.1/0.3	6	800	2.7	±5	170
LTC1967	0.02/0.15	0.1/0.3	200	4MHz	4.5	5.5	390
LTC1968	0.02/0.15	0.1/0.3	500	15MHz	4.5	5.5	2.3mA

Figure 1. Primary Differences in RMS to DC Converter Family are Bandwidth and Supply Requirements. All Devices Have Rail-to-Rail Differential Inputs and Output

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AN106-1

Application Note 106

POSITIVE SUPPLY

2.7V TO 5.5V

DEPENDING ON DEVICE CHOICE

DIFFERENTIAL

+V

OUTPUT TO FILTER CAPACITOR

INPUTS.

INPUT 1

LTC1966

OUTPUT

OUTPUT

MAX COMMON MODE

RANGE = ±V SUPPLY.

LTC1967

OUTPUT REFERRED

INPUT 2

LTC1968

OUTPUT RETURN

MAX DIFFERENTIAL = 1V.

TO THIS PIN.

MINIMUM INPUT = 5mV

ENABLE

–V*

GND

NORMALLY GROUNDED

SET HIGH TO

LTC1966 ONLY.

POWER

*NO –V PIN ON THE LTC1967/LTC1968

SHUT DOWN

0V TO –5V

GROUND

SUPPLY

±5V Supplies, Differential, DC-Coupled

5V Single Supply, Differential, AC-Coupled

RMS-to-DC Converter

RMS-to-DC Converter

5V

5V

VDD

VDD

DC + AC

LTC1966

DC OUTPUT

LTC1966

DC OUTPUT

IN1

VOUT

AC INPUTS

IN1

VOUT

INPUTS

CAVE ZO = 85kΩ

(1VPEAK

CAVE

ZO = 85kΩ

(1VPEAK

IN2

OUT RTN

IN2

OUT RTN

1µF

DIFFERENTIAL)

1µF

DIFFERENTIAL)

VSS GND

VSS GND

EN

EN

CC

0.1µF

AN106 F02

–5V

Figure 2. RMS Converter Pin Functions (Top) and Basic Circuits (Bottom).

Pin Descriptions are Common to All Devices, with Minor Differences

1	T1	5
LINE INPUT		A
90VAC
TO 140VAC
4		6
		7

B 8

BIAS SUPPLY

1k

–5V

	100µF
	+	1k
			5V	DANGER! Lethal Potentials Present—See Text
+
100Ω	100µF
1W
ISOLATED LINE SENSE				5V		–5V		RMS CONVERTER
		100Ω
		1W		+V		–V
1k	100Ω	0.25%			C1				+
				IN1			OUT
				IN2	LTC1966			1µF	A1
120VAC		10Ω				OUT RTN
									LT®1006
TRIM		0.25%		EN		GND			–
									100k*

T1 = TAMURA-PAN MAG 3FS-212					100k*
	= 1N4148

AN106 F03

= 1N4689 5.1V

*1% METAL FILM RESISTOR

1µF = WIMA MKS-2

RMS OUT 0.9V TO 1.4V =

90VAC TO 140VAC

Figure 3. Isolated Power Line Monitor Senses Via Transformer with 0.5% Accuracy Over 90VAC to 130VAC Input. Secondary Loading Optimizes Transformer Voltage Conversion Linearity

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AN106-2

Application Note 106

TheAClinevoltageisdivideddownbyT1’sratio.Anisolated and reduced potential appears across T1’s secondary B, where it is resistively scaled and presented to C1’s input. Power for C1 comes from T1’s secondary A, which is rectified, filtered and zener regulated to DC. A1 takes gain and provides a numerically convenient output. Accuracy is increased by biasing T1 to an optimal loading point, facilitated by the relatively low resistance divider values. Similarly, although C1 and A1 are capable of single supply operation, split supplies maintain symmetrical T1 loading. The circuit is calibrated by adjusting the 1k trim for 1.20V output with the AC line set at 120VAC. This adjustment is madeusing avariableAC line transformer and a wellfloated (use a line isolation transformer) RMS voltmeter.3

Figure 4’s error plot shows 0.5% accuracy from 90VAC to 130VAC, degrading to 1.4% at 140VAC. The beneficial effect of trimming at 120VAC is clearly evident; trimming at full scale would result in larger overall error, primarily due to non-ideal transformer behavior. Note that the data is specific to the transformer specified. Substitution for T1 necessitates circuit value changes and recharacterization.

	2.0
(%)	1.5
ERROR	1.0
READING	0.5
	0		120VAC TRIM POINT
OUTPUT
	–1.0
RMS	–0.5
	–1.5
	–2.0
											130		140
	90				100		110		120

RMS INPUT VOLTAGE (V)

AN106 F04

Figure 4. Error Plot for Isolated Line Monitor Shows 0.5% Accuracy from 90VAC to 130VAC, Degrading to 1.4% at 140VAC. Transformer Parasitics Account for Almost All Error

Fully Isolated 2500V Breakdown,

Wideband RMS-to-DC Converter

NOTE: BEFORE PROCEEDING ANY FURTHER, THE READER IS WARNED THAT CAUTION MUST BE USED IN THE CONSTRUCTION, TESTING AND USE OF THIS CIRCUIT. HIGH VOLTAGE, LETHAL POTENTIALS ARE PRESENT IN THIS CIRCUIT. EXTREME CAUTION MUST BE USED IN WORKING WITH, AND MAKING CONNECTIONS TO, THIS CIRCUIT. REPEAT: THIS CIRCUIT CONTAINS DANGEROUS, HIGH VOLTAGE POTENTIALS. USE CAUTION.

Accurate RMS amplitude measurement of SCR chopped AC line related waveforms is a common requirement. This measurement is complicated by the SCR’s fast switching of a sine wave, introducing odd waveshapes with high frequency harmonic content. Figure 5’s conceptual SCRbased AC/DC converter is typical. The SCRs alternately chop the 220VAC line, responding to a loop enforced, phase modulated trigger to maintain a DC output. Figure 6’s waveforms are representative of operation. Trace A is one AC line phase, trace B the SCR cathodes. The SCR’s irregularly shaped waveform contains DC and high frequency harmonic, requiring wideband RMS conversion for measurement. Additionally, for safety and system interface considerations, the measurement must be fully isolated.

3See Appendix B, “AC Measurement and Signal Handling Practice,” for recommendations on RMS voltmeters and other AC measurement related gossip.

DANGER! Lethal Potentials Present—See Text

220VAC
INPUT	DC
	OUTPUT
NEUTRAL
220VAC
AC LINE SYNC
AND PHASE	TRIGGER
MODULATION
	AN106 F05
	REF

Figure 5. Conceptual AC/DC Converter is Typical of SCR-Based Configurations. Feedback Directed, AC Line Synchronized Trigger Phase Modulates SCR Turn-On, Controlling DC Output

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AN106-3

Application Note 106

A = 100V/DIV

B = 50V/DIV

ON 170 VDC

LEVEL

1ms/DIV	AN106 F06

Figure 6. Typical SCR-Based Converter Waveforms Taken at AC Line (Trace A) and SCR Cathodes (Trace B). SCR’s Irregularly Shaped Waveform Contains DC and High Frequency Harmonic, Requiring Wideband RMS Converter for Measurement

Figure 7 provides isolated power and data output paths to anRMS-to-DCconverter,permittingsafe,wideband,digital output RMS measurement. A pulse generator configured comparator combines with Q1 and Q2 to drive T1, resulting in isolated 5V power at T1’s rectified, filtered and zener regulated output. The RMS-to-DC converter senses either 135VAC or 270VAC full-scale inputs via a resistive divider. The converter’s DC output feeds a self-clocked, serially interfaced A/D converter; optocouplers convey output data across the isolation barrier. The LTC6650 provides a 1V reference to the A/D and biases the RMS-to-DC converter’s inputs to accommodate the voltage divider’s AC swing. Calibration is accomplished by adjusting the 20k trim while noting output data agreement with the input AC voltage. Circuit accuracy is within 1% in a 200kHz bandwidth.

	PULSE GENERATOR		POWER	5V
	1N4148	1k	DRIVER
	220k			1k
					Q2		ISOLATED POWER SUPPLY
					ZTX-749
	–			100Ω	2Ω		1N5817
0.001µF			1k	Q1
	LT1671									5V ISO
					3				+
				2N2369		1		1N4689
	+						1N4148			100µF
			750k			6		5.1V
					4
			750k
			5V		T1
			750k
				ISOLATION/POWER
				TRANSFORMER

2500V BREAKDOWN	DANGER! Lethal Potentials Present—See Text
ISOLATION BARRIER

DATA	5V	4.7k
ISOLATORS		5V ISO	A/D			RMS CONVERTER
SCK		5V ISO		5V ISO			CALIBRATE
						5V ISO
			+V		–		20k
							0.1µF
		SCK	VIN	LT1006		+V

DATA

5V

LTC2400

+

OUT

IN1

OUTPUTS

SDO

1µF†

FO

GND CS REF

LTC1967

OUT RTN

IN2

SDO

EN

GND

		+
	5V ISO	10µF
	4.7k	5V ISO
	1µF			LT6650	1VDC
	ISOLATORS = AGILENT-HCPL-2300-010	400mV	+
	T1 = BI TECHNOLOGIES HM-41-11510			OUT	AN106 F07
		REFERENCE
	* = 1% METAL FILM RESISTOR		–
	† = WIMA MKS-2			FB	15k*
	= CIRCUIT COMMON				1µF
	= AC LINE GROUND				10k*

182k*

182k*

INPUT 135VAC OR 270VAC FULL SCALE

1k*

Figure 7. Isolated RMS Converter Permits Safe, Digital Output, Wideband RMS Measurement. T1-Based Circuitry Supplies Isolated Power. RMS-to-DC Converter Senses High Voltage Input via Resistive Divider. A/D Converter Provides Digital Output Through Optoisolators. Accuracy is 1% in 200kHz Bandwidth

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AN106-4

Application Note 106

Low Distortion AC Line RMS Voltage Regulator

NOTE: BEFORE PROCEEDING ANY FURTHER, THE READER IS WARNED THAT CAUTION MUST BE USED IN THE CONSTRUCTION, TESTING AND USE OF THIS CIRCUIT. HIGH VOLTAGE, LETHAL POTENTIALS ARE PRESENT IN THIS CIRCUIT. EXTREME CAUTION MUST BE USED IN WORKING WITH, AND MAKING CONNECTIONS TO, THIS CIRCUIT. REPEAT: THIS CIRCUIT CONTAINS DANGEROUS, HIGH VOLTAGE POTENTIALS. USE CAUTION.

Almost all AC line voltage regulators rely on some form of waveform chopping, clipping or interruption to function. This is efficient, but introduces waveform distortion, which is unacceptable in some applications. Figure 8 regulates the AC line’s RMS value within 0.25% over wide input swings and does not introduce distortion. It does this by continuously controlling the conductivity of a series pass MOSFET in the AC lines path. Enclosing the MOSFET in a diode bridge permits it to operate during both AC line polarities.

AC SERIES PASS

AND CURRENT LIMIT

DANGER! Lethal Potentials Present—See Text

AC	1.5A
HIGH	SB
			+V REGULATORS
				170V
	470k		100k	Q4
	0.47µF			MPSA42
			BAT85	5V
	1N5281B		1N4690	0.1µF
	200V		5.6V
UNREGULATED
AC LINE
INPUT
	200k	Q3	OVERVOLTAGE
0.22µF	200k	2N5210 PROTECTION

AC

LOW

HEAT SINK IRF-840

*1% METAL FILM RESISTOR OPTODRIVER = TOSHIBA TLP190B

= 1N4005

				Q2	AC HIGH
				IRF-840
	0.03µF	430k		10k	Q6
		0.7Ω			2N3904
					2.2M*
10k	170V				FEEDBACK	VAC
					SENSE	REGULATED
2W
	ISOLATED	0.1µF	CONTROL			OUTPUT
	GATE BIAS		AMPLIFIER
	1µF		22k		RMS-TO-DC CONVERTER

				5V		C1	5V
										105VAC
				10k					1k
							+V
										TO 120VAC
				–		OUT		IN1
		5.6k					LTC1966			ADJUST
Q1				A1	1µF			IN2
						OUT RTN
2N3440				LT1077
						EN	–V	GND
				+						7.32k*
		REFERENCE				1N4689
5V	IN	0.4V OUT		330k	2.2µF	5.1V			AC LOW
		LT6650		1µF	MYLAR
		GND	FB
		1M		Q5
			0.22µF	2N4393
				100k
							AN106 F08
			SOFT-START/–VBIAS

Figure 8. Adjustable AC Line Voltage Regulator Introduces No Waveform Distortion. Line Voltage RMS Value is Sensed and Compared to a Reference by A1. A1 Biases Photovoltaic Optocoupler via Q1, Setting Q2-Diode Bridge Conductivity and Closing a Control Loop. VIN Must be ≥2V Above VOUT to Maintain Regulation

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AN106-5

Application Note 106

The AC line voltage is applied to the Q2-diode bridge. The Q2-diode bridge output is sensed by a calibrated variable voltage divider which feeds C1. C1’s output, representing the regulated lines RMS value, is routed to control amplifier A1 and compared to a reference. A1’s output biases Q1, controlling drive to a photovoltaic optoisolator. The optoisolator’s output voltage provides level-shifted bias to diode bridge enclosed Q2, closing a control loop which regulates the output’s RMS voltage against AC line and load shifts. RC components in A1’s local feedback path stabilize the control loop. The loop operates Q2 in its linear region, much like a common low voltage DC linear regulator. The result is absence of introduced distortion at the expense of lost power. Available output power is constrained by heat dissipation. For example, with the output adjustment set to regulate 10V below the normal input, Q2 dissipates about 10W at 100W output. This figure can be improved upon. The circuit regulates for VIN ≥ 2V above VOUT, but operation in this region risks regulation dropout as VIN varies.

CircuitdetailsincludeJFETQ5andassociatedcomponents. The passive components associated with Q5’s gate form a slow turn-on negative supply for C1. They also provide gate bias for Q5. Q5, a soft-start, prevents abrupt AC power application to the output at start-up. When power is off, Q5 conducts, holding A1’s “+” input low. When power is applied, A1 initially has a zero volt reference, causing the control loop to set the output at zero. As the 1MΩ 0.22µF combination charges, Q5’s gate moves negative, causing its channel conductivity to gradually decay. Q5 ramps off, A1’s positive input moves smoothly towards the LT6650’s 400mV reference, and the AC output similarly ascends towards its regulation point. Current sensor Q6, measuring across the 0.7Ω shunt, limits output current to about 1A. At normal line inputs (90VAC to 135VAC) Q4 supplies 5V operating bias to the circuit. If line voltage rises beyond this point, Q3 comes on, turning off Q4 and shutting down the circuit.

X1000 DC Stabilized Millivolt Preamplifier

The preceding circuits furnish high level inputs to the RMS converter. Many applications lack this advantage and some formofpreamplifierisrequired.Highgainpre-amplification for the RMS converter requires more attention than might be supposed. The preamplifier must have low offset error because the RMS converter (desirably) processes DC as

legitimate input. More subtly, the preamplifier must have far more bandwidth than is immediately apparent. The amplifiers –3db bandwidth is of interest, but its closed loop 1% amplitude error bandwidth must be high enough to maintain accuracy over the RMS converter’s 1% error passband. This is not trivial, as very high open-loop gain at the maximum frequency of interest is required to avoid inaccurate closed-loop gain.

Figure 9 shows an x1000 preamplifier which preserves the LTC1966’s DC-6kHz 1% accuracy. The amplifier may be either AC or DC coupled to the RMS converter. The 1mV full-scale input is split into high and low frequency paths. AC coupled A1 and A2 take a cascaded, high frequency gain of 1000. DC coupled, chopper stabilized A3 also has X1000 gain, but is restricted to DC and low frequency by its RC input filter. Assuming the switch is set to “DC + AC,” high and low frequency path information recombine at the RMS converter. The high frequency paths 650kHz –3db responsecombineswiththelowfrequencysectionsmicrovolt level offset to preserve the RMS converters DC-6kHz 1% error. If only AC response is desired, the switch is set to the appropriate position. The minimum processable input, set by the circuits noise floor, is 15µV.

Wideband Decade Ranged X1000 Preamplifier

The LTC1968, with a 500kHz, 1% error bandwidth, poses a significant challenge for an accurate preamplifier, but Figure 10 meets the requirement. This design features decade ranged gain to X1000 with a 1% error bandwidth beyond 500kHz, preserving the RMS converters 1% error bandwidth. Its 20µV noise floor maintains wideband performance at microvolt level inputs.

Q1A and Q1B form a low noise buffer, permitting high impedance inputs. A1 and A2, both gain switchable, take cascaded gain in accordance with the figure’s table. The gains are settable via reed relays controlled by a 2-bit code. A2’s output feeds the RMS converter and the converter’s output is smoothed by a Sallen-Keys active filter. The circuit maintains 1% error over a 10Hz to 500kHz bandwidth at all gains due to the preamplifiers –3db, 10MHz bandwidth. The 10Hz low frequency restriction could be eliminated with a DC stabilization path similar to Figure 9’s but its gain would have to be switched in concert with the A1-A2 path.

an106f

AN106-6