Analog Devices AD6600ST-PCB, AD6600AST Datasheet

Dual Channel, Gain-Ranging

a

FEATURES
Dual IF Inputs, 70 MHz–250 MHz

Diversity or Two Independent IF Signals
Separate Attenuation Paths

Oversample RF Channels

20 MSPS on a Single Carrier
10 MSPS/Channel in Diversity Mode

Total Signal Range 90+ dB

30 dB from Automatic Gain-Ranging (AGC)
60 dB from A/D Converter
Range >100 dB After Processing Gain

Digital Outputs

11-Bit ADC Word
3-Bit RSSI Word
2 Clock, A/B Indicator

Single 5 V Power Supply

Output DVCC 3.3 V or 5 V

775 mW Power Dissipation

APPLICATIONS
Communications Receivers
PCS/Cellular Base Stations

GSM, CDMA, TDMA

Wireless Local Loop, Fixed Access

PRODUCT DESCRIPTION

The AD6600 mixed-signal receiver chip directly samples signals
at analog input frequencies up to 250 MHz. The device includes

ADC with RSSI

AD6600

two input channels, each with 1 GHz input amplifiers and
30 dB of automatic gain-ranging circuitry. Both channels are
sampled with a 450 MHz track-and-hold followed by an 11-bit,
20 MSPS analog-to-digital converter. Digital RSSI outputs, an
A/B channel indicator, a 2× Clock output, references, and con­trol circuitry are all on-chip. Digital output signals are two’s
complement, CMOS-compatible and interface directly to

3.3 V or 5 V digital processing chips.

The primary use for the dual analog input structure is sampling
both antennas in a two-antenna diversity receiver. However,
Channels A and B may also be used to sample two independent
IF signals. Diversity, or dual-channel mode, is limited to 10 MSPS
per channel. In single-channel mode, the full clock rate of
20 MSPS may be applied to a single carrier.

The AD6600 may be used as a stand-alone sampling chip, or it
may be combined with the AD6620 Digital Receive Signal Pro­cessor. The AD6620 provides 10 dB–25 dB of additional pro­cessing gain before passing data to a fixed- or floating-point DSP.

Driving the AD6600 is simplified by using the AD6630 differen­tial IF amplifier. The AD6630 is easily matched to inexpensive
SAW filters from 70 MHz to 250 MHz.

Designed specifically for cellular/PCS receivers, the AD6600
supports GSM, IS-136, CDMA and Wireless LANs, as well as
proprietary air interfaces used in WLL/fixed-access systems.

Units are available in plastic, surface-mount packages (44-lead
LQFP) and specified over the industrial temperature range
(–40°C to +85°C).

FUNCTIONAL BLOCK DIAGRAM

0dB, –12dB, –24dB

AIN

AIN

DETECT

BIN

BIN

ATTEN

SET

PEAK

RSSI

ATTEN

0dB, –12dB, –24dB

GAIN

3

GAIN

RSSI

+12, +18dB

ANALOG MUX

SELECT GAIN

GAIN

AD6600

REV. 0

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

NOISE FILTER

FLT FLT

630

ENCODE

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] World Wide Web Site: http://www.analog.com
Fax: [phone] © Analog Devices, Inc., 2000

RESONANT

ENCODE

PORT

A/D

CONVERTER

TIMING

TWO'S

COMPLEMENT

11

3

RSSI

DVCCENCENCGNDAVCCB_SELA_SEL

AB_OUT

D10–D0

RSSI [2:0]

CLK2

AD6600–SPECIFICATIONS

DC SPECIFICATIONS

(AVCC = 5 V, DVCC = 3.3 V; T

= –40C, T

MIN

= +85C unless otherwise noted.)

MAX

Test AD6600AST

Parameter Temp Level Min Typ Max Unit

ANALOG INPUTS (AIN, AIN/BIN, BIN)

Differential Analog Input Voltage Range
Differential Analog Input Resistance

1

2

Full V 2.0 V p-p
Full IV 160 200 240 Ω

Differential Analog Input Capacitance 25°C V 1.5 pF

PEAK DETECTOR (Internal), RSSI

Resolution 3 Bits
RSSI Gain Step Full V 6 dB
RSSI Hysteresis

3

Full V 6 dB

RESONANT PORT (FLT, FLT)

Differential Port Resistance Full V 630 Ω
Differential Port Capacitance Full V 1.75 pF

A/D CONVERTER

Resolution Full IV 11 Bits

ENCODE INPUTS (ENC, ENC)

Differential Input Voltage (AC-Coupled)

4

Full IV 0.4 V p-p

Differential Input Resistance 25°CV 11 kΩ
Differential Input Capacitance 25°C V 2.5 pF

A/B MODE INPUTS (A_SEL, B_SEL)

5

Input High Voltage Range Full IV 4.75 5.25 V
Input Low Voltage Range Full IV 0.0 0.5 V

POWER SUPPLY

Supply Voltages

AVCC Full II 4.75 5.0 5.25 V
DVCC Full IV 3.0 3.3 5.25 V

Supply Current

I

(AVCC = 5.0 V) Full II 145 182 mA

AVCC

I

(DVCC = 3.3 V) Full II 15 20 mA

DVCC

POWER CONSUMPTION

NOTES

1

Analog Input Range is a function of input frequency. See ac specifications for 70 MHz–250 MHz inputs.

2

Analog Input Impedance is a function of input frequency. See ac specifications for 70 MHz–450 MHz inputs.

3

Six dB of digital hysteresis is used to eliminate level uncertainty at the RSSI threshold points due to noise and amplitude variations.

4

Encode inputs should be ac-coupled and driven differentially. See Encoding the AD6600 for details.

5

A_SEL and B_SEL should be tied directly to ground or AVCC.

6

Maximum power consumption is computed as maximum current at nominal supplies.

Specifications subject to change without notice.

6

Full II 775 976 mW

DIGITAL SPECIFICATIONS

(AVCC = 5 V, DVCC = 3.3 V; T

= –40C, T

MIN

= +85C unless otherwise noted.)

MAX

Test AD6600AST

Parameter Temp Level Min Typ Max Unit

LOGIC OUTPUTS (D10–D0, AB_OUT, RSSI2–0)

1

Logic Compatibility CMOS
Logic “1” Voltage (DVCC = 3.3 V) Full II 2.8 DVCC – 0.2 V
Logic “0” Voltage (DVCC = 3.3 V) Full II 0.2 0.5 V
Logic “1” Voltage (DVCC = 5.0 V) Full IV 4.0 DVCC – 0.35 V
Logic “0” Voltage (DVCC = 5.0 V) Full IV 0.35 0.5 V
Output Coding (D10–D0) Two’s Complement

CLK2× OUTPUT

1, 2

Logic “1” Voltage (DVCC = 3.3 V) Full II 2.8 DVCC – 0.2 V
Logic “0” Voltage (DVCC = 3.3 V) Full II 0.2 0.5 V
Logic “1” Voltage (DVCC = 5.0 V) Full IV 4.0 DVCC – 0.3 V
Logic “0” Voltage (DVCC = 5.0 V) Full IV 0.35 0.5 V

NOTES

1

Digital output load is one LCX gate.

2

CLK2× output voltage levels, high and low, tested at switching rate of 10 MHz.

Specifications subject to change without notice.

–2–

REV. 0

AD6600

TIMING REQUIREMENTS AND SWITCHING SPECIFICATIONS

(AVCC = 5 V, DVCC = 3.3 V; ENC and ENC = 20 MSPS; T

Parameter Name Temp Level Min Typ Max Unit

A/D CONVERTER

Conversion Rate f
Maximum Conversion Rate Full II 20 MSPS
Minimum Conversion Rate Full IV 6 MSPS
Aperture Uncertainty t

ENCODE INPUTS (ENC, ENC)

Period t
Pulsewidth High
Pulsewidth Low

3

4

2× CLOCK OUTPUT (CLK2×)

Output Frequency 2× f
Output Period

CLK2× Pulsewidth Low
Output Risetime
Output Falltime

6

6

7

7

OUTPUT RISE/FALL TIMES

2

t
t

5

t
t
t

8

Output Risetime (D10:D0, RSSI2:0) Full V 8 ns
Output Falltime (D10:D0, RSSI2:0) Full V 8.4 ns
Output Risetime (AB_OUT) Full V 6 ns
Output Falltime (AB_OUT) Full V 6.2 ns

NOTES

1

See AD6600 Timing Diagrams.

2

All switching specifications tested by driving ENC and ENC differentially.

3

Several timing specifications are a function of Encode high time, t
should be kept as close to 50% as possible.

4

Encode pulse low directly affects the amount of settling time available at FLT resonant port. See External Analog (Resonant) Filter section for details.

5

The 2× Clock is generated internally, therefore some specifications are functions of encode period and duty cycle. All timing measurements to or from CLK2 × are
referenced to 2.0 V crossing.

6

This specification IS a function of Encode period and duty cycle; reference timing diagrams Figure 8.

7

Output rise time is measured from 20% point to 80% point of total CLK2× voltage swing; output fall time is measured from 80% point to 20% point of total CLK2×
voltage swing.

8

Output rise time is measured from 20% point to 80% point of total data voltage swing; output fall time is measured from 80% point to 20% point of total data voltage
swing. All outputs specified with 10 pF load.

Specifications subject to change without notice.

MIN

ENC

j

ENC

ENCH

ENCL

CLK2×_1
CLK2×_2
CLK2×L

= –40C, T

= +85C unless otherwise noted.)

MAX

Test AD6600AST

25°C V 0.3 ps rms

Full II 50 ns
Full IV 20 ns
Full IV 20 ns

Full V t
Full V t
Full V t
Full V 3 ns
Full V 2.6 ns

; these specifications are shown in the data tables and timing diagrams. Encode duty cycle

ENCH

1

1/(t

) MSPS

ENC

ENC

ENCL

ENCH

/2 ns

ENCH

MSPS
ns
ns

REV. 0 –3–

AD6600–SPECIFICATIONS

TIMING REQUIREMENTS AND SWITCHING SPECIFICATIONS

(AVCC = 5 V, DVCC = 3.3 V; ENC and ENC = 20 MSPS, Duty Cycle = 50%; T

Parameter Name Temp Level Min Typ Max Unit

ENCODE/CLK2×

Encode Rising to CLK2× Falling
Encode Rising to CLK2× Rising

3

4

t

CF

t

CR

@ Encode = 13 MSPS, 50% Duty Cycle Full IV 25.7 27.2 28.7 ns
@ Encode = 20 MSPS, 50% Duty Cycle Full IV 19.0 20.5 22.0 ns

CLK2×/DATA (D10:0, RSSI2:0)

CLK2× to DATA Rising Low Delay
CLK2× to DATA Hold Time
CLK2× to DATA Falling Low

CLK2× to DATA Setup Time

@ Encode = 13 MSPS, 50% Duty Cycle Full IV 16.5 23.0 ns
@ Encode = 20 MSPS, 50% Duty Cycle

CLK2×/AB_OUT

5

CLK2× to AB_OUT Rising Low Delay
CLK2× to AB_OUT Hold Time
CLK2× to AB_OUT Falling Low Delay

CLK2× to AB_OUT Setup Time

@ Encode = 13 MSPS, 50% Duty Cycle Full IV 12.5 19.5 ns
@ Encode = 20 MSPS, 50% Duty Cycle

ENCODE/DATA (D10:0, RSSI2:0)

ENCODE to DATA Rising Low Delay
ENCODE to DATA Hold Time

5

3

3

3, 6

4

3

3

3, 6

4

4

4

t

2×_DRL

t

H_D2×

t

2×_DFL

t

S_D2×

6

t

2×_ARL

t

H_A2×

t

2×_AFL

t

S_A2×

6

t

EN_DRL

t

H_DEN

@ Encode = 13 MSPS, 50% Duty Cycle Full IV 28.7 33.7 ns

@ Encode = 20 MSPS, 50% Duty Cycle Full IV 22.0 27.0 ns
ENCODE to DATA Falling Low Delay
ENCODE to DATA Delay (Setup)

@ Encode = 13 MSPS, 50% Duty Cycle Full IV 26.2 34.2 ns

@ Encode = 20 MSPS, 50% Duty Cycle

4

4

t

EN_DFL

t

S_DEN

6

ENCODE/AB_OUT

ENCODE to AB_OUT Rising Low Delay4t
ENCODE to AB_OUT Delay (Hold)

4

EN_ARL

t

H_AEN

@ Encode = 13 MSPS, 50% Duty Cycle Full IV 32.7 38.2 ns

@ Encode = 20 MSPS, 50% Duty Cycle Full IV 26.0 31.5 ns
ENCODE to AB_OUT Falling Low Delay
ENCODE to AB_OUT Delay (Setup)

@ Encode = 13 MSPS, 50% Duty Cycle Full IV 22.2 30.7 ns

@ Encode = 20 MSPS, 50% Duty Cycle

NOTES

1

See AD6600 Timing Diagrams.

2

All switching specifications tested by driving ENC and ENC differentially.

3

This specification IS NOT a function of Encode period and duty cycle.

4

This specification IS a function of Encode period and duty cycle.

5

CLK2× referenced to 2.0 V crossing; digital output levels referenced to 0.8 V and 2.0 V crossings; all outputs with 10 pF load.

6

For these particular specifications, the 25°C specification is valid from 25°C to 85°C. The Full temperature specification includes cold temperature extreme and
covers the entire range, –40°C to +85°C.

Specifications subject to change without notice.

4

t

4

EN_AFL

t

S_AEN

6

= –40C, T

MIN

Test AD6600AST

Full IV 6.5 8.0 9.5 ns
Full IV tCF + (t

Full IV 3.0 6.5 ns
Full IV 3.0 6.5 ns
25°C IV 10.0 15.0 20.0 ns
Full IV 11.0 15.5 22.0 ns
Full IV t

25°C IV 5.0 10.0 ns
Full IV 3.0 9.5 ns

Full IV 7.0 11.0 ns
Full IV 7.0 11.0 ns
25°C IV 12.0 18.0 23.0 ns
Full IV 10.7 19.0 26.0 ns
Full IV t

25°C IV 2.0 7.0 ns
Full IV –1.0 6.0 ns

Full IV tCR + t
Full IV t

Full IV tCR + t
Full IV t

25°C IV 8.0 14.5 ns
Full IV 6.0 14.0 ns

Full IV tCR + t
Full IV t

Full IV tCR + t
Full IV t

25°C IV 5.0 11.5 ns
Full IV 2.0 10.5 ns

1, 2

= +85C unless otherwise noted.)

MAX

ENCH

– t

ENCH

– t

ENCH

EN_DRL

– t

ENC

EN_ARL

– t

ENC

2×_DFL

2×_AFL

2×_DRL

2×_DFL

EN_DFL

2×_ARL

2×_AFL

EN_AFL

)/2 ns

ns

ns

ns
ns

ns
ns

ns
ns

ns
ns

–4–

REV. 0

AD6600

(AVCC = 5 V, DVCC = 3.3 V; ENC and ENC = 20 MSPS, Duty Cycle = 50%; T

AC SPECIFICATIONS

otherwise noted.)

Test AD6600AST

Parameter Temp Level Min Typ Max Unit

ANALOG INPUTS

Analog Input 3 dB Bandwidth

1

2

Full V 450 MHz

Differential Analog Input Voltage Range

70 MHz Full V 2.45 V p-p
150 MHz Full V 2.57 V p-p
200 MHz Full V 2.62 V p-p
250 MHz Full V 2.86 V p-p

Differential Analog Input Impedance

3

70 MHz 25°C V 197–j24 Ω
150 MHz 25°C V 188–j48 Ω
200 MHz 25°C V 175–j57 Ω
250 MHz 25°C V 161–j67 Ω
300 MHz 25°C V 151–j73 Ω
350 MHz 25°C V 140–j80 Ω
400 MHz 25°C V 141–j75 Ω
450 MHz 25°C V 173–j107 Ω

Full-Scale Input Power

70 MHz Full V 5.8 dBm
150 MHz Full V 6.3 dBm
200 MHz Full V 6.7 dBm
250 MHz Full V 7.7 dBm

Full-Scale Gain Tolerance

70 MHz–250 MHz Full V ± 0.5 dB
200 MHz

5

4

25°C I –1.0 ±0.1 +1.0 dB

Gain Error

AIN = 200 MHz

@ –76 dBFS 25°C I –1.5 +1.5 dB

Gain Matching (Input A:B)

70 MHz–250 MHz Full V ± 0.1 dB
200 MHz Full II –0.5 ±0.05 +0.5 dB

Range-to-Range Gain Tolerance

70 MHz–250 MHz Full V ± 0.1 dB

Range-to-Range Phase Tolerance

70 MHz Full V 0.2 Degree
250 MHz Full V 0.5 Degree

Channel Isolation

70 MHz–250 MHz Full IV 45 50 dB

7

Noise

6

Minimum Attenuation Level Full V 34 µV rms
Maximum Attenuation Level Full V 869 µV rms

Attenuator 3OIP
Signal-to-Noise Ratio (SNR)

8

9, 10, 11

Full V +33 dBm

AIN = 70 MHz

@ –1 dBFS 25°CIV5559 dB
@ –6 dBFS 25°C V 54.5 dB
@ –10 dBFS 25°CIV4549 dB
@ –12 dBFS to –42 dBFS 25°C IV 41 48 ±6dB
@ –54 dBFS 25°CIV3134 dB

AIN = 150 MHz

@ –1 dBFS 25°CIV5558 dB
@ –6 dBFS 25°CV 54 dB
@ –10 dBFS 25°CIV4549 dB
@ –12 dBFS to –42 dBFS 25°C IV 41 48 ±6dB
@ –54 dBFS 25°CIV3134 dB

= –40C, T

MIN

= +85C unless

MAX

REV. 0 –5–

AD6600–SPECIFICATIONS

AC SPECIFICATIONS (continued)

Test AD6600AST

Parameter Temp Level Min Typ Max Unit

ANALOG INPUTS (Continued)

Signal-to-Noise Ratio (Continued)

AIN = 200 MHz

@ –1 dBFS 25°C I 55 57.5 dB
@ –6 dBFS 25°C V 53.5 dB
@ –10 dBFS 25°C I 45 49 dB
@ –12 dBFS to –42 dBFS 25°C I 40.5 48 ± 6dB
@ –54 dBFS 25°C I 31 34 dB

AIN = 250 MHz

@ –1 dBFS 25°CIV5256 dB
@ –6 dBFS 25°C V 53.5 dB
@ –10 dBFS 25°CIV4349 dB
@ –12 dBFS to –42 dBFS 25°C IV 40 48 ±6dB
@ –54 dBFS 25°CIV3034 dB

SECOND HARMONIC

AIN = 70 MHz

@ –1 dBFS Full V 69 dBc

@ –6 dBFS Full V 68 dBc

@ –12 dBFS to –42 dBFS Full V 68 ±6 dBc
AIN = 150 MHz

@ –1 dBFS Full V 60 dBc

@ –6 dBFS Full V 59 dBc

@ –12 dBFS to –42 dBFS Full V 67 ±6 dBc
AIN = 200 MHz

@ –1 dBFS 25°C I 50 60 dBc

@ –6 dBFS Full V 56 dBc

@ –10 dBFS 25°C I 48 55 dBc

@ –12 dBFS to –42 dBFS Full V 65 ±6 dBc

@ –54 dBFS Full V 50 dBc
AIN = 250 MHz

@ –1 dBFS Full V 54 dBc

@ –6 dBFS Full V 62 dBc

@ –12 dBFS to –42 dBFS Full V 65 ±6 dBc

THIRD HARMONIC

AIN = 70 MHz

@ –1 dBFS Full V 77 dBc

@ –6 dBFS Full V 76 dBc

@ –12 dBFS to –42 dBFS Full V 67 ±6 dBc
AIN = 150 MHz

@ –1 dBFS Full V 65 dBc

@ –6 dBFS Full V 70 dBc

@ –12 dBFS to –42 dBFS Full V 66 ±6 dBc
AIN = 200 MHz

@ –1 dBFS 25°C I 50 55 dBc

@ –6 dBFS Full V 58 dBc

@ –10 dBFS 25°C I 55 66 dBc

@ –12 dBFS to –42 dBFS Full V 65 ±6 dBc

@ –54 dBFS Full V 62 dBc
AIN = 250 MHz

@ –1 dBFS Full V 50 dBc

@ –6 dBFS Full V 56 dBc

@ –12 dBFS to –42 dBFS Full V 65 ±6 dBc
AIN = 70 MHz–250 MHz

@ –75 dBFS Full IV 28 35 dBc

9, 10, 11

9, 10, 11

–6–

REV. 0

AD6600

AC SPECIFICATIONS (continued)

Test AD6600AST

Parameter Temp Level Min Typ Max Unit

WORST OTHER SPUR (4th or Higher)

AIN = 70 MHz

@ –1 dBFS Full V 74.5 dBc
@ –6 dBFS Full V 71 dBc
@ –12 dBFS to –42 dBFS Full V 68 ±6 dBc

AIN = 150 MHz

@ –1 dBFS Full V 67 dBc
@ –6 dBFS Full V 65 dBc
@ –12 dBFS to –42 dBFS Full V 67 ±6 dBc

AIN = 200 MHz

@ –1 dBFS 25°C I 60 67 dBc
@ –6 dBFS Full V 66 dBc
@ –10 dBFS 25°C I 55 66 dBc
@ –12 dBFS to –42 dBFS Full V 65 ±6 dBc

AIN = 250 MHz

@ –1 dBFS Full V 66.5 dBc
@ –6 dBFS Full V 65 dBc
@ –12 dBFS to –42 dBFS Full V 65 ±6 dBc

NOTES

1

AIN, AIN/BIN, BIN: The AD6600 analog inputs are unconditionally stable and guarantee proper operation over the 70 MHz–250 MHz specified operating range.

Circuit board layout is critical on this device, and proper PCB layout must be employed to achieve specified results.

2

Analog Input 3 dB Bandwidth is determined by internal track-and-hold. The front-end attenuators have a bandwidth of 1 GHz.

3

Measured real and imaginary values using Network Analyzer.

4

Full-scale gain tolerance is the typical variation in gain at a given IF input frequency. The nominal value for full-scale input power is a function of frequency as

shown in previous specification.

5

Full-scale gain tolerance measured at 200 MHz analog input referenced to 6.7 dBm nominal full-scale input power. For the gain measurement test, the input signal

level is set to –6 dBFS. Tuning port bandwidth is set to 50 MHz.

6

Main channel set to full-scale input power. Diversity channel swept from –20 dBFS to –90 dBFS.

7

Measurement includes thermal and quantization noise at 70 MHz analog input. Tuning port bandwidth is set to 50 MHz.

8

Test tones at 160.05 MHz and 170.05 MHz.

9

Measurements at –1 dFBS, –6 dBFS, and –10 dBFS are in highest attenuation mode, RSSI = 101.

10

Each gain-range is checked at ~3 dB from RSSI trip point (not in hysteresis); nominally –16 dBFS (RSSI = 100), –22 dBFS (RSSI = 011), –28 dBFS (RSSI = 010),

–35 dBFS (RSSI = 001).

11

Measurement at –54 dBFS is in the lowest attenuation mode, RSSI = 000.

Specifications subject to change without notice.

REV. 0 –7–

AD6600

WARNING!

ESD SENSITIVE DEVICE

ABSOLUTE MAXIMUM RATINGS

1

Parameter Min Max Unit

ELECTRICAL

AVCC Voltage 0 7 V
DVCC Voltage 0 7 V
Analog Input Voltage
Analog Input Current
Digital Input Voltage
Output Current
Resonant Port Voltage

ENVIRONMENTAL

2

2

3

4

5

6

0 AVCC V

25 mA

0 AVCC V

4mA

0 AVCC V

Operating Temperature Range

(Ambient) –40 +85 °C
Maximum Junction Temperature 150 °C
Lead Temperature (Soldering, 10 sec) 300 °C
Storage Temperature Range (Ambient) –65 +150 °C

NOTES

1

Absolute maximum ratings are limiting values to be applied individually, and
beyond which the serviceability of the circuit may be impaired. Functional
operability is not necessarily implied. Exposure to absolute maximum rating
conditions for an extended period of time may affect device reliability.

2

Pins AIN, AIN, BIN, BIN.

3

Pins ENC, ENC, A_SEL, B_SEL.

4

Pins D10:0, RSSI2:0, AB_OUT, CLK2×.

5

Pins FLT, FLT.

6

Typical thermal impedance (44-lead LQFP); θJC = 16°C/W, θJA = 55°C/W.

EXPLANATION OF TEST LEVELS
Test Level

I. 100% Production Tested.
II. 100% Production Tested at 25°C and guaranteed by design

and characterization at temperature extremes.

IV. Parameter is guaranteed by design and characterization

testing.

V. Parameter is a typical value only.

ORDERING GUIDE

Temperature Package Package

Model Range Description Option

AD6600AST –40°C to 44-Terminal LQFP ST-44

+85°C (Low-Profile Quad
(Ambient) Plastic Flatpack)

AD6600ST/PCB Evaluation Board

with AD6600AST

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the AD6600 features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are recom­mended to avoid performance degradation or loss of functionality.

–8–

REV. 0

AD6600

PIN FUNCTION DESCRIPTIONS

Pin Number Name Function

1, 33 DVCC Digital VCC for Digital Outputs. Can be 3.3 V.
2, 5, 13, 19, 21, 24, 30, 32 GND Ground.
3 C1 Internal Bias Point. Bypass by 0.01 µF to GND.
4, 14, 15, 18, 20, 25, 31 AVCC 5 V Power Supply.
6–8 RSSI[2:0] RSSI Digital Output Bits.
9, 10 B_SEL, A_SEL Mode Select Pins for Analog Input Channel A and B Sampling.
11 AIN True Analog Input Channel A.
12 AIN Complementary Analog Input Channel A.
16, 17 FLT, FLT Resonant Filter Pins for External LC Noise Filter.
22 BIN Complementary Analog Input Channel B.
23 BIN True Analog Input Channel B.
26 ENC Complementary Encode Input.
27 ENC True Encode Input.
28 CLK2× 2× Clock Output Used for Clocking Digital Filter Chips.
29 AB_OUT Digital Output Flag Indicating Whether Output Is Input A (High) or B (Low).
34 D0 Digital Data Output Bit (Least Significant Bit)*.
35–43 D1–D9 Digital Data Output Bits*.
44 D10 Digital Data Output Bit (Most Significant Bit)*.

*Digital Outputs (D10:D0) in Two’s Complement Format.

PIN CONFIGURATION

DVCC

GND

AVCC

GND

RSSI2

RSSI1

RSSI0

B_SEL

A_SEL

AIN

D8

D6

D9

GND

D7

40 39 3841424344 36 35 3437

TOP VIEW

(Not to Scale)

AVCC

AVCC

AD6600

FLT

D10 (MSB)

1

PIN 1
IDENTIFIER

2

3

C1

4

5

6

7

8

9

10

11

121314 15 1 6 17 18 192021 22

AIN

D5

FLT

D4

AVCC

D3

GND

D2

AVCC

D1

GND

D0 (LSB)

33

32

31

30

29

28

27

26

25

24

23

BIN

DVCC

GND

AVCC

GND

AB_OUT

CLK2

ENC

ENC

AVCC

GND

BIN

REV. 0 –9–

AD6600

DEFINITIONS OF SPECIFICATIONS
Analog Bandwidth

The analog input frequency at which the spectral power of the
fundamental frequency (as determined by the FFT analysis) is
reduced by 3 dB. The bandwidth is determined by the internal
track-and-hold when the filter node is resonated.

Aperture Delay

The delay between the 50% point of the rising edge of the
ENCODE command and the instant at which the analog input­is sampled.

Aperture Uncertainty (Jitter)

The sample-to-sample variation in aperture delay.

Attenuator 3OIP

The third order intercept point of the front end of the AD6600.
It is the point at which the third order products would theoreti­cally intercept the input signal level if the input level could increase
without bounds. This is measured using the ADC within the
AD6600 while the input is stimulated with dual tones in the
minimum attenuation (i.e., maximum gain) range.

Channel Isolation

The amount of signal leakage from one channel to the next
when one channel is driven with a full-scale input, and the other
channel is swept from –20 dBFS to –90 dBFS with a frequency
offset. The leakage is measured on the side with the smaller signal.

Differential Analog Input Resistance, Differential Analog
Input Capacitance and Differential Analog Input Impedance

The real and complex impedances measured at each analog
input port. The resistance is measured statically and the capaci­tance and differential input impedances are measured with a
network analyzer.

Differential Analog Input Voltage Range

The peak-to-peak differential voltage that must be applied to the
converter to generate a full-scale response. Peak differential
voltage is computed by observing the voltage on a single pin and
subtracting the voltage from the other pin, which is 180 degrees
out of phase. Peak-to-peak differential is computed by rotating
the inputs phase 180 degrees and taking the peak measurement
again. The difference is then computed between both peak
measurements.

Differential Nonlinearity

The deviation of any code width from an ideal 1 LSB step.

Differential Resonant Port Resistance

The resistance shunted across the resonant port (nominally
630 Ω). Used to determine the filter bandwidth and gain of
that stage.

Encode Pulsewidth/Duty Cycle

Pulsewidth high is the minimum amount of time that the
ENCODE pulse should be left in logic “1” state to achieve rated
performance; pulsewidth low is the minimum time ENCODE
pulse should be left in low state. See timing implications of
changing t

in text. At a given clock rate, these specifications

ENCH

define an acceptable Encode duty cycle.

Full-Scale Gain Tolerance

Unit-to-unit variation in full-scale input power.

Full-Scale Input Power

Expressed in dBm. Computed using the following equation:

2

Power

FULL SCALE

Gain Matching (Input A:B)

=

10

log



V

FULL SCALE rms



Z







INPUT

0 001

.










Variation in full-scale power between A and B inputs.

Harmonic Distortion, 2nd

The ratio of the rms signal amplitude to the rms value of the
second harmonic component, reported in dBc.

Harmonic Distortion, 3rd

The ratio of the rms signal amplitude to the rms value of the
third harmonic component, reported in dBc.

Integral Nonlinearity

The deviation of the transfer function from a reference line
measured in fractions of 1 LSB using a “best straight line”
determined by a least square curve fit.

Minimum Conversion Rate

The encode rate at which the SNR of the lowest analog signal fre­quency drops by no more than 3 dB below the guaranteed limit.

Maximum Conversion Rate

The encode rate at which parametric testing is performed.

Noise (For Any Range Within the ADC)

VZ

=× ×

0 001 10

NOISE

.



−−

FS SNR Signal

dBm dBc dBFS




10






where:

Z is the input impedance,
FS is the full-scale of the device for the frequency in question,
SNR is the value for the particular input level,
Signal is the signal level within the ADC reported in dB below

full scale. This value includes both thermal and quanti­zation noise.

Range-Range Gain Tolerance

The gain error in the RSSI attenuator ladder from one range to
the next.

Range-Range Phase Tolerance

The phase error in the RSSI attenuator ladder from one range
to the next.

Differential Resonant Port Capacitance

The capacitance between the two resonant pins. Used to deter­mine filter bandwidth and resonant frequency.

–10–

REV. 0

AD6600

RSSI Gain Step

The input amplitude span between taps of the RSSI (received
signal strength) attenuator ladder. Ideally each stage should
span 6 dB of input power.

RSSI Hysteresis

The amount of movement in the RSSI switch points, depending
on the direction of approach. Hysteresis prevents unnecessary
RSSI toggling when input signal power is near a threshold.

Signal-to-Noise Ratio (Without Harmonics)

The ratio of the rms signal amplitude (set at 1 dB below full
scale) to the rms value of the sum of all other spectral compo­nents, excluding the first five harmonics and dc.

Worst Other Spur

The ratio of the rms signal amplitude to the rms value of the
worst spurious component (excluding the second and third
harmonic) reported in dBc.

AD6600 TRANSFER FUNCTION

60

54

48

42

36

30

SNR – dB

24

18

12

6

0

–100

–80 –70 –60 –50 –40 –30 –20 –10

–90

AIN LEVEL – dBFS

Figure 1. SNR vs. Input Power

0

REV. 0

–11–

AD6600

EQUIVALENT CIRCUITS

AVCC

AIN

GND

AVCC

AIN

GND

ATTENUATOR STAGE

EQUIVALENT INPUT R

SHOWN ONLY

100

BUF GAINBUF

100

4/8 GAIN STAGE

V

REF

Figure 2. Analog Input Stage (Channel A Shown;
Channel B Is Equivalent)

AVCC

AVCC

A_SEL

GND

ISEL_A ISEL_B

BIAS

AVCC

B_SEL

GND

AVCC

GND

GND

AVCC

AVCC

GND

FROM

GAIN STAGE

EXTERNAL LC FILTER

FLT FLT

315 315

Figure 5. Resonant (LC Noise Filter) Port

AVCC

AVCC

R1

17k

1/2

ENCODE ENCODE

8k

1/2

R2

TIMING

CIRCUITS

R1
17k
1/2

R2
8k
1/2

AVCC

TO T/H

GND

Figure 3. A_SEL, B_SEL Input Mode Pins

DVCC

CURRENT

MIRROR

DVCC

V

CURRENT

MIRROR

REF

500

D10–D0
RSSI [2:0]

Figure 4. Digital Outputs

Figure 6. Encode Inputs

DVCC

CURRENT

MIRROR

DVCC

V

CURRENT

MIRROR

REF

CLK2
AB_OUT

Figure 7. CLK2⫻, AB_OUT Outputs

–12–

REV. 0

AD6600 TIMING DIAGRAMS

AD6600

ENCODE

CLK2

D [10:0]

RSSI [2:0]

AB_OUT

ENCODE

t

ENCH

t

CR1

t

t

CLK2L

CF1

CLK22 CLK21 CLK22 CLK21 CLK22

t

t

21_DRL

21_DFL

t

21_ARL

t

ENCL

t

CR2

t

t

CLK2L

CF2

t

CLK2H2

t

CLK22

t

ENC

t

CLK2H1

t

21_AFL

t

CLK21

Figure 8. Encode to CLK2⫻ Delays and CLK2⫻ Propagation Delays

t

ENCH

t

CR1

t

t

CLK2L

CF1

t

ENCL

t

CR2

t

t

CLK2L

CF2

t

CLK2H2

t

CLK22

t

ENC

t

CLK2H1

t

CLK21

CLK2

D [10:0]

RSSI [2:0]

AB_OUT

ENCODE

CLK2

D [10:0]

RSSI [2:0]

AB_OUT

CLK22 CLK21 CLK22 CLK21 CLK22

t

H_D2

t

H_A2

t

S_D2

t

S_A2

t

H_D2

t

H_A2

t

S_D2

t

S_A2

Figure 9. CLK2⫻ Setup-and-Hold Time Characteristics

t

ENCH

ENCODE ENCODE ENCODE

t

CR1

t

t

CLK22 CLK21 CLK22 CLK21 CLK22

CF1

CLK2L

t

EN_DRL

t

t

EN_DFL

EN_ARL

t

ENCL

t

CR2

t

t

CF2

CLK2L

t

CLK2H2

t

CLK22

t

EN_AFL

t

ENC

t

CLK2H1

t

CLK21

REV. 0

Figure 10. Encode to CLK2⫻ Delays and Encode Propagation Delays

–13–

AD6600

ENCODE

CLK2

D [10:0]

RSSI [2:0]

AB_OUT

t

ENCH

ENCODE ENCODE ENCODE

t

CR1

CF1

t

CLK2L

t

H_DEN

t

H_AEN

t

CLK22 CLK21 CLK22 CLK21 CLK22

t

ENCL

t

CR2

t

t

CF2

CLK2L

t

S_DEN

t

S_AEN

t

CLK2H2

t

CLK22

t

H_DEN

t

H_AEN

t

ENC

t

CLK2H1

t

CLK21

t

S_DEN

t

S_AEN

Figure 11. Encode Setup-and-Hold Time Characteristics

3 2.6

CLK2

8 8.4

D [10:0]

RSSI [2:0]

ENCODE

CLK2

ENCODE

CLK2

AB_OUT

Figure 12. Typical Output Rise and Fall Times

20 30 50

40%

8 8

Figure 13. Encode = 20 MSPS, Duty Cycle = 40%

60%

8 8

6

2030 50

2323

20 30

6.2

20301818

Figure 14. Encode = 20 MSPS, Duty Cycle = 60%

–14–

REV. 0

AIN

SNR – dB

–12

–48

24

28

AIN – dBFS

32 36 40 44 48 52 56 60

–18

–42

–54

–30
–36

–60

–24

–66

–72

–78

–84

–90

–96

0

–90

201612840

12dB SNR WINDOW

101

100

011

010

001

000

101

100

011

010

001

000

AIN

BIN

BIN

0dB, –12dB, –24dB

ATTEN

DETECT

SET

PEAK

RSSI

ATTEN

0dB, –12dB, –24dB

GAIN

3

GAIN

RSSI

+12, +18dB

ANALOG MUX

SELECT GAIN

FLT FLT

GAIN

AD6600

NOISE FILTER

630

ENCODE

RESONANT

ENCODE

PORT

A/D

CONVERTER

TIMING

TWO'S

COMPLEMENT

11

RSSI

AD6600

AB_OUT

D10–D0

3

RSSI [2:0]

CLK2

B_SELA_SEL

AVCC

Figure 15. Functional Block Diagram

THEORY OF OPERATION

The AD6600, dual-channel, gain-ranging ADC integrates ana­log IF circuitry with high speed data conversion. Each analog
input stage is a 1 GHz, 0 dB to –24 dB, phase-compensated step
attenuator; the step size in each attenuator is 12 dB. Both input
stages drive an analog multiplex function followed by a 12 dB/
18 dB gain amplifier. A simple LC noise filter at the output of
the gain amplifier is required to resonate at the desired IF. This
resonant filter port precedes a wide input bandwidth (450 MHz)
track-and-hold followed by an 11-bit analog-to-digital converter
(ADC). A high speed synchronous peak detector monitors sig­nal strength at both input channels. The peak detector drives
RSSI circuitry that automatically adjusts attenuation and gain
on a clock-by-clock basis. The three RSSI indicator bits and the
eleven ADC bits are available at the output providing an exponent
and mantissa data format. Together these integrated components
form an IF sampling, high dynamic range ADC system.

It is helpful to view this device as a stand-alone ADC using
automatic gain control. The gain control referred to in this data
sheet as “gain-ranging” works to maintain a constant SNR over
as wide a range as possible.

As stated previously, the AD6600 has a floating-point output:
eleven mantissa bits and three exponent bits. As shown in Fig­ure 16, at the lowest input levels SNR increases 1 dB for a 1 dB
increase in input power. In this range, the AD6600 is set for
maximum gain. However, when the input signal level reaches

–15–

the gain-ranging section (approximately –42 dBFS), the SNR is
contained between about 50 dB and 56 dB or between 44 dB and
56 dB including the effects of hysteresis. Although Figure 16
does not indicate so, there are slight differences between the
SNR from one gain range to the next as the gain amp switches
between 12 dB and 18 dB. Once the final RSSI range has been
exceeded (approximately –12 dBFS), SNR again increases 1 dB
per 1 dB input power increase until converter full scale is reached.
Again, this performance is very much like the effects of a typical
analog AGC loop.

REV. 0

ENC

ENCGND

DVCC

Figure 16. SNR for Gain-Ranging ADC

AD6600 SUBCIRCUITS
Input Step Attenuator and Gain Stage

The AD6600 has two identical input attenuators, Channel A
and Channel B. These dual inputs are typically used as diversity
channels but may also process two independent IF signals. For
maximum oversampling the device is used in single channel mode;
in this case only one input channel is required. The attenuator
steps are 0 dB, –12 dB and –24 dB. The attenuator settings are
based on the decisions of the RSSI stage (see Peak Detector/
RSSI section). The outputs of the attenuators connect to an
analog multiplexer that selects either Channel A or B for subse­quent processing (see Input Mode). The selected signal drives
a dual-gain amplifier set to either 12 dB or 18 dB; the selected
gain is also determined by the RSSI stage. Therefore, based on
all possible combinations of attenuation and gain, the input
signal receives –12 dB to +18 dB of voltage gain in 6 dB steps
(Table I). Overall gain-matching is typically within 0.1 dB. With
a bandwidth of 1 GHz, the phase delay through the front-end
ranges from 0.2 degrees to 0.5 degrees, depending on input
frequency. Additionally, the input impedance does not change
with attenuator settings so there is no AM-to-PM distortion.

AD6600

Table I. Attenuator and Gain Settings

Attenuator Gain Amp Total RSSI Word

0 dB +18 dB +18 dB 000
0 dB +12 dB +12 dB 001

–12 dB +18 dB +6 dB 010
–12 dB +12 dB 0 dB 011
–24 dB +18 dB –6 dB 100
–24 dB +12 dB –12 dB 101

High-Speed Peak Detector and RSSI Circuitry

The peak detector along with the attenuator and dual gain
amplifier form the control loop within the AD6600.

The peak detector is designed to follow the analog input one clock
cycle before the conversion is actually made. Therefore, while the
converter section of the AD6600 is converting sample “n,” the
peak detector is already looking at sample “n+1.” While look­ing at the “n+1” sample (the calibration period), the peak detec­tor examines the envelope of the input signal. The more of an
envelope that is tracked, the more accurate the gain setting. At
the very least, the peak detector must be presented either a positive
or negative sinusoidal peak, which represents about one-half of a
sine wave cycle. Since the peak detector works for a complete cycle
prior to conversion, the absolute minimum IF frequency that can
be determined is twice the sample rate per channel. Therefore,
at 15 MSPS, the minimum IF frequency that can be sampled
would be 30 MHz.

Note that the more cycles of the input that are monitored by the
peak detector, the more accurate the gain setting will be. There­fore, the actual minimum IF frequency recommended is higher
than this. The minimum specified frequency is 70 MHz. Since the
RSSI control loop is performed on a sample-by-sample basis,
the AD6600 very accurately follows the signals into and out of a
deep fade.

Hysteresis

The AD6600 employs hysteresis to prevent the gain-ranging from
unnecessarily changing when the signal envelope is near an RSSI
threshold. The hysteresis is digital and will account for exactly
6 dB of shift, depending on whether the signal is increasing or
decreasing. This effect is shown in the dashed lines of the over­all transfer function, Figure 16.

External LC Noise Filter, Resonant Port

The output of the attenuator/gain stage drives the wide bandwidth
track-and-hold (T/H), followed by the ADC encoder. Because the
attenuator/gain stage has a very wide bandwidth (~1 GHz), an
LC filter or “resonant port” is provided to limit the amount of
wideband noise delivered to the ADC. The simple LC filter does
not provide signal selectivity and should typically be 35 MHz to
50 MHz wide. However, because the ADC’s track-and-hold itself
has a wide bandwidth (~450 MHz), this noise-limiting filter is
critical to meeting overall sensitivity. Specific details on select­ing components for the resonant port are provided later in the
text (Understanding the External Analog Filter).

ADC Encoder

After the calibration period is complete (one clock cycle), the
appropriate gain and attenuator settings are determined and set.
Once settled, the internal track-and-hold freezes the input signal
so that the ADC encoder may digitize the signal. During digiti­zation, the peak detector/RSSI circuitry is already looking at the
next sample. When the AD6600 is in dual channel mode, the
process is interleaved: while Channel B is monitored for signal
strength, Channel A is digitized. This allows the RSSI to update
on a clock-by-clock basis.

ENCODE

IF INPUT

INTERNAL

2 CLOCK

RSSI

CALIBRATION

AMPLIFIER

CONTROL

T/H INPUT

DIGITIZE

OLD DATA

T-AND-H HOLD

RSSI
CAL.

NOISE FILTER

DISCHARGE

4/8 AMP

CLAMPED

T-AND-H TRACK

RSSI SET

NOISE FILTER

SETTLING

NOISE FILTER

SETTLING

ADC DIGITIZE

T-AND-H HOLD

Figure 17. Internal Timing

Figure 17 shows the internal timing of the chip. The encode
applied to the device initiates several actions. The first and most
important is that the track-and-hold is placed in hold, thus
sampling the analog input at that instant. The second action is that
the peak detector of the RSSI circuitry is initialized. During this
period, the analog input envelope is monitored to determine signal
power. The AD6600 is in calibration mode for about one­quarter of the encode period.

While the AD6600 is in calibration, the external noise filter is
discharged and the amplifier driving the filter disabled. Since this
filter is shared between the two input channels in dual channel
mode, this greatly reduces the feedthrough between the channels
that would otherwise exist. One-quarter of an encode period after
the calibration is complete, the amplifier is re-enabled and allowed
to settle to its new signal conditions for sampling by the wideband
T/H on the next encode signal. The final action is that the signal
on the resonant port is sampled by the track-and-hold. This
happens on the next rising edge of the encode.

Input Mode Select

The AD6600 has two operating modes: single channel and dual
channel. In single channel mode, the ADC always samples Chan­nel A or always samples Channel B. In dual channel mode, the
ADC converter is sampling Channel A and Channel B on alter­nating Encode cycles. Two control pins are provided to select
the desired mode of operation. A_SEL and B_SEL arbitrate the
selection of how these input channels are connected to the out­put. Table II shows the truth table for selection of the input.

–16–

REV. 0

AD6600

Table II. Selecting AD6600 Operating Mode

Output vs. Encode Clock

Mode A_SEL B_SEL n n+1 n+2 n+3

Dual: A/B 1 1 A B A B
Single: A 1 0 A A A A
Single: B 0 1 B B B B
Not Valid 00–– – –

RSSI 11-Bit Word Format Shift Right of

101 DATA DATA× 32 5
100 DATA DATA× 16 4
011 DATA DATA× 83
010 DATA DATA× 42
001 DATA DATA× 21

A_SEL and B_SEL are not logic inputs and should be tied

000 DATA DATA× 10

directly to ground or analog VCC (5 V analog).

In dual channel mode, the AB_OUT signal indicates which
input is currently available on the digital output. When the
AB_OUT is 1, the digital output is the digitized version of
Channel A. Likewise, when AB_OUT is 0, the Channel B is
available on the digital output (Table III).

Table III. AB_OUT for Dual Channel Operation

Output Data vs. Encode Clock

A_SEL and B_SEL = 1 n n+1 n+2 n+3

D[10:0], RSSI[2:0] A B A B
AB_OUT 1010

Data Output Stage

The output stage provides data in the form of mantissa, D[10:0],
and exponent, RSSI[2:0], where D[10:0] represents the output
of the 11-bit ADC coded as two’s complement, and RSSI[2:0]
represents the gain-range setting coded in offset binary. Table
IV shows the nominal gain-ranges for a nominal 2 V p-p differ­ential full-scale input. Keep in mind that the actual full-scale
input voltage and power will vary with input frequency.

When mated with the AD6620, Digital Receive Processor Chip,
the AD6600 floating point data (mantissa + exponent) is automati­cally converted to 16-bit two’s complement format by the AD6620.

APPLYING THE AD6600
Encoding the AD6600

The AD6600 encode signal must be a high quality, extremely
low phase noise source to prevent degradation of performance.
Digitizing high frequency signals (IF range 70 MHz–250 MHz)
places a premium on encode clock phase noise. SNR perfor­mance can easily degrade by 3 dB–4 dB with 70 MHz input
signals when using a high-jitter clock source. At higher IFs (up
to 250 MHz), and with high-jitter clock sources, the higher
slew rates of the input signals reduce performance even further.
See AN-501, Aperture Uncertainty and ADC System Performance
for complete details.

For optimum performance, the AD6600 must be clocked differ­entially. The encode signal is usually ac-coupled into the ENC
and ENC pins via a transformer or capacitors. These pins are
biased internally and require no additional bias.

Figure 18 shows one preferred method for clocking the AD6600.
The sine source (low jitter) is converted from single-ended to

Table IV. Interpreting the RSSI Bits

Differential RSSI [2:0]
Analog Input Voltage Decimal Attenuation
(V p-p) Binary Equiv. or Gain (dB)

0.5 < V

IN

0.25 < V

0.125 < V

< 0.5 100 4 –6

IN

IN

0.0625 < V

0.03125 < V

< 0.25 011 3 0

< 0.125 010 2 +6

IN

< 0.0625 001 1 +12

IN

101 5 –12

differential using an RF transformer. The back-to-back Schottky
diodes across the transformer secondary limit clock excursions
into the AD6600 to approximately 0.8 V p-p differential. This
helps prevent the larger voltage swings of the clock from feeding
through to other portions of the AD6600, and limits the noise
presented to the encode inputs. A crystal clock oscillator can
also be used to drive the RF transformer if an appropriate
limiting resistor (typically 100 Ω) is placed in the series with
the primary.

VIN < 0.03125 000 0 +18

The digital processing chip which follows the AD6600 can com­bine the 11 bits of two’s complement data with the 3 RSSI bits
to form a 16-bit equivalent output word. Table V explains how
the RSSI data can be interpreted when using a PLD or ASIC.
Basically, the circuit performs right shifts of the data depending
on the RSSI word. This can also be performed in software using
the following pseudo code fragment:

r0 = dm (rssi);
r2 = 5;
r0 = r2–r0;
r1 = dm (adc); (11 bits, MSB justified into DSP word)
rshift r1, r0; (arithmetic shift to extend the sign bit)

The result of the shifted data is a 16-bit fixed-point word that
can be used as any normal 16-bit word.

Table V. 16-Bit, Fixed-Point Data Format

16-Bit Data Corresponds to a

SINE

SOURCE

5082–2810

DIODES

ENCODE

AD6600

ENCODE

T1–1T

100

Figure 18. Transformer-Coupled Sine Source

REV. 0

–17–

AD6600

If a low jitter ECL/PECL clock is available, another option is to
ac-couple a differential ECL/PECL signal to the encode input
pins as shown in Figure 19.

VT

ECL/

PECL

0.1F

0.1F

VT

ENCODE

AD6600

ENCODE

Figure 19. AC-Coupled ECL/PECL Encode

Driving the Analog Inputs

As with most new high-speed, high dynamic range analog-to-digital
converters, the analog input to the AD6600 is differential. Differ­ential inputs allow much improvement in performance on-chip
as IF signals are processed through attenuation and gain stages.
Most of the improvement is a result of differential analog stages
having high rejection of even-order harmonics. There are also
benefits at the PCB level. First, differential inputs have high
common-mode rejection to stray signals such as ground and
power noise. They also provide good rejection to common-mode
signals such as local oscillator feedthrough.

Driving a differential analog input introduces some new chal­lenges. Most RF/IF amplifiers are single-ended and may not
obviously interface to the AD6600. However, using simple
techniques, a clean interface is possible. The recommended
method to drive the analog input port is shown in Figure 20.
The AD6600 input is actually designed to match easily to a
SAW filter such as SAWTEK 855297. This allows the SAW
filter to be used in a differential mode, which often improves the
operations of a SAW filter. Using network analyzer data for
both the SAW filter output and the AD6600 input ports (see
data tables for AD6600 S

data), a conjugate match can be

11

used for maximum power transfer. Often an adequate match
can be achieved simply by using a shunt inductor to make the
port look real (Figure 20). For more details on how to exactly
match networks, see RF Circuit Design by Chris Bowick, ISBN:
0-672-21868-2.

FROM

MIXER

OUTPUT

SAW #1

AD6630

SAW #2

AD6600

ADC

Figure 20. Cascaded SAW Filters with AD6630

Where gain is required, the AD6630 differential, low noise, IF
gain block is recommended. This amplifier provides 24 dB of
gain and provides limiting to prevent damage to the SAW filter
and AD6600. The AD6630 is designed to reside between two
SAW filters. This low noise device is ideally suited to many
applications of the AD6600. For more information on the
AD6630, reference the AD6630 data sheet.

When general purpose gain blocks are used, matching can easily
be achieved using a transformer. Most gain blocks are available
with 50 Ω input and output ports. Thus matching to the 200 Ω
impedance of the AD6600 requires only a 1:4 (impedance ratio)
transformer as shown in Figure 21.

FROM

MIXER

OUTPUT

50 GAIN
BLOCK

AD6600

ADC

Figure 21. Transformer-Coupled Gain Block

In the rare case that better matching is required, a conjugate
match between the amplifier selected and the transformer­coupled analog input can be achieved by placing the matching
network between the amplifier and the transformer (Figure 22).
For more details on matching, see the reference mentioned
previously for more details.

FROM

MIXER

OUTPUT

50 GAIN
BLOCK

MATCHING
NETWORK

AD6600

ADC

Figure 22. Gain Block and Matching Network

Understanding the External Analog Filter

Two primary trade-offs must be made when designing the exter­nal resonant filter. The obvious one is the bandwidth of the
filter. The second, not so obvious, trade-off is settling time of
the filter nodes.

Resonant Filter Bandwidth determines the amount of noise that
is limited at the center frequency chosen. If the resonant filter is
too wide, little noise improvement is seen. If the resonant filter
is too narrow, amplitude variation can be seen due to the toler­ance of filter components. If the narrow filter is off center due to
these tolerances (or drift), the 4×/8× signal will fall on the transi­tion band of the filter. An optimum starting point for this filter
is approximately 50 MHz.

Resonant Filter Settling limits the amount of capacitance of this
filter. The output of the 4×/8× amplifier is clamped when the
ADC is processing its input (encode high time). This prevents the
amp output from feeding through to the ADC (T/H) and cor­rupting the ADC results. But, upon the falling edge of encode,
the amp must now come out of clamp and present an accurate
signal to the ADC T/H. The RC of the external filter deter­mines the settling of the amp. If the amp output does not settle,
the ADC sees an attenuated signal. So obviously, a narrow
bandwidth is desired to improve noise performance; but if the
filter is too narrow, the amp will not settle and the ADC will see
an attenuated signal.

Figure 23 shows a simplified model of the 4×/8× amplifier. A
key point to note is that the resistor values in the collector legs
are 315 Ω nominal with a tolerance of ±20%. The filter perfor-
mance is determined by these values in conjunction with the
internal parasitic capacitance, board parasitics and the external
filter components.

–18–

REV. 0

AD6600

AVCC

ENCODE

RESONANT

FILTER PORT

CLAMP

FLT

FLT

FROM

GAIN STAGE

315 315

GND

Figure 23. 4 ×/8× Amplifier Clamp Circuitry

Figure 24 shows why settling is important for this circuit. If the
4×/8× amp does not settle (come out of clamp), the amplitude
presented to the ADC will be decreased. This results in decreased
gain when the filter capacitance is too high.

ENCODE

RESONANT

FILTER

HOLD TRACK HOLD

CLAMPED

SETTLING

Figure 24. 4 ×/8× Amplifier Settling

This explains why the total capacitance allowed for the external
filter varies depending on the clock rate (actually encode clock
high time). If the encode is 13 MSPS and the duty cycle is 50%,
the allowable settling time is 38.5 ns (1/2 of the encode time).
Our assumption is that the amp should be allowed to settle to
1/4 LSB in this time period. This has been proven with both
simulation and empirical analysis. If the settling is assumed to
be an RC circuit, then:

T = RC; t = time; n = number of bits

tT

/

1

()

ntT

/

21

tT

e

1

/

tT





1





n





2

/

()

/

t

n

2

()

ns

38 5

=

315 8192

.

ln

×

Ω

()

=

13 6

.

pF

C

TOTAL

=

In this case, C

VAe

=−

O

−=−

AA A e

1

−=−

1

n

2

1

=

e

n

2

t

=

ln

T

=

T

ln

T

()

ENCODE

Rln

TOTAL

×

05

.

8192

×

()

includes all parasitics and external capaci-

tance. R is nominally 315 Ω. The 8192 is (4 × 2048), which is
1/4 LSB of the converter (11 bits, 2048).

So for settling purposes, with 13 MSPS encode and 50% duty
cycle, the maximum allowable capacitance for proper settling is
C

= 13.6 pF.

TOTAL

As stated above, this C

includes the external capacitors,

TOTAL

the board parasitics, and the AD6600 parasitics. The parasitics
of the AD6600 (lead, internal bond pad and internal connec­tions) at FLT and FLT are 1.75 pF ±0.35 pF (differential).

If the resistors are at maximum value (315 + 20%), the maxi­mum allowable capacitance is C

= 11.3 pF. If the duty

TOTAL

cycle is less than 50%, the maximum allowable capacitance is
further decreased to allow for settling.

Power Supplies

Care should be taken when selecting a power source. Linear
supplies are strongly recommended. Switching supplies tend to
have radiated components that may be “received” by the AD6600.
Each of the power supply pins should be decoupled as closely to
the package as possible using 0.1 µF chip capacitors.

The AD6600 has separate digital and analog power supply pins.
The analog supplies are denoted AVCC and the digital supply
pins are denoted DVCC. Although analog and digital supplies
may be tied together, best performance is achieved when the
supplies are separate. This is because the fast digital output
swings can couple switching current back into the analog sup­plies. Note that AVCC must be held within 5% of 5 Volts; how­ever, the DVCC supply may be varied according to output
digital logic family. The AD6600 is specified for DVCC = 3.3 V
as this is a common supply for digital ASICS.

Output Loading

Care must be taken when designing the data receivers for the
AD6600. Note from the equivalent circuits shown earlier (see
Equivalent Circuits) that D[10:0] and RSSI[2:0] contain a
500 Ω output series resistor. To minimize capacitive loading,
there should only be one gate on each output pin. Extra capaci­tive loading will increase output timing and invalidate timing
specifications. CLK2× and AB_OUT do not contain the output
series resistors. Testing for digital output timing is performed
with 10 pF loads.

Layout Information

The schematic of the evaluation board (Figure 25) represents a
typical implementation of the AD6600. A multilayer board is
recommended to achieve best results. It is highly recommended
that high quality, ceramic chip capacitors be used to decouple
each supply pin to ground directly at the device. The pinout of
the AD6600 facilitates ease of use in the implementation of high
frequency, high resolution design practices. All of the digital
outputs are segregated to two sides of the chip, with the inputs
on the opposite side for isolation purposes.

Care should be taken when routing the digital output traces. To
prevent coupling through the digital outputs into the analog
portion of the AD6600, minimal capacitive loading should be
placed on these outputs. It is recommended that a fanout of
only one be used for all AD6600 digital outputs.

The layout of the analog inputs and the external resonant filter
are critical. No digital traces must be routed near, under, or
above these portions of the circuit. The transformers used for
coupling into the analog inputs must be located as close as
possible to the analog inputs of the AD6600. The external reso­nant filter components must be physically close to the filter­input pins, yet separated from the analog inputs.

REV. 0

–19–

AD6600

The layout of the Encode circuit is equally critical. Any noise
received on this circuitry will result in corruption in the digitiza­tion process and lower overall performance. The Encode clock
must be isolated from the digital outputs and the analog inputs.

Evaluation Board

The evaluation board for the AD6600 is straightforward, con­taining all required circuitry for evaluating the device. The only
external connections required are power supplies, clock and the
analog inputs. The evaluation board includes the option for an
on-board, clock oscillator for encode.

Power to the analog supply pins of the AD6600 is connected via
the power terminal block (TB1). Power for the digital interface
is supplied via Pin 1 of J201, or the VDD e-hole located adja­cent to J201. The VDD supply can vary between 3.3 V to 5.0 V
and sets the level for the output digital data (J201). The J201
connector mates directly with the AD6620 (Receive Signal
Processor) evaluation board, Part # AD6620S/PCB, allowing
complete evaluation of system performance.

The two analog inputs are connected via SMA connectors
AIN and BIN, which are transformer-coupled to the AD6600
inputs. The transformers have a turns-ratio of 1:4 to match
the input resistance of the AD6600 (200 Ω) to 50 Ω at the
SMA connectors.

Table VI. AD6600ST/PCB Bill of Material

The Encode signal may be generated using an on-board crystal
oscillator, U100. If an on-board crystal is used, R104 must be
removed from the board to prevent loading of the oscillator’s
output. The on-board oscillator may be replaced by an external
encode source via the SMA connector labeled ENCODE. If an
external source is used, it must be a high quality and very low
phase noise source. The high IF range of the AD6600 (70 MHz
–250 MHz) demands that the Encode clock be sufficiently pure
to maintain performance.

The AD6600 output data is latched using 74LCX574 (U201,
U202) latches. The clock for these latches is determined by
jumper selection on header J1. The clock can be a delayed ver­sion of the encode clock (CLKA, CLKB), or the CLK2× gener­ated by the AD6600. A clock is also distributed with the output
data (J201) that is labeled CLKX (Pin 11, J201). The CLK× is
selected with jumpers on header J1 and can be CLKA, CLKB,
or CLK2×.

The resonant LC filter components (SEL2, C2 and C3) are
omitted. The user must install proper values based on the IF
chosen. See Understanding the External Analog Filter section of
the data sheet for guidelines on selecting these components.

Item Quantity Reference Description

1 3 AIN, BIN, ENCODE SMA Connector
2 14 C1, C102–108, C114, C117–118, Ceramic Chip Capacitor 1206, 0.1 µF

C120–121, C299

3 2 C100–101 Tantalum Chip Capacitor, 10 µF
4 1 C111 Ceramic Chip Capacitor 0805, 0.1 µF
54C112–C113, C115–116 Ceramic Chip Capacitor 0508, 0.1 µF
6 2 CR1–2 1N2810 Schottky Diode
7 1 DUT AD6600AST
8 1 J1 20-Pin Double Row Male Header
9 1 J201 50-Pin Double Row Male Header, Right Angle
10 2 R1–2 Omitted
11 2 R100–R101 Surface Mount Resistor 1206, 10 kΩ
12 1 R103 Surface Mount Resistor 1206, 100 Ω
13 1 R104 Surface Mount Resistor 1206, 50 Ω
14 2 R298–R299 Surface Mount Resistor 1206, 2 kΩ
15 3 T1–T2, T4 Surface Mount Transformer Mini-Circuits T4–1T
16 1 TB1 PCTB2 Terminal Block
17 2 U201–U202 74LCX574 Octal Latch
18 1 U204 74LVQ00 Two Input NAND Gate

–20–

REV. 0

J201

H50DM

DD

V

501

GND

249

GND

GND

348

BIT9

BIT10

GND

447

546

BIT8

GND

BIT7

GND

GND

GND

GND

GND

GND

GND

GND

GND

GND

GND

645

744

843

942

10411140123913381437153616351734183319322031213022292328242725

CLK

BIT6

BIT5

BIT4

BIT3

BIT2

BIT1

BIT0

GND

CLKX

GND

GND

GND

GND

GND

GND

GND

GND

GND

GND

GND

GND

GND

RSSIB2

RSSIB1

A/B

GND

26

REMOVE 21 AND 30

CLK

GND

RSSIB0

AD6600

R101

10k

B

A

R100

10k

VCC

VCC

CLKA

6

11

GND

GND

201918171615141312

J1

123456789

H20DM

A

B

CR2

CR1

C1

0.1F

R1 R2

2

3

T4

TI–4T

4

CLKREF

U100

VCC

14

CLKX

CLKA

CLKB

11

10

CLKA

CLKB

CLK_2X

CLK

D10D9D8D7D6D5D4D3D2D1D0

1N2810

1N2810

C113

0.01F

1

1:4

6

C114

0.1F

R103

100

8

CC

OUT

V

VEE

K1115
7

SMA

ENCODE

R104

A/B

BIT0

BIT1

BIT2

BIT3

BIT4

BIT5

BIT6

12131415161718

8D 8Q

7D 7Q

74LCX574

9876543

D0D1D2D3D4D5D6

RSSI2

A/B A/B

40 39 3841

42

4344 36 35 3437

6D 6Q

D0

D1

D2

D3

D4

D5

D6

D7

D8

D9

D10

C111

U201

RSSI0

RSSI1

50

5D 5Q

(LSB)

(MSB)

0.1F

4D 4Q

3D 3Q

VDD

33

DVCC

DVCC
1

VDD

C112

2D 2Q

GND

32

GND

GND
2

GND

0.01F

19

2

1D 1Q

VCC

31

AVCC

C1

3

OE

CK

GND

30

AVCC
4

VCC

11 1

29

GND

AB_OUT

AD6600AST

GND
567

GND

BIT7

BIT8

12131415161718

8D 8Q

7D 7Q

U202

74LCX574

9876543

D7D8D9

2CLK

CLK_2

VCC

27

2825262324

ENC

ENC

RSSI1

RSSI0
8

AVCC

B_SEL
9

B

CLK2X

RSSI2

BIT9

BIT10

6D 6Q

D10

GND

GND

A_SEL

10

A

RSSIB2

5D 5Q

4D 4Q

RSSI2

BIN

AIN

11

T1

RSSIB1

3D 3Q

RSSI1

RSSI0

3

TI–4T

4

RSSIB0

19

2D 2Q

1D 1Q

2

GND

T2

T1–4T

BIN

GND

AVCC

GND

AVCC

FLT

FLT

AVCC

AVCC

GND

AIN

2

AIN

SMA

CK

12 13 14 15 16 17 18 19 20 21 22

OE

3

U204

5

2

CLKREF

TB1

VCC

U204

2k

PCTB2

74LVQ00

13

12

GND

8

74LVQ00

9

10

GND

VCC

C100 C102 C104 C106 C108 C120 C121

+

1

2

VDD

C101 C103 C105 C107 C117 C118

+

GND

GND

U204

74LVQ00

4

11 1

C116

1

2

4

6

BIN

SMA

GND

VCC

GND

VCC

VCC

VCC

GND

C115

0.01F

1

1:4

6

0.01F

DD

V

U204

1:4

SEL2

3

74LVQ00

1

R2982R299

C299

0.1F

C3

SEL

C2

SEL

REV. 0

Figure 25. AD6600ST/PCB Schematic Diagram

–21–

AD6600

Figure 26. AD6600ST/PCB Top Side Silk Screen

Figure 27. AD6600ST/PCB Top Side Copper

Figure 28. AD6600ST/PCB Bottom Side Copper

Figure 29. AD6600ST/PCB Power Supply Layer (Negative)

Figure 30. AD6600ST/PCB Ground Layer (Negative)

–22–

REV. 0

AD6600

Connecting the AD6600 with the AD6620

The AD6600 interfaces directly to the AD6620 Digital Receive
Signal Processor as shown in Figure 31. No additional external
components are required. Note that the layout requirements dis­cussed previously do apply and deviations can result in degraded
performance. The digital outputs of the AD6600 must connect
directly to the AD6620 inputs with no additional fanout. Addi­tional loading on the outputs will compromise timing performance.

ENC

ENC

(MSB) D10

AD6600

D9
D8
D7
D6
D5
D4
D3
D2
D1

(LSB) D0

RSSI2
RSSI1
RSSI0

AB_OUT

CLK2

IN15
IN14
IN13
IN12
IN11
IN10
IN9
IN8
IN7
IN6
IN5
IN4
IN3
IN2
IN1
IN0
EXP2
EXP1
EXP0
A/B
CLK

AD6620

Figure 31. AD6600/AD6620 Connections

Figure 32 shows the timing details between the AD6600 and the
AD6620. On Clock 1, D[10:0], RSSI[2:0], and AB_OUT are
captured by the AD6620. Since AB_OUT has changed state from
the previous clock, the D[10:0] and RSSI[2:0] are processed by
the AD6620. This clock allows adequate setup and hold time
for AB_OUT, D[10:0], and RSSI[2:0] to be captured by the
AD6620.

On Clock2, D[10:0], RSSI[2:0], and AB_OUT are captured
by the AD6620. Since AB_OUT has not changed from the
previous clock, the D[10:0] and RSSI[2:0] are ignored by the
AD6620. This clock is concerned only with the AB_OUT setup­and-hold time.

CLK2

D [10:0]

RSSI [2:0]

AB_OUT

38.5 38.5

CLOCK1 CLOCK2

3.0 16.5

7.0

Figure 32. AD6600 to AD6620 Timing at 13 MSPS

3.0 16.5

12.5

REV. 0

–23–

AD6600

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm).

44-Terminal LQFP (Low-Profile Quad Plastic Flatpack)

(ST-44)

0.063 (1.60)

0.030 (0.75)

0.018 (0.45)

SEATING

PLANE

MAX

33

34

0.472 (12.00) SQ

TOP VIEW

(PINS DOWN)

23

22

0.394
(10.0)

SQ

C00966–2.5–7/00 (rev. 0)

0.006 (0.15)

0.002 (0.05)

0.057 (1.45)

0.053 (1.35)

44

1

0.031 (0.80)
BSC

12

11

0.018 (0.45)

0.012 (0.30)

–24–

PRINTED IN U.S.A.

REV. 0