TEXAS INSTRUMENTS THS3001, THS3002 Technical data

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THS3001, THS3002

420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

High Speed – 420 MHz Bandwidth (G = 1, –3 dB) – 6500 V/µs Slew Rate – 40-ns Settling Time (0.1%)

High Output Drive, IO = 100 mA

Excellent Video Performance – 115 MHz Bandwidth (0.1 dB, G = 2) – 0.01% Differential Gain – 0.02° Differential Phase

Low 3-mV (max) Input Offset Voltage

Very Low Distortion – THD = –96 dBc at f = 1 MHz – THD = –80 dBc at f = 10 MHz

Wide Range of Power Supplies – V

Evaluation Module Available

= ±4.5 V to ±16 V

description

The THS300x is a high-speed current-feedback operational amplifier, ideal for communication, imaging, and high-quality video applications. This device offers a very fast 6500-V/µs slew rate, a 420-MHz bandwidth, and 40-ns settling time for large-signal applications requiring excellent transient response. In addition, the THS300x operates with a very low distortion of – 96 dBc, making it well suited for applications such as wireless communication basestations or ultrafast ADC or DAC buffers.

THS3001

D AND DGN† PACKAGE

(TOP VIEW)

NULL

NC – No internal connection †

IN–

IN+

3 4

CC–

The THS3001 implemented in the DGN package is in the product preview stage of development. Contact your local TI sales office for availability.

NULL

CC+

OUT

6 5

1OUT

–V

THS3002

D AND DGN PACKAGE

(TOP VIEW)

1IN– 1IN+

1 2 3 4

8 7 6 5

OUTPUT AMPLITUDE

FREQUENCY

Output Amplitude – dB

G = 2

RL = 150 Ω VI = 200 mV RMS

–1

1M 100M

10M 1G100k

f – Frequency – Hz

VCC = ±15 V RF = 680 Ω

VCC = ±5 V RF = 750 Ω

CC+

2OUT 2IN– 2IN+

HIGH-SPEED AMPLIFIER FAMILY

DEVICE

THS3001/02 • • • 420 6500 –96 40 0.01% 0.02° 1.6 THS4001 • • • • 270 400 –72 40 0.04% 0.15° 12.5

THS4011/12 • • • 290 310 –80 37 0.006% 0.01° 7.5 THS4031/32 • • • 100 100 –72 60 0.02% 0.03° 1.6 THS4061/62 • • • 180 400 –72 40 0.02% 0.02° 14.5

PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters.

ARCHITECTURE

VFB CFB 5 V ±5 V ±15 V

CAUTION: The THS300x provides ESD protection circuitry. However, permanent damage can still occur if this device is subjected to high-energy electrostatic discharges. Proper ESD precautions are recommended to avoid any performance degradation or loss of functionality.

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

SUPPLY

VOLTAGE

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

THD

f = 1 MHz

(dB) (ns)

0.1%

DIFF.

THS3001, THS3002

†

MODULE

PACKAGE

Suppl

oltage, V

and V

Operating free-air temperature, T

°C

420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

AVAILABLE OPTIONS

PACKAGED DEVICE

0°C to 70°C

–40°C to 85°C

†

The D package is available taped and reeled. Add an R suffix to the device type (i.e., THS3001CDR)

‡

Product Preview

SOIC

(D)

THS3001CD

THS3002CD

THS3001ID

THS3002ID

THS3001CDGN

‡

THS3002CDGN

THS3001IDGN

‡

THS3002IDGN

MSOP (DGN)

DEVICE SYMBOL

‡

TIADP

‡

TIADQ

‡

TIADI

TIADJ

EVALUATION

THS3001EVM

THS3002EVM

—

‡

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)

Supply voltage, V Input voltage, V Output Current, I

to V

CC+

±V

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

175 mA. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Differential input voltage, V

Continuous total power dissipation See Dissipation Rating Table. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Operating free-air temperature, T Storage temperature, T

stg

Lead temperature, 1,6 mm (1/16 inch) from case for 10 seconds 300°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

†

Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

POWER RATING ABOVE TA = 25°C

D 740 mW 6 mW/°C 470 mW 380 mW

33 V. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

CC–

±6 V. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

, THS300xC 0°C to 70°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

THS300xI –40°C to 85°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

–65°C to 125°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

DISSIPATION RATING TABLE

≤ 25°C DERATING FACTOR T

= 70°C T

POWER RATING

= 85°C

POWER RATING

recommended operating conditions

MIN NOM MAX UNIT

y v

CC+

CC–

Split supply ±4.5 ±16 Single supply 9 32 THS300xC 0 70

THS300xI –40 85

†

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

VCCPower supply operating range

ICCQuiescent current

±15 V

±5 V

VOOutput voltage swing

±15 V

IOOutput current (see Note 1)

VIOInput offset voltage

±15 V

In ut

IIBIn ut bias current

±15 V

µA

Input

Common-mode input voltage range

Oen loo transresistance

MΩ

CMRR

Common-mode rejection ratio

±5 V

PSRR

Power supply rejection ratio

±15 V

RIInput resistance

InInput current noise

A/√H

THS3001, THS3002

420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

electrical characteristics, TA = 25°C, RL = 150 Ω, RF = 1 kΩ (unless otherwise noted)

PARAMETER TEST CONDITIONS

Input offset voltage drift VCC = ±5 V or ±15 V 5 µV/°C

–

ICR

†

Full range = 0°C to 70°C for the THS300xC and –40°C to 85°C for the THS300xI.

NOTE 1: Observe power dissipation ratings to keep the junction temperature below absolute maximum when the output is heavily loaded or

Differential input capacitance 7.5 pF Output resistance Open loop at 5 MHz 10 Ω

Input voltage noise

shorted. See absolute maximum ratings section.

+Input 1.5 MΩ –Input 15 Ω

Positive (IN+) Negative (IN–)

Split supply ±4.5 ±16.5 Single supply 9 33

= ±5

VCC = ±5 V, RL = 20 Ω 100 VCC = ±15 V, RL = 75 Ω 85 120

= ±5 V or

VCC = ±5 V ±3 ±3.2 VCC = ±15 V ±12.9 ±13.2 VCC = ±5 V,

RL = 1 kΩ VCC = ±15 V,

RL = 1 kΩ VCC = ±5 V, VCM = ±2.5 V 62 70 VCC = ±15 V, VCM = ±10 V 65 73

VCC = ±5 V or ±15 V, f = 10 kHz, G = 2

= ±5 V or ±15 V, f = 10 kHz,

G = 2

†

TA = 25°C 5.5 7.5 TA = full range 8.5

TA = 25°C 6.6 9 TA = full range 10 RL = 150 Ω ±2.9 ±3.2 RL = 1 kΩ ±3 ±3.3 RL = 150 Ω ±12.1 ±12.8 RL = 1 kΩ ±12.8 ±13.1

TA = 25°C 1 3 TA = full range 4

TA = 25°C 2 10 TA = full range 15 TA = 25°C 1 10 TA = full range 15

VO = ±2.5 V ,

VO = ±7.5 V ,

TA = 25°C 65 76 TA = full range 63 TA = 25°C 69 76 TA = full range 67

MIN TYP MAX UNIT

1.3

2.4

1.6 nV/√Hz 13

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

THS3001, THS3002

Slew rate (see Note 2)

V/µs

ADDifferential gain error

θDDifferential phase error

Bandwidth for 0.1 dB flatness

O(PP)

Full ower bandwidth (see Note 3)

O(PP)

420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

operating characteristics, TA = 25°C, RL = 150 Ω, RF = 1 kΩ (unless otherwise noted)

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

= ±5 V,

= 4 V

O(PP)

= ±15 V,

= 20 V

O(PP)

Settling time to 0.1%

THD Total harmonic distortion

Small signal bandwidth (–3 dB)

Crosstalk (THS3002 only) TBD dB

NOTES: 2. Slew rate is measured from an output level range of 25% to 75%.

3. Full power bandwidth is defined as the frequency at which the output has 3% THD.

VCC = ±15 V, 0 V to 10 V Step

VCC = ±5 V, 0 V to 2 V Step,

VCC = ±15 V, fc = 10 MHz,

G = 2, 40 IRE modulation, ±100 IRE Ramp, NTSC and PAL

= 1,

G = 2, RF = 750 Ω, VCC = ±5 V 300 G = 2, RF = 680 Ω, VCC = ±15 V 385 G = 5, RF = 560 Ω, VCC = ±15 V 350 G = 2, RF = 750 Ω, VCC = ±5 V 85 G = 2, RF = 680 Ω, VCC = ±15 V 115 VCC = ±5 V,

RL = 500 Ω VCC = ±15 V,

RL = 500 Ω

= 1 kΩ,

= 4 V,

= 20

G = –5 1700 G = 5 1300 G = –5 6500 G = 5 6300 Gain = –1,

Gain = –1,

= 2 V,

O(PP)

G = 2 VCC = ±5 V 0.015%

VCC = ±15 V 0.01%

VCC = ±5 V 0.01°

VCC = ±15 V 0.02° VCC = ±5 V, 330 MHz

VCC = ±15 V, 420 MHz

G = –5 65 MHz G = 5 62 MHz G = –5 32 MHz G = 5 31 MHz

–80 dBc

MHz

PARAMETER MEASUREMENT INFORMATION

50 Ω

Figure 1. Test Circuit, Gain = 1 + (RF/RG)

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

VCC+

– +

VCC–

PSRR

Power supply rejection ratio

Slew rate

Harmonic distortion

THS3001, THS3002

420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

TYPICAL CHARACTERISTICS

Table of Graphs

FIGURE

|VO| Output voltage swing vs Free-air temperature 2 I

CMRR Common-mode rejection ratio

Current supply vs Free-air temperature 3 Input bias current vs Free-air temperature 4 Input offset voltage vs Free-air temperature 5

vs Common-mode input voltage 6 vs Common-mode input voltage 7

vs Frequency 8 Transresistance vs Free-air temperature 9 Closed-loop output impedance vs Frequency 10 Voltage noise vs Frequency 11 Current noise vs Frequency 11

Normalized slew rate vs Gain 17

Differential gain vs Loading 22, 23 Differential phase vs Loading 24, 25 Output amplitude vs Frequency 26–30 Normalized output response vs Frequency 31–34 Small and large signal frequency response 35, 36 Small signal pulse response 37, 38 Large signal pulse response 39 – 46

vs Frequency 12

vs Free-air temperature 13

vs Supply voltage 14

vs Output step peak-to-peak 15, 16

vs Peak-to-peak output voltage swing 18, 19

vs Frequency 20, 21

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

TYPICAL CHARACTERISTICS

OUTPUT VOLTAGE SWING

FREE-AIR TEMPERATURE

3.5 3

2.5 2

VCC = ±15 V No Load

VCC = ±5 V No Load

–20 20

0 40 100–40

TA – Free-Air Temperature – ° C

VCC = ±15 V RL = 150 Ω

VCC = ±5 V RL = 150 Ω

60 80

13.5

12.5

– Output Voltage Swing – VV

– Supply Current – mA

–20 20

Figure 2

INPUT BIAS CURRENT

FREE-AIR TEMPERATURE

–0.5

CURRENT SUPPLY

FREE-AIR TEMPERATURE

VCC = ±15 V

VCC = ±10 V

VCC = ±5 V

0 40 100–40

TA – Free-Air Temperature – ° C

60 80

Figure 3

INPUT OFFSET VOLTAGE

FREE-AIR TEMPERATURE

–1

Aµ

–1.5

–2

– Input Bias Current –

–2.5

–3

–40 –20 0 20 80 100

TA – Free-Air Temperature – ° C

VCC = ±5 V

VCC = ±15 V

VCC = ±5 V

VCC = ±15 V

6040

Figure 4

IB+

IB–

–0.2

–0.4

–0.6

–0.8

– Input Offset Voltage – mV

–1

–1.2

Gain = 1 RF = 1 kΩ

–20 20

0 40 100–40

TA – Free-Air Temperature – ° C

Figure 5

VCC = ±5 V

VCC = ±15 V

60 80

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

THS3001, THS3002

420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

TYPICAL CHARACTERISTICS

COMMON-MODE REJECTION RATIO

COMMON-MODE INPUT VOLTAGE

TA = –40°C

TA = 85°C

TA = 25°C

CMRR – Common-Mode Rejection Ratio – dB

VCC = ±15 V

2648 14

|VIC| – Common-Mode Input Voltage – V

Figure 6

COMMON-MODE REJECTION RATIO

FREQUENCY

VCC = ±15 V

VCC = ±5 V

CMRR – Common-Mode Rejection Ratio – dB

1k 10k 10M 100M1M100k

1 kΩ

– +

1 kΩ

f – Frequency – Hz

Figure 8

10 12

COMMON-MODE REJECTION RATIO

COMMON-MODE INPUT VOLTAGE

TA = –40°C

TA = 85°C

CMRR – Common-Mode Rejection Ratio – dB

VCC = ±5 V

0.5 1.512 4

|VIC| – Common-Mode Input Voltage – V

TA = 25°C

2.5 3

Figure 7

TRANSRESISTANCE

FREE-AIR TEMPERATURE

2.8

2.6

2.4

2.2

1.8

1.6

Transresistance – MΩ

1.4

1.2

VO = VCC/2 RL = 1 kΩ

–40

–20 20

0 40 100

TA – Free-Air Temperature – ° C

VCC = ±15 V

VCC = ±10 V

VCC = ±5 V

Figure 9

3.5

60 80

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

TYPICAL CHARACTERISTICS

CLOSED-LOOP OUTPUT IMPEDANCE

FREQUENCY

100

VCC = ±15 V RF = 750 Ω

Ω

Closed-Loop Output Impedance –

Gain = +2 TA = 25°C

= 2 V

I(PP)

750 Ω

0.1

0.01 100k 10M 100M

f – Frequency – Hz

Figure 10

POWER SUPPLY REJECTION RATIO

FREQUENCY

VCC = ±5 V

50 Ω

750 Ω

– +

THS300x

(

1 kΩ

1000

– 1

VOLTAGE NOISE AND CURRENT NOISE

FREQUENCY

1000

and

nV/ Hz– Voltage Noise –V

– Current Noise – pA/

)

VCC = ±15 V and ±5 V TA = 25°C

100

100 10k1k 100k10

f – Frequency – Hz

n–

Figure 11

POWER SUPPLY REJECTION RATIO

FREE-AIR TEMPERATURE

VCC = ±15 V

PSRR – Power Supply Rejection Ratio – dB

1k 10k 10M 100M1M100k

VCC = ±5 V

G = 1 RF = 1 kΩ

f – Frequency – Hz

VCC = ±15 V

+PSRR

Figure 12

–PSRR

PSRR – Power Supply Rejection Ratio – dB

–20 20

0 40 100–40

TA – Free-Air Temperature – ° C

VCC = –15 V

VCC = +15 V

Figure 13

VCC = –5 V

VCC = +5 V

60 80

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

THS3001, THS3002

420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

TYPICAL CHARACTERISTICS

SR – Slew Rate – V/µs

SLEW RATE

SUPPLY VOLTAGE

7000

G = +5 RL = 150 Ω

6000

5000

4000

3000

2000

1000

tr/tf = 300 ps RF = 1 kΩ

+SR

–SR

5 15

711913

|VCC| – Supply Voltage – V

Figure 14

SLEW RATE

OUTPUT STEP

2000

+SR

1000

–SR

10000

SR – Slew Rate – V/µs

1000

100

1.5

1.4

1.3

1.2

1.1

SLEW RATE

OUTPUT STEP

+SR

–SR

VCC = ±15 V G = +5 RL = 150 Ω tr/tf = 300 ps RF = 1 kΩ

515

O(PP)

10 200

– Output Step – V

Figure 15

NORMALIZED SLEW RATE

GAIN

VCC = ±5 V V

= 4 V

O(PP)

RL = 150 Ω RF = 1 kΩ tr/tf = 300 ps

–Gain

SR – Slew Rate – V/µs

100

240

– Output Step – V

O(PP)

= ±5 V

G = +5 RL = 150 Ω tr/tf = 300 ps RF= 1 kΩ

0.9

SR – Normalized Slew Rate – V/µs

0.8

0.7 24

Figure 16

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

+Gain

35 101

Figure 17

G – Gain – V/V

THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

TYPICAL CHARACTERISTICS

HARMONIC DISTORTION

PEAK-TO-PEAK OUTPUT VOLTAGE SWING

–50

8 MHz Gain = 2

–55

VCC = ±15 V RL = 150 Ω RF = 750 Ω

–60

–65

–70

–75

Harmonic Distortion – dBc

–80

–85

0 2 4 6 12 14108 16

– Peak-to-Peak Output Voltage Swing – V

O(PP)

2nd Harmonic

Figure 18

HARMONIC DISTORTION

FREQUENCY

–70

Gain = 2 VCC = ±15 V

–75

–80

–85

VO = 2 V RL = 150 Ω RF = 750 Ω

3rd Harmonic

HARMONIC DISTORTION

PEAK-TO-PEAK OUTPUT VOLTAGE SWING

–50

4 MHz Gain = 2

–55

VCC = ±15 V RL = 150 Ω

–60

RF = 750 Ω

–65

–70

–75

–80

Harmonic Distortion – dBc

–85

–90 –95

0 2 4 6 12 14108 16

– Peak-to-Peak Output Voltage Swing – V

O(PP)

2nd Harmonic

Figure 19

HARMONIC DISTORTION

FREQUENCY

–60

Gain = 2 VCC = ±5 V

–65

–70

–75

–80

VO = 2 V RL = 150 Ω RF = 750 Ω

3rd Harmonic

–90

Harmonic Distortion – dBc

–95

–100

100k 1M 10M

2nd Harmonic

f – Frequency – Hz

Figure 20

–85

–90

Harmonic Distortion – dBc

–95

–100

100k 1M 10M

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2nd Harmonic

3rd Harmonic

f – Frequency – Hz

Figure 21

THS3001, THS3002

420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

TYPICAL CHARACTERISTICS

DIFFERENTIAL GAIN

0.04 Gain = 2 RF = 750 Ω 40 IRE NTSC Modulation Worst Case: ±100 IRE Ramp

0.03

VCC = ±15 V

0.02

VCC = ±5 V

Differential Gain – %

0.01

1234 78

Number of 150 Ω Loads

Figure 22

DIFFERENTIAL PHASE

0.3 Gain = 2 RF = 750 Ω 40 IRE NTSC Modulation

0.25

Worst Case: ±100 IRE Ramp

0.2

DIFFERENTIAL GAIN

0.04 Gain = 2 RF = 750 Ω 40 IRE PAL Modulation Worst Case: ±100 IRE Ramp

0.03

VCC = ±15 V

0.02

VCC = ±5 V

Differential Gain – %

0.01

1234 78

Number of 150 Ω Loads

Figure 23

DIFFERENTIAL PHASE

0.35 Gain = 2 RF = 750 Ω

0.3

40 IRE PAL Modulation Worst Case: ±100 IRE Ramp

0.25

0.15 VCC = ±15 V

0.1

Differential Phase – Degrees

0.05

1234 78

Number of 150 Ω Loads

VCC = ±5 V

Figure 24

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

0.2

0.15

0.1

Differential Phase – Degrees

0.05

1234 78

VCC = ±15 V

VCC = ±5 V

Number of 150 Ω Loads

Figure 25

THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

TYPICAL CHARACTERISTICS

Gain = 1 VCC = ±15 V

RL = 150 Ω VI = 200 mV RMS

–1

–2

–3

Output Amplitude – dB

–4

–5

–6

Gain = 2 VCC = ±15 V

RL = 150 Ω

VI = 200 mV RMS

OUTPUT AMPLITUDE

FREQUENCY

RF = 750 Ω

RF = 1 kΩ

RF = 1.5 kΩ

1M 100M

10M 1G100k

f – Frequency – Hz

Figure 26

OUTPUT AMPLITUDE

FREQUENCY

RF = 560 Ω

Gain = 1 VCC = ±5 V

RL = 150 Ω VI = 200 mV RMS

–1

–2

–3

Output Amplitude – dB

–4

–5

–6

Gain = 2 VCC = ±5 V

RL = 150 Ω

VI = 200 mV RMS

OUTPUT AMPLITUDE

FREQUENCY

RF = 750 Ω

RF = 1 kΩ

RF = 1.5 kΩ

1M 100M

10M 1G100k

f – Frequency – Hz

Figure 27

OUTPUT AMPLITUDE

FREQUENCY

RF = 560 Ω

6 5

Output Amplitude – dB

–1

RF = 680 Ω

RF = 1 kΩ

1M 100M

10M 1G100k

f – Frequency – Hz

Figure 28

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

6 5

Output Amplitude – dB

–1

RF = 750 Ω

RF = 1 kΩ

1M 100M

10M 1G100k

f – Frequency – Hz

Figure 29

THS3001, THS3002

420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

TYPICAL CHARACTERISTICS

OUTPUT AMPLITUDE

FREQUENCY

Output Amplitude – dB

–10

NORMALIZED OUTPUT RESPONSE

FREQUENCY

Gain = –1 VCC = ±15 V

RL = 150 Ω VI = 200 mV RMS

RF = 560 Ω

VCC = ±5 V

G = +1000 RF = 10 kΩ RL = 150 Ω VO = 200 mV RMS

1M 100M

10M 1G100k

f – Frequency – Hz

Figure 30

VCC = ±15 V

NORMALIZED OUTPUT RESPONSE

FREQUENCY

Gain = –1 VCC = ±5 V RL = 150 Ω VI = 200 mV RMS

RF = 560 Ω

–1

–2

–3

–4

Normalized Output Response – dB

–5

–6

1M 100M

f – Frequency – Hz

RF = 680 Ω

10M 1G100k

Figure 31

RF = 1 kΩ

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

–1

–2

–3

–4

Normalized Output Response – dB

–5

–6

1M 100M

f – Frequency – Hz

RF = 750 Ω

10M 1G100k

Figure 32

RF = 1 kΩ

THS3001, THS3002 420-MHz HIGH-SPEED CURRENT-FEEDBACK AMPLIFIERS

SLOS217A – JULY 1998 – REVISED JUNE 1999

TYPICAL CHARACTERISTICS

NORMALIZED OUTPUT RESPONSE

FREQUENCY

RF = 390 Ω

–3

–6

–9

Normalized Output Response – dB

Gain = +5

–12

VCC = ±15 V RL = 150 Ω VO = 200 mV RMS

–15

1M 100M

f – Frequency – Hz

RF = 560 Ω

10M 1G100k

Figure 33

SMALL AND LARGE SIGNAL

FREQUENCY RESPONSE

–3

VI = 500 mV

–6

RF = 1 kΩ

NORMALIZED OUTPUT RESPONSE

FREQUENCY

–2

–4

–6

–8

–10

Normalized Output Response – dB

Gain = +5 VCC = ±5 V RL = 150 Ω

–12

VO = 200 mV RMS

–14

1M 100M

f – Frequency – Hz

RF = 620 Ω

10M 1G100k

Figure 34

SMALL AND LARGE SIGNAL

FREQUENCY RESPONSE

VI = 500 mV

RF = 390 Ω

RF = 1 kΩ

–9

VI = 250 mV

–12

–15

VI = 125 mV

–18

–21

Output Level – dBV

–24

–27

–30

VI = 62.5 mV

Gain = 1 VCC = ±15 V RF = 1 kΩ RL = 150 Ω

1M 100M

10M 1G100k

f – Frequency – Hz

Figure 35

–3

VI = 250 mV

–6

–9

VI = 125 mV

–12

–15

Output Level – dBV

–18

–21

–24

VI = 62.5 mV

Gain = 2 VCC = ±15 V RF = 680 Ω RL = 150 Ω

1M 100M

10M 1G100k

f – Frequency – Hz

Figure 36

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TYPICAL CHARACTERISTICS

SMALL SIGNAL PULSE RESPONSE

300

100

–100

– Input Voltage – mV

–200

200

100

–100

– Output Voltage – V

–200

–300

0302010 40 50 7060 80 90 100

Gain = 1 VCC = ±5 V RL = 150 Ω RF = 1 kΩ tr/tf = 300 ps

t – Time – ns

Figure 37

LARGE SIGNAL PULSE RESPONSE

SMALL SIGNAL PULSE RESPONSE

–20

– Input Voltage – mV

–60

200 100

–100

–200

– Output Voltage – mV

–300

0302010 40 50 7060 80 90 100

Gain = 5 VCC = ±5 V RL = 150 Ω RF = 1 kΩ tr/tf = 300 ps

t – Time – ns

Figure 38

LARGE SIGNAL PULSE RESPONSE

–1

– Input Voltage – V

–3

–1

– Output Voltage – V

–2

–3

0302010 40 50 7060 80 90 100

Gain = +1 VCC = ±15 V RL = 150 Ω RF = 1 kΩ tr/tf= 2.5 ns

t – Time – ns

Figure 39

–1

– Input Voltage – V

–3

–1

– Output Voltage – V

–2

–3

0302010 40 50 7060 80 90 100

Gain = 1 VCC = ±5 V RL = 150 Ω RF = 1 kΩ tr/tf= 2.5 ns

t – Time – ns

Figure 40

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TYPICAL CHARACTERISTICS

LARGE SIGNAL PULSE RESPONSE

–1

– Input Voltage – V

–3

5 0

–5

– Output Voltage – V

–10

–15

0302010 40 50 7060 80 90 100

Gain = +5 VCC = ±15 V RL = 150 Ω RF = 1 kΩ tr/tf= 300 ps

t – Time – ns

Figure 41

LARGE SIGNAL PULSE RESPONSE

600

200

–200

– Input Voltage – mV

–600

–1

– Output Voltage – V

–2

–3

0302010 40 50 7060 80 90 100

Gain = 5 VCC = ±5 V RL = 150 Ω RF = 1 kΩ tr/tf= 300 ps

t – Time – ns

Figure 42

LARGE SIGNAL PULSE RESPONSE

–1

– Input Voltage – V

–1

–2

– Output Voltage – V

–3

0302010 40 50 7060 80 90 100

Gain = –1 VCC = ±15 V RL = 150 Ω RF = 1 kΩ tr/tf= 2.5 ns

t – Time – ns

Figure 43

–1

– Input Voltage – V

–1

–2

– Output Voltage – V

–3

0302010 40 50 7060 80 90 100

Gain = –1 VCC = ±5 V RL = 150 Ω RF = 1 kΩ tr/tf= 300 ps

t – Time – ns

Figure 44

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TYPICAL CHARACTERISTICS

LARGE SIGNAL PULSE RESPONSE

600

200

–200

– Input Voltage – mV

–600

1 0

–1

– Output Voltage – V

–2

–3

0302010 40 50 7060 80 90 100

Gain = –5 VCC = ±5 V RL = 150 Ω RF = 1 kΩ tr/tf= 300 ps

t – Time – ns

Figure 45

LARGE SIGNAL PULSE RESPONSE

–1

– Input Voltage – V

–2

5 0

–5

– Output Voltage – V

–10

–15

0302010 40 50 7060 80 90 100

Gain = –5 VCC = ±15 V RL = 150 Ω RF = 1 kΩ tr/tf= 300 ps

t – Time – ns

Figure 46

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APPLICATION INFORMATION

theory of operation

The THS300x is a high-speed, operational amplifier configured in a voltage-feedback architecture. The device is built using a 30-V, dielectrically isolated, complementary bipolar process with NPN and PNP transistors possessing f has a wide bandwidth, high slew rate, fast settling time, and low distortion. A simplified schematic is shown in Figure 47.

s of several GHz. This configuration implements an exceptionally high-performance amplifier that

CC+

IN+ IN–

Figure 47. Simplified Schematic

CC–

OUT

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recommended feedback and gain resistor values

The THS300x is fabricated using Texas Instruments 30-V complementary bipolar process, HVBiCOM. This process provides the excellent isolation and extremely high slew rates that result in superior distortion characteristics.

As with all current-feedback amplifiers, the bandwidth of the THS300x is an inversely proportional function of the value of the feedback resistor (see Figures 26 to 34). The recommended resistors for the optimum frequency response are shown in Table 1. These should be used as a starting point and once optimum values are found, 1% tolerance resistors should be used to maintain frequency response characteristics. For most applications, a feedback resistor value of 1 kΩ is recommended – a good compromise between bandwidth and phase margin that yields a very stable amplifier.

Consistent with current-feedback amplifiers, increasing the gain is best accomplished by changing the gain resistor, not the feedback resistor . This is because the bandwidth of the amplifier is dominated by the feedback resistor value and internal dominant-pole capacitor. The ability to control the amplifier gain independent of the bandwidth constitutes a major advantage of current-feedback amplifiers over conventional voltage-feedback amplifiers. Therefore, once a frequency response is found suitable to a particular application, adjust the value of the gain resistor to increase or decrease the overall amplifier gain.

Finally, it is important to realize the effects of the feedback resistance on distortion. Increasing the resistance decreases the loop gain and increases the distortion. It is also important to know that decreasing load impedance increases total harmonic distortion (THD). Typically, the third-order harmonic distortion increases more than the second-order harmonic distortion.

Table 1. Recommended Resistor Values for Optimum Frequency Response

GAIN RF for VCC = ±15 V RF for VCC = ±5 V

1 1 kΩ 1 kΩ

2, –1 680 Ω 750 Ω

–2 620 Ω 620 Ω

5 560 Ω 620 Ω

offset voltage

The output offset voltage, (VOO) is the sum of the input offset voltage (VIO) and both input bias currents (IIB) times the corresponding gains. The following schematic and formula can be used to calculate the output offset voltage:

IB–

– +

VOO+

IB+

Figure 48. Output Offset Voltage Model

) ǒ

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)

) ǒ

IB–RF

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noise calculations and noise figure

Noise can cause errors on very small signals. This is especially true for amplifying small signals coming over a transmission line or an antenna. The noise model for current-feedback amplifiers (CFB) is the same as for voltage feedback amplifiers (VFB). The only difference between the two is that CFB amplifiers generally specify different current-noise parameters for each input, while VFB amplifiers usually only specify one noise-current parameter. The noise model is shown in Figure 49. This model includes all of the noise sources as follows:

• e

= amplifier internal voltage noise (nV/√Hz)

• IN+ = noninverting current noise (pA/√Hz)

• IN– = inverting current noise (pA/√Hz)

• e

The total equivalent input noise density (eni) is calculated by using the following equation:

Where:

= thermal voltage noise associated with each resistor (eRx = 4 kTRx)

eni+

)ǒIN

)

k = Boltzmann’s constant = 1.380658 × 10 T = temperature in degrees Kelvin (273 +°C) R

|| RG = parallel resistance of RF and R

IN+

IN–

Figure 49. Noise Model

)

IN–

Noiseless

+ _

ǒRFø

–23

)

4kTRs)

4kT

RFø

To get the equivalent output noise of the amplifier, just multiply the equivalent input noise density (eni) by the overall amplifier gain (A

eno+

eniAV+

)

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APPLICATION INFORMATION

noise calculations and noise figure (continued)

As the previous equations show, to keep noise at a minimum, small value resistors should be used. As the closed-loop gain is increased (by reducing R resistance term. This leads to the general conclusion that the most dominant noise sources are the source resistor (R method, noise sources smaller than 25% of the largest noise source can be effectively ignored. This can greatly simplify the formula and make noise calculations much easier.

This brings up another noise measurement usually preferred in RF applications, the noise figure (NF). Noise figure is a measure of noise degradation caused by the amplifier. The value of the source resistance must be defined and is typically 50 Ω in RF applications.

Because the dominant noise components are generally the source resistance and the internal amplifier noise voltage, we can approximate noise figure as:

) and the internal amplifier noise voltage (en). Because noise is summed in a root-mean-squares

10log

ȱ ȧ Ȳ

), the input noise is reduced considerably because of the parallel

)

4kTR

)

10log

ȱ ȧ

ȧ ȧ ȧ ȧ

ȡ ȧ Ȣ

)

The Figure 50 shows the noise figure graph for the THS300x.

NOISE FIGURE

SOURCE RESISTANCE

f = 10 kHz

TA = 25°C

16 14 12

Noise Figure – dB

ȧ ȧ ȧ ȧ

2 0

10 100 10k

RS – Source Resistance – Ω

Figure 50. Noise Figure vs Source Resistance

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slew rate

The slew rate performance of a current-feedback amplifier, like the THS300x, is affected by many different factors. Some of these factors are external to the device, such as amplifier configuration and PCB parasitics, and others are internal to the device, such as available currents and node capacitance. Understanding some of these factors should help the PCB designer arrive at a more optimum circuit with fewer problems.

Whether the THS300x is used in an inverting amplifier configuration or a noninverting configuration can impact the output slew rate. As can be seen from the specification tables as well as some of the figures in this data sheet, slew-rate performance in the inverting configuration is faster than in the noninverting configuration. This is because in the inverting configuration the input terminals of the amplifier are at a virtual ground and do not significantly change voltage as the input changes. Consequently , the time to charge any capacitance on these input nodes is less than for the noninverting configuration, where the input nodes actually do change in voltage an amount equal to the size of the input step. In addition, any PCB parasitic capacitance on the input nodes degrades the slew rate further simply because there is more capacitance to charge. Also, if the supply voltage (V

) to the amplifier is reduced, slew rate decreases because there is less current available within the amplifier

to charge the capacitance on the input nodes as well as other internal nodes. Internally , the THS300x has other factors that impact the slew rate. The amplifier’s behavior during the slew-rate

transition varies slightly depending upon the rise time of the input. This is because of the way the input stage handles faster and faster input edges. Slew rates (as measured at the amplifier output) of less than about 1500 V/µs are processed by the input stage in a very linear fashion. Consequently, the output waveform smoothly transitions between initial and final voltage levels. This is shown in Figure 51. For slew rates greater than 1500 V/µs, additional slew-enhancing transistors present in the input stage begin to turn on to support these faster signals. The result is an amplifier with extremely fast slew-rate capabilities. Figures 41 and 52 show waveforms for these faster slew rates. The additional aberrations present in the output waveform with these faster-slewing input signals are due to the brief saturation of the internal current mirrors. This phenomenon, which typically lasts less than 20 ns, is considered normal operation and is not detrimental to the device in any way . If for any reason this type of response is not desired, then increasing the feedback resistor or slowing down the input-signal slew rate reduces the effect.

SLEW RATE

– Input Voltage – V

SR = 1500 V/µs

–5

–10

– Output Voltage – V

–15

0604020 80 100 140120 160 180 200

Gain = 5 VCC = ±15 V RL = 150 Ω RF = 1 kΩ tr/tf = 10 ns

t – Time – ns

Figure 51

SLEW RATE

– Input Voltage – V

–2

–5

–10

– Output Voltage – V

–15

0604020 80 100 140120 160 180 200

SR = 2400 V/µs Gain = 5 VCC = ±15 V RL = 150 Ω RF = 1 kΩ tr/tf = 5 ns

t – Time – ns

Figure 52

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APPLICATION INFORMATION

driving a capacitive load

Driving capacitive loads with high-performance amplifiers is not a problem as long as certain precautions are taken. The first is to realize that the THS300x has been internally compensated to maximize its bandwidth and slew-rate performance. When the amplifier is compensated in this manner, capacitive loading directly on the output will decrease the device’s phase margin leading to high-frequency ringing or oscillations. Therefore, for capacitive loads of greater than 10 pF, it is recommended that a resistor be placed in series with the output of the amplifier, as shown in Figure 53. A minimum value of 20 Ω should work well for most applications. For example, in 75-Ω transmission systems, setting the series resistor value to 75 Ω both isolates any capacitance loading and provides the proper line impedance matching at the source end.

1 kΩ

Input

THS300x

20 Ω

LOAD

Output

Figure 53. Driving a Capacitive Load

PCB design considerations

Proper PCB design techniques in two areas are important to assure proper operation of the THS300x. These areas are high-speed layout techniques and thermal-management techniques. Because the THS300x is a high-speed part, the following guidelines are recommended.

Ground plane – It is essential that a ground plane be used on the board to provide all components with a low inductive ground connection. Although a ground connection directly to a terminal of the THS300x is not necessarily required, it is recommended that the thermal pad of the package be tied to ground. This serves two functions: it provides a low inductive ground to the device substrate to minimize internal crosstalk, and it provides the path for heat removal.

Input stray capacitance – To minimize potential problems with amplifier oscillation, the capacitance at the inverting input of the amplifiers must be kept to a minimum. T o do this, PCB trace runs to the inverting input must be as short as possible, the ground plane must be removed under any etch runs connected to the inverting input, and external components should be placed as close as possible to the inverting input. This is especially true in the noninverting configuration. An example of this can be seen in Figure 54, which shows what happens when a 1-pF capacitor is added to the inverting input terminal. The bandwidth increases at the expense of peaking. This is because some of the error current is flowing through the stray capacitor instead of the inverting node of the amplifier. Although, while the device is in the inverting mode, stray capacitance at the inverting input has a minimal effect. This is because the inverting node is at a

ground

seen in Figure 55, where a 10-pF capacitor adds only 0.35 dB of peaking. In general, as the gain of the system increases, the output peaking due to this capacitor decreases. While this can initially look like a faster and better system, overshoot and ringing are more likely to occur under fast transient conditions. So proper analysis of adding a capacitor to the inverting input node should be performed for stable operation.

and the voltage does not fluctuate nearly as much as in the noninverting configuration. This can be

virtual

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APPLICATION INFORMATION

PCB design considerations (continued)

OUTPUT AMPLITUDE

FREQUENCY

7 6

4 3 2 1 0

Output Amplitude – dB

–1 –2

Gain = 1 VCC = ±15 V

–3

VO = 200 mV RMS

–4

100k 10M 100M

1 kΩ

– +

50 Ω

out

RL =

150 Ω

f – Frequency – Hz

CI = 1 pF

CI = 0 pF

(Stray C Only)

–1

–2

–3

–4

–5

Output Amplitude – dB

–6

Gain = –1 VCC = ±15 V

–7

VO = 200 mV RMS

–8

100k 10M 100M

Figure 54

Proper power-supply decoupling – Use a minimum 6.8-µF tantalum capacitor in parallel with a 0.1-µF

OUTPUT AMPLITUDE

FREQUENCY

CI = 10 pF

CI = Stray C Only

1 kΩ

50 Ω

1 kΩ

– +

f – Frequency – Hz

RL =

150 Ω

Figure 55

out

ceramic capacitor on each supply terminal. It may be possible to share the tantalum among several amplifiers depending on the application, but a 0.1-µF ceramic capacitor should always be used on the supply terminal of every amplifier. In addition, the 0.1- µF capacitor should be placed as close as possible to the supply terminal. As this distance increases, the inductance in the connecting etch makes the capacitor less effective. The designer should strive for distances of less than 0.1 inches between the device power terminal and the ceramic capacitors.

thermal information

The THS300x incorporates output-current-limiting protection. Should the output become shorted to ground, the output current is automatically limited to the value given in the data sheet. While this protects the output against excessive current, the device internal power dissipation increases due to the high current and large voltage drop across the output transistors. Continuous output shorts are not recommended and could damage the device. Additionally, connection of the amplifier output to one of the supply rails (±V of the device is possible under this condition and should be avoided. But, the THS300x does not incorporate thermal-shutdown protection. Because of this, special attention must be paid to the device’s power dissipation or failure may result.

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APPLICATION INFORMATION

thermal information (continued)

The thermal coefficient θJA is approximately 169°C/W for the SOIC 8-pin D package. For a given θJA, the maximum power dissipation, shown in Figure 56, is calculated by the following formula:

MAX–TA

Where:

PD+

= Maximum power dissipation of THS300x (watts)

= Absolute maximum junction temperature (150°C)

MAX

= Free-ambient air temperature (°C)

= Thermal coefficient from die junction to ambient air (°C/W)

1.5

MAXIMUM POWER DISSIPATION

FREE-AIR TEMPERATURE

SOIC-D Package:

θJA = 169°C/W

TJ = 150°C No Airflow

0.5

– Maximum Power Dissipation – W

–20 20

Figure 56. Maximum Power Dissipation vs Free-Air Temperature

0 40 100–40

TA – Free-Air Temperature – ° C

60 80

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APPLICATION INFORMATION

general configurations

A common error for the first-time CFB user is the creation of a unity gain buffer amplifier by shorting the output directly to the inverting input. A CFB amplifier in this configuration will oscillate and is not recommended. The THS300x, like all CFB amplifiers, must have a feedback resistor for stable operation. Additionally, placing capacitors directly from the output to the inverting input is not recommended. This is because, at high frequencies, a capacitor has a very low impedance. This results in an unstable amplifier and should not be considered when using a current-feedback amplifier. Because of this, integrators and simple low-pass filters, which are easily implemented on a VFB amplifier, have to be designed slightly dif ferently . If filtering is required, simply place an RC-filter at the noninverting terminal of the operational-amplifier (see Figure 57).

– +

–3dB

+ ǒ

)

2pR1C1

1)sR1C1

Figure 57. Single-Pole Low-Pass Filter

If a multiple-pole filter is required, the use of a Sallen-Key filter can work very well with CFB amplifiers. This is because the filtering elements are not in the negative feedback loop and stability is not compromised. Because of their high slew-rates and high bandwidths, CFB amplifiers can create very accurate signals and help minimize distortion. An example is shown in Figure 58.

R2R1

+ _

R1 = R2 = R C1 = C2 = C Q = Peaking Factor (Butterworth Q = 0.707)

2pRC R

2 –

(

)

–3dB

Figure 58. 2-Pole Low-Pass Sallen-Key Filter

There are two simple ways to create an integrator with a CFB amplifier. The first, shown in Figure 59, adds a resistor in series with the capacitor. This is acceptable because at high frequencies, the resistor is dominant and the feedback impedance never drops below the resistor value. The second, shown in Figure 60, uses positive feedback to create the integration. Caution is advised because oscillations can occur due to the positive feedback.

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general configurations (continued)

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APPLICATION INFORMATION

– +

THS300x

Figure 59. Inverting CFB Integrator

– +

R2R1

THS300x

Figure 60. Noninverting CFB Integrator

+ ǒ

For Stable Operation:

R1 || R

≅ V

F G

(

ȧ Ȣ

1 +

sR1C1

≥

RFC1

ȣ ȧ Ȥ

)

The THS300x may also be employed as a very good video distribution amplifier. One characteristic of distribution amplifiers is the fact that the differential phase (DP) and the differential gain (DG) are compromised as the number of lines increases and the closed-loop gain increases (see Figures 22 to 25 for more information). Be sure to use termination resistors throughout the distribution system to minimize reflections and capacitive loading.

750 Ω750 Ω

–

+ THS300x

75 Ω

N Lines

75 Ω

75-Ω Transmission Line

75 Ω

Figure 61. Video Distribution Amplifier Application

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evaluation board

Evaluation boards are available for the THS3001 (literature #SLOP130) and the THS3002 (literature #SLOP241). The boards have been configured for very low parasitic capacitance in order to realize the full performance of the amplifier. Schematics of the evaluation boards are shown in Figures 62 and 63. The circuitry has been designed so that the amplifier may be used in either an inverting or noninverting configuration. T o order the evaluation board contact your local TI sales office or distributor . For more detailed information, refer to the

THS3001 EVM User’s Manual

To order the evaluation board, contact your local TI sales office or distributor.

(literature #SLOV021) or the

VCC+

0.1 µF

THS3002 EVM User’s Guide

6.8 µF

(literature #SLOVxxx).

IN+

IN–

1 kΩ

49.9 Ω

1 kΩ

49.9 Ω

0.1 µF

THS3001

VCC–

6.8 µF

Figure 62. THS3001 Evaluation Board Schematic

49.9 Ω

OUT

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evaluation board (continued)

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APPLICATION INFORMATION

100 Ω

R13

R2 0 Ω

–

VCC+

VCC–

0.1 µF

301 Ω

THS3002 U1:A

0.1 µF

R14

301 Ω

49.9 Ω

OUT1

100 Ω

R12

100 Ω

R11

100 Ω

R10 0 Ω

–

THS3002 U1:B

R15

49.9 Ω

Figure 63. THS3002 Evaluation Board Schematic

OUT2

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MECHANICAL INFORMATION

D (R-PDSO-G**) PLASTIC SMALL-OUTLINE PACKAGE

14 PIN SHOWN

0.069 (1,75) MAX

0.050 (1,27)

0.020 (0,51)

0.014 (0,35)

0.010 (0,25)

0.004 (0,10)

0.010 (0,25)

0.157 (4,00)

0.150 (3,81)

0.244 (6,20)

0.228 (5,80)

Seating Plane

0.004 (0,10)

PINS **

DIM

A MAX

A MIN

0.008 (0,20) NOM

Gage Plane

0°–8°

0.197

(5,00)

0.189

(4,80)

0.344 (8,75)

0.337

(8,55)

0.010 (0,25)

0.044 (1,12)

0.016 (0,40)

4040047/D 10/96

0.394

(10,00)

0.386

(9,80)

NOTES: A. All linear dimensions are in inches (millimeters).

B. This drawing is subject to change without notice. C. Body dimensions do not include mold flash or protrusion, not to exceed 0.006 (0,15). D. Falls within JEDEC MS-012

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MECHANICAL INFORMATION

DGN (S-PDSO-G8) PowerPAD PLASTIC SMALL-OUTLINE PACKAGE

0,65

1,07 MAX

3,05 2,95

0,38 0,25

3,05 2,95

Seating Plane

0,15 0,05

0,25

4,98 4,78

0,10

Thermal Pad (See Note D)

0,15 NOM

Gage Plane

0°–6°

0,25

0,69 0,41

NOTES: A. All linear dimensions are in millimeters.

B. This drawing is subject to change without notice. C. Body dimensions include mold flash or protrusions. D. The package thermal performance may be enhanced by attaching an external heat sink to the thermal pad. This pad is electrically

and thermally connected to the backside of the die and possibly selected leads.

E. Falls within JEDEC MO-187

PowerPAD is a trademark of Texas Instruments Incorporated.

4073271/A 01/98

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TEXAS INSTRUMENTS THS3001, THS3002 Technical data

Specifications and Main Features

Frequently Asked Questions

User Manual