22 High Frequency Electronics
High Frequency Design
RF POWER AMPLIFIERS
RF and Microwave Power
Amplifier and Transmitter
Technologies —
Part 2
By Frederick H. Raab, Peter Asbeck, Steve Cripps, Peter B. Kenington,
Zoya B. Popovich, Nick Pothecary, John F. Sevic and Nathan O. Sokal
P
art 1 of this series
introduced basic
concepts, discussed
the characteristics of signals to be amplified, and
gave background information on RF power
devices. Part 2 reviews
the basic techniques, ratings, and implementation
methods for power amplifiers operating at HF
through microwave frequencies.
6a. BASIC TECHNIQUES FOR
RF POWER AMPLIFICATION
RF power amplifiers are commonly designated as classes A, B, C, D, E, and F [19]. All
but class A employ various nonlinear, switching, and wave-shaping techniques. Classes of
operation differ not in only the method of
operation and efficiency, but also in their
power-output capability. The power-output
capability (“transistor utilization factor”) is
defined as output power per transistor normalized for peak drain voltage and current of
1 V and 1 A, respectively. The basic topologies
(Figures 7, 8 and 9) are single-ended, transformer-coupled, and complementary. The
drain voltage and current waveforms of selected ideal PAs are shown in Figure 10.
Class A
In class A, the quiescent current is large
enough that the transistor remains at all
times in the active region and acts as a current source, controlled by the drive.
Consequently, the drain voltage and current
waveforms are (ideally) both sinusoidal. The
power output of an ideal class-A PA is
P
o
= V
om
2
/ 2R (5)
where output voltage V
om
on load R cannot
exceed supply voltage V
DD
. The DC-power
input is constant and the efficiency of an ideal
PA is 50 percent at PEP. Consequently, the
instantaneous efficiency is proportional to the
power output and the average efficiency is
inversely proportional to the peak-to-average
ratio (e.g., 5 percent for x = 10 dB). The utilization factor is 1/8.
For amplification of amplitude-modulated
signals, the quiescent current can be varied in
proportion to the instantaneous signal envelope. While the efficiency at PEP is
unchanged, the efficiency for lower ampli-
Our multi-part series on
power amplifier tech-
nologies and applications
continues with a review of
amplifier configurations,
classes of operation,
device characterization
and example applications
This series of articles is an expanded version of the paper, “Power Amplifiers and Transmitters for RF and
Microwave” by the same authors, which appeared in the the 50th anniversary issue of the IEEE Transactions on
Microwave Theory and Techniques, March 2002. © 2002 IEEE. Reprinted with permission.
Figure 7 · A single-ended power amplifier.
From May 2003 High Frequency Electronics
Copyright © 2003 Summit Technical Media, LLC
24 High Frequency Electronics
High Frequency Design
RF POWER AMPLIFIERS
tudes is considerably improved. In an
FET PA, the implementation
requires little more than variation of
the gate-bias voltage.
The amplification process in class
A is inherently linear, hence increasing the quiescent current or decreasing the signal level monotonically
decreases IMD and harmonic levels.
Since both positive and negative
excursions of the drive affect the
drain current, it has the highest gain
of any PA. The absence of harmonics
in the amplification process allows
class A to be used at frequencies close
to the maximum capability (fmax) of
the transistor. However, the efficiency
is low. Class-A PAs are therefore typically used in applications requiring
low power, high linearity, high gain,
broadband operation, or high-frequency operation.
The efficiency of real class-A PAs
is degraded by the on-state resistance
or saturation voltage of the transistor. It is also degraded by the presence of load reactance, which in
essence requires the PA to generate
more output voltage or current to
deliver the same power to the load.
Class B
The gate bias in a class-B PA is
set at the threshold of conduction so
that (ideally) the quiescent drain current is zero. As a result, the transistor is active half of the time and the
drain current is a half sinusoid.
Since the amplitude of the drain current is proportional to drive amplitude and the shape of the drain-current waveform is fixed, class-B provides linear amplification.
The power output of a class-B PA
is controlled by the drive level and
varies as given by eq. (5). The DCinput current is, however, proportional to the drain current which is in
turn proportional to the RF-output
current. Consequently, the instantaneous efficiency of a class-B PA
varies with the output voltage and
for an ideal PA reaches π/4 (78.5 percent) at PEP. For low-level signals,
class B is significantly more efficient
than class A, and its average efficiency can be several times that of class A
at high peak-to-average ratios (e.g.,
28 vs. 5 percent for ξ = 10 dB). The
utilization factor is the same 0.125 of
class A.
In practice, the quiescent current
is on the order of 10 percent of the
peak drain current and adjusted to
minimize crossover distortion caused
by transistor nonlinearities at low
outputs. Class B is generally used in
a push-pull configuration so that the
two drain-currents add together to
produce a sine-wave output. At HF
and VHF, the transformer-coupled
push-pull topology (Figure 8) is generally used to allow broadband operation with minimum filtering. The
use of the complementary topology
(Figure 9) has generally been limited
to audio, LF, and MF applications by
the lack of suitable p-channel transistors. However, this topology is
attractive for IC implementation and
has recently been investigated for
low-power applications at frequencies to 1 GHz [20].
Class C
In the classical (true) class-C PA,
the gate is biased below threshold so
that the transistor is active for less
than half of the RF cycle (Figure 10).
Linearity is lost, but efficiency is
increased. The efficiency can be
increased arbitrarily toward 100 percent by decreasing the conduction
angle toward zero. Unfortunately,
this causes the output power (utilization factor) to decrease toward zero
and the drive power to increase
toward infinity. A typical compromise
is a conduction angle of 150° and an
ideal efficiency of 85 percent.
The output filter of a true class-C
PA is a parallel-tuned type that
Figure 8 · Transformer-coupled
push-pull PA.
Figure 9 · Complementary PA. Figure 10 · Wavefrorms for ideal PAs.
26 High Frequency Electronics
High Frequency Design
RF POWER AMPLIFIERS
bypasses the harmonic components
of the drain current to ground without generating harmonic voltages.
When driven into saturation, efficiency is stabilized and the output
voltage locked to supply voltage,
allowing linear high-level amplitude
modulation.
Classical class C is widely used in
high-power vacuum-tube transmitters. It is, however, little used in
solid-state PAs because it requires
low drain resistances, making implementation of parallel-tuned output
filters difficult. With BJTs, it is also
difficult to set up bias and drive to
produce a true class-C collector-current waveform. The use of a seriestuned output filter results in a
mixed-mode class-C operation that is
more like mistuned class E than true
class C.
Class D
Class-D PAs use two or more transistors as switches to generate a
square drain-voltage waveform. A
series-tuned output filter passes only
the fundamental-frequency component to the load, resulting in power
outputs of (8/π
2
)V
DD
2
/R and
(2/π
2
)V
DD
2
/R for the transformer-coupled and complementary configurations, respectively. Current is drawn
only through the transistor that is
on, resulting in a 100-percent efficiency for an ideal PA. The utilization
factor (1/2π = 0.159) is the highest of
any PA (27 percent higher than that
of class A or B). A unique aspect of
class D (with infinitely fast switching) is that efficiency is not degraded
by the presence of reactance in the
load.
Practical class-D PAs suffer from
losses due to saturation, switching
speed, and drain capacitance. Finite
switching speed causes the transistors to be in their active regions while
conducting current. Drain capacitances must be charged and discharged once per RF cycle. The associated power loss is proportional to
V
DD
3
/2 [21] and increases directly
with frequency.
Class-D PAs with power outputs
of 100 W to 1 kW are readily implemented at HF, but are seldom used
above lower VHF because of losses
associated with the drain capacitance. Recently, however, experimental class-D PAs have been tested with
frequencies of operation as high as 1
GHz [22].
Class E
Class E employs a single transistor operated as a switch. The drainvoltage waveform is the result of the
sum of the DC and RF currents
charging the drain-shunt capacitance. In optimum class E, the drain
voltage drops to zero and has zero
slope just as the transistor turns on.
The result is an ideal efficiency of 100
percent, elimination of the losses
associated with charging the drain
capacitance in class D, reduction of
switching losses, and good tolerance
of component variation.
Optimum class-E operation
requires a drain shunt susceptance
0.1836/R and a drain series reactance 1.15R and delivers a power output of 0.577V
DD
2
/R for an ideal PA
[23]. The utilization factor is 0.098.
Variations in load impedance and
shunt susceptance cause the PA to
deviate from optimum operation [24,
25], but the degradations in performance are generally no worse than
those for class A and B.
The capability for efficient operation in the presence of significant
drain capacitance makes class E useful in a number of applications. One
example is high-efficiency HF PAs
with power levels to 1 kW based upon
low-cost MOSFETs intended for
switching rather than RF use [26].
Another example is the switchingmode operation at frequencies as
high as K band [27]. The class-DE PA
[28] similarly uses dead-space
between the times when its two transistors are on to allow the load network to charge/discharge the drain
capacitances.
Class F
Class F boosts both efficiency and
output by using harmonic resonators
in the output network to shape the
drain waveforms. The voltage waveform includes one or more odd harmonics and approximates a square
wave, while the current includes even
harmonics and approximates a half
sine wave. Alternately (“inverse class
F”), the voltage can approximate a
half sine wave and the current a
square wave. As the number of harmonics increases, the efficiency of an
ideal PA increases from the 50 percent (class A) toward unity (class D)
and the utilization factor increases
from 1/8 (class A) toward 1/2π (class
D) [29].
The required harmonics can in
principle be produced by currentsource operation of the transistor.
However, in practice the transistor is
driven into saturation during part of
the RF cycle and the harmonics are
produced by a self-regulating mechanism similar to that of saturating
class C. Use of a harmonic voltage
requires creating a high impedance
(3 to 10 times the load impedance) at
the drain, while use of a harmonic
current requires a low impedance
(1/3 to 1/10 of the load impedance).
While class F requires a more complex output filter than other PAs, the
impedances must be correct at only a
few specific frequencies. Lumped-element traps are used at lower frequencies and transmission lines are
used at microwave frequencies.
Typically, a shorting stub is placed a
quarter or half-wavelength away
from the drain. Since the stubs for
different harmonics interact and the
open or short must be created at a
“virtual drain” ahead of the drain
capacitance and bond-wire inductance, implementation of suitable
networks is a bit of an art.
Nonetheless, class-F PAs are successfully implemented from MF through
Ka band.
A variety of modes of operation inbetween class C, E, and F are possi-
28 High Frequency Electronics
High Frequency Design
RF POWER AMPLIFIERS
ble. The maximum achievable efficiency [30] depends upon the number
of harmonics, (0.5, 0.707, 0.8165,
0.8656, 0.9045 for 1 through 5 harmonics, respectively). The utilization
factor depends upon the harmonic
impedances and is highest for ideal
class-F operation.
6b. LOAD-PULL
CHARACTERIZATION
RF-power transistors are characterized by breakdown voltages and
saturated drain currents. The combination of the resultant maximum
drain voltage and maximum drain
current dictates a range of load
impedances into which useful power
can be delivered, as well as an
impedance for delivery of the maximum power. The load impedance for
maximum power results in drain
voltage and current excursions from
near zero to nearly the maximum
rated values.
The load impedances corresponding to delivery of a given amount of
RF power with a specified maximum
drain voltage lie along parallel-resis-
tance lines on the Smith chart. The
impedances for a specified maximum
current analogously follow a seriesresistance line. For an ideal PA, the
resultant constant-power contour is
football-shaped as shown in Figure
11.
In a real PA, the ideal drain is
embedded behind the drain capacitance and bond-wire/package inductance. Transformation of the ideal
drain impedance through these elements causes the constant-power
contours to become rotated and distorted [31]. With the addition of second-order effects, the contours
become elliptical. A set of power contours for a given PA somewhat
resembles a set of contours for a conjugate match. However, a true conjugate match produces circular contours. With a power amplifier, the
process is more correctly viewed as
loading to produce a desired power
output. As shown in the example of
Figure 12, the power and efficiency
contours are not necessarily aligned,
nor do maximum power and maximum efficiency necessarily occur for
the same load impedance. Sets of
such “load-pull” contours are widely
used to facilitate design trade-offs.
Load-pull analyses are generally
iterative in nature, as changing one
parameter may produce a new set of
contours. A variety of different
parameters can be plotted during a
load-pull analysis, including not only
power and efficiency, but also distortion and stability. Harmonic
impedances as well as drive
impedances are also sometimes varied.
A load-pull system consists essentially of a test fixture, provided with
biasing capabilities, and a pair of lowloss, accurately resettable tuners,
usually of precision mechanical construction. A load-pull characterization procedure consists essentially of
measuring the power of a device, to a
given specification (e.g., the 1-dB
compression point) as a function of
impedance. Data are measured at a
large number of impedances and
plotted on a Smith chart. Such plots
are, of course, critically dependent on
the accurate calibration of the tuners,
both in terms of impedance and losses. Such calibration is, in turn, highly
dependent on the repeatability of the
tuners.
Precision mechanical tuners, with
micrometer-style adjusters, were the
traditional apparatus for load-pull
analysis. More recently, a new generation of electronic tuners has
emerged that tune through the use
varactors or transmission lines
switched by pin diodes. Such electronic tuners [32] have the advantage
of almost perfect repeatability and
high tuning speed, but have much
higher losses and require highly complex calibration routines. Mechanical
tuners are more difficult to control
using a computer, and move very
slowly from one impedance setting to
another.
In an active load-pull system, a
second power source, synchronized in
frequency and phase with the device
input excitation, is coupled into the
output of the device. By controlling
the amplitude and phase of the
injected signal, a wide range of
impedances can be simulated at the
output of the test device [33]. Such a
Figure 11 · Contant power contours
and transformation.
Figure 12 · Example load-pull contours for a 0.5-W, 836 MHz PA.
(Courtesy Focus Microwaves and
dBm Engineering)
30 High Frequency Electronics
High Frequency Design
RF POWER AMPLIFIERS
system eliminates the expensive
tuners, but creates a substantial calibration challenge of its own. The wide
availability of turn-key load-pull systems has generally reduced the application of active load-pull to situations
where mechanical or electronic tuning becomes impractical (e.g., millimeter-wave frequencies).
6c. STABILITY
The stability of a small-signal RF
amplifier is ensured by deriving a set
of S-parameters from using measured data or a linear model, and
then establishing the value of the kfactor stability parameter. If the kfactor is greater than unity, at the
frequency and bias level in question,
then expressions for matching
impedances at input and output can
be evaluated to give a perfect conjugate match for the device. Amplifier
design in this context is mainly a
matter of designing matching networks which present the prescribed
impedances over the necessary specified bandwidth. If the k factor is less
than unity, negative feedback or lossy
matching must be employed in order
to maintain an unconditionally stable
design.
A third case is relevant to PA
design at higher microwave frequencies. There are cases where a device
has a very high k-factor value, but
very low gain in conjugate matched
condition. The physical cause of this
can be traced to a device which has
gain roll-off due to carrier-mobility
effects, rather than parasitics. In
such cases, introduction of some positive feedback reduces the k-factor
and increases the gain in conjugately
matched conditions, while maintaining unconditional stability. This technique was much used in the early era
of vacuum-tube electronics, especially
in IF amplifiers.
6d. MICROWAVE IMPLEMENTATION
At microwave frequencies, lumped
elements (capacitors, inductors)
become unsuitable as tuning compo-
nents and are used primarily as
chokes and by-passes. Matching, tuning, and filtering at microwave frequencies are therefore accomplished
with distributed (transmission-line)
networks. Proper operation of power
amplifiers at microwave frequencies
is achieved by providing the required
drain-load impedance at the fundamental and a number of harmonic
frequencies.
Class F
Class-F operation is specified in
terms of harmonic impedances, so it
is relatively easy to see how transmission-line networks are used.
Methods for using transmission lines
in conjunction with lumped-element
tuned circuits appear in the original
paper by Tyler [34]. In modern
microwave implementation, however,
it is generally necessary to use transmission lines exclusively. In addition,
the required impedances must be
produced at a virtual ideal drain that
is separated from the output network
by drain capacitance, bond-wire/lead
inductance.
Typically, a transmission line
between the drain and the load provides the fundamental-frequency
drain impedance of the desired value.
A stub that is a quarter wavelength
at the harmonic of interest and open
at one end provides a short circuit at
the opposite end. The stub is placed
along the main transmission line at
either a quarter or a half wavelength
from the drain to create either an
open or a short circuit at the drain
[35]. The supply voltage is fed to the
drain through a half-wavelength line
bypassed on the power-supply end or
alternately by a lumped-element
choke. When multiple stubs are used,
the stub for the highest controlled
harmonic is placed nearest the drain.
Stubs for lower harmonics are placed
progressively further away and their
lengths and impedances are adjusted
to allow for interactions. Typically,
“open” means three to ten times the
fundamental-frequency impedance,
and “shorted” means no more 1/10 to
1/3 of the fundamental-frequency
impedance [FR17].
A wide variety of class-F PAs have
been implemented at UHF and
microwave frequencies [36-41].
Generally, only one or two harmonic
impedances are controlled. In the Xband PA from [42], for example, the
output circuit provides a match at the
fundamental and a short circuit at
the second harmonic. The third-harmonic impedance is high, but not
explicitly adjusted to be open. The 3dB bandwidth of such an output network is about 20 percent, and the efficiency remains within 10 percent of
its maximum value over a bandwidth
of approximately 10 to 15 percent.
Dielectric resonators can be used
in lieu of lumped-element traps in
class-F PAs. Power outputs of 40 W
have been obtained at 11 GHz with
efficiencies of 77 percent [43].
Class E
The drain-shunt capacitance and
series inductive reactance required
for optimum class-E operation result
in a drain impedance of R + j0.725R
at the fundamental frequency,
–j1.7846R at the second harmonic,
and proportionately smaller capacitive reactances at higher harmonics.
At microwave frequencies, class-E
operation is approximated by providing the drain with the fundamentalfrequency impedance and preferably
one or more of the harmonic
impedances [44].
An example of a microwave
approximation of class E that provides the correct fundamental and
second-harmonic impedances [44] is
shown in Figure 13. Line l2 is a quarter-wavelength long at the second
harmonic so that the open circuit at
its end is transformed to a short at
plane AA'. Line l1 in combination
with L and C is designed to be also a
quarter wavelength to translate the
short at AA' to an open at the transistor drain. The lines l1 to l4 provide
the desired impedance at the funda-
July 2003 31
mental. The implementation using an
FLK052 MESFET is shown in Figure
14 produces 0.68 W at X band with a
drain efficiency of 72 percent and
PAE of 60 percent [42].
Methods exist for providing the
proper impedances through the
fourth harmonic [45]. However, the
harmonic impedances are not critical
[30], and many variations are therefore possible. Since the transistor
often has little or no gain at the higher harmonic frequencies, those
impedances often have little or no
effect upon performance. A singlestub match is often sufficient to provide the desired impedance at the
fundamental while simultaneously
providing an adequately high
impedance at the second harmonic,
thus eliminating the need for an
extra stub and reducing a portion of
the losses associated with it. Most
microwave class-E amplifiers operate
in a suboptimum mode [46].
Demonstrated capabilities range
from 16 W with 80-percent efficiency
at UHF (LDMOS) to 100 mW with
60-percent efficiency at 10 GHz [47],
[48], [44], [49], [50], [51]. Optical sampling of the waveforms [52] has verified that these PAs do indeed operate
in class E.
Comparison
PAs configured for classes A (AB),
E, and F are compared experimentally in [50] with the following conclusions. Classes AB and F have essentially the same saturated output
power, but class F has about 15 percent higher efficiency. Class E has the
highest efficiency. Gain compression
occurs at a lower power level for class
E than for class F. For a given efficiency, class F produces more power.
For the same maximum output
power, the third order intermodulation products are about 10 dB lower
for class F than for class E. Lowerpower PAs implemented with smaller
RF power devices tend to be more
efficient than PAs implemented with
larger devices [42].
Millimeter-Wave PAs
Solid-state PAs for millimeterwave (mm-W) frequencies (30 to 100
GHz) are predominantly monolithic.
Most Ka-band PAs are based upon
pHEMT devices, while most W-band
PAs are based upon InP HEMTs.
Some use is also made of HBTs at the
lower mm-W frequencies. Class A is
used for maximum gain. Typical performance characteristics include 4 W
with 30-percent PAE at Ka band [53],
250 mW with 25-percent PAE at Q
band [54], and 200 mW with 10-percent PAE at W band [55]. Devices for
operation at mm-W are inherently
small, so large power outputs are
obtained by combining the outputs of
multiple low-power amplifiers in corporate or spatial power combiners.
6e. EXAMPLE APPLICATIONS
The following examples illustrate
the wide variety of power amplifiers
in use today:
HF/VHF Single Sideband
One of the first applications of
RF-power transistors was linear
amplification of HF single-sideband
signals. Many PAs developed by
Helge Granberg have been widely
adapted for this purpose [56, 57]. The
300-W PA for 2 to 30 MHz uses a pair
of Motorola MRF422 Si NPN transistors in a push-pull configuration. The
PA operates in class AB push-pull
from a 28-V supply and achieves a
collector efficiency of about 45 percent (CW) and a two-tone IMD ratio
of about –30 dBc. The 1-kW amplifier
is based upon a push-pull pair of
MRF154 MOSFETs and operates
from a 50-V supply. Over the frequency range of 2 to 50 MHz it achieves a
drain efficiency of about 58 percent
(CW) with an IMD rating of –30 dBc.
13.56-MHz ISM Power Sources
High-power signals at 13.56 MHz
are needed for a wide variety of
Industrial, Scientific, and Medical
(ISM) applications such as plasma
generation, RF heating, and semiconductor processing. A 400-W class-E
PA uses an International Rectifier
IRFP450LC MOSFET (normally
used for low-frequency switchingmode DC power supplies) operates
from a 120-V supply and achieves a
drain efficiency of 86 percent [58, 26].
Industrial 13.56-MHz RF power generators using class-E output stages
have been manufactured since 1992
by Dressler Hochfrequenztechnik
(Stolberg, Germany) and Advanced
Figure 13 · Idealized microwave class-E PA circuit. Figure 14 · Example X-band class-E PA.
32 High Frequency Electronics
High Frequency Design
RF POWER AMPLIFIERS
Energy Industries (Ft. Collins, CO).
They typically use RF-power
MOSFETs with 500- to 900-V breakdown voltages made by Directed
Energy or Advanced Power
Technology and produce output powers of 500 W to with 3 kW with drain
efficiencies of about 90 percent. The
Advanced Energy Industries amplifier (Figure 15) uses thick-film-hybrid
circuits to reduce size. This allows
placement inside the clean-room
facilities of semiconductor-manufacturing plants, eliminating the need
for long runs of coaxial cable from an
RF-power generator installed outside
the clean-room.
VHF FM Broadcast Transmitter
FM-broadcast transmitters (88 to
108 MHz) with power outputs from
50 W to 10 kW are manufactured by
Broadcast Electronics (Quincy,
Illinois). These transmitters use up to
32 power-combined PAs based upon
Motorola MRF151G MOSFETs. The
PAs operate in class C from a 44-V
supply and achieve a drain efficiency
of 80 percent. Typically, about 6 percent of the output power is dissipated
in the power combiners, harmonicsuppression filter, and lightning-protection circuit.
MF AM Broadcast Transmitters
Since the 1980s, AM broadcast
transmitters (530 to 1710 kHz) have
been made with class-D and -E RFoutput stages. Amplitude modulation
is produced by varying the supply
voltage of the RF PA with a high-efficiency amplitude modulator.
Transmitters made by Harris
(Mason, Ohio) produce peak-envelope
output powers of 58, 86, 150, 300, and
550 kW (unmodulated carrier powers
of 10, 15, 25, 50, and 100 kW). The
100-kW transmitter combines the
output power from 1152 transistors.
The output stages can use either
bipolars or MOSFETs, typically operate in class DE from a 300-V supply,
and achieve an efficiency of 98 percent. The output section of the Harris
3DX50 transmitter is shown in
Figure 16.
Transmitters made by Broadcast
Electronics (Quincy, IL) use class-E
RF-output stages based upon
APT6015LVR MOSFETs operating
from 130-V maximum supply voltages. They achieve drain efficiencies
of about 94 percent with peak-envelope output powers from 4.4 to 44 kW.
The 44-kW AM-10A transmitter combines outputs from 40 individual output stages.
900-MHz Cellular-Telephone
Handset
Most 900-MHz CDMA handsets
use power-amplifier modules from
vendors such as Conexant and RF
Micro Devices. These modules typically contain a single GaAs-HBT
RFIC that includes a single-ended
class-AB PA. Recently developed PA
modules also include a silicon control
IC that provides the base-bias reference voltage and can be commanded
to adjust the output-transistor base
bias to optimize efficiency while
maintaining acceptably low amplifier
distortion. over the full ranges of
temperature and output power. A typical module (Figure 17) produces 28
dBm (631 mW) at full output with a
PAE of 35 to 50 percent.
Cellular-Telephone Base
Station Transmitter
The Spectrian MCPA 3060 cellular base-station transmitter for 18401870 MHz CDMA systems provides
up to 60-W output while transmitting
a signal that may include as many 9
modulated carriers. IMD is minimized by linearizing a class-AB main
amplifier with both adaptive predistortion and adaptive feed-forward
cancellation. The adaptive control
Figure 15 · 3-kW high efficiency PA for 13.56 ISM-band operation.
(Courtesy Advanced Energy)
Figure 16 · Output section of a 50kW AM broadcast transmitter.
(Courtesy Harris)
34 High Frequency Electronics
High Frequency Design
RF POWER AMPLIFIERS
system adjusts operation as needed
to compensate for changes due to
temperature, time, and output power.
The required adjustments are
derived from continuous measurements of the system response to a
spread-spectrum pilot test signal.
The amplifier consumes a maximum
of 810 W from a 27-V supply.
S-Band Hybrid Power Module
A thick-film-hybrid power-amplifier module made by UltraRF (now
Cree Microwave) for 1805 to 1880
MHz DCS and 1930-1960 MHz PCS
is shown in Figure 18. It uses four
140-mm LDMOS FETs operating
from a 26-V drain supply. The individual PAs have 11-dB power gain
and are quadrature-combined to produce a 100-W PEP output. The average output power is 40 W for EDGE
and 7 W for CDMA, with an ACPR of
–57 dBc for EDGE and –45 dBc for
CDMA. The construction is based
upon 0.02-in. thick film with silver
metalization.
GaAs MMIC Power Amplifier
A MMIC PA for use from 8 to 14
GHz is shown in Figure 19. This
amplifier is fabricated with GaAs
HBTs and intended for used in
phased-array radar. It produces a 3W output with a PAE of approximately 40 percent [59].
References
19. H. L. Krauss, C. W. Bostian, and
F. H. Raab, Solid State Radio
Engineering, New York: Wiley, 1980.
20. R. Gupta and D. J. Allstot, “Fully
monolithic CMOS RF power amplifiers:
Recent advances,” IEEE Communi-
cations Mag., vol. 37, no. 4, pp. 94-98,
April 1999.
21. F. H. Raab and D. J. Rupp, “HF
power amplifier operates in both class
B and class D,” Proc. RF Expo West ’93,
San Jose, CA, pp. 114-124, March 17-19,
1993.
22. P. Asbeck, J. Mink, T. Itoh, and G.
Haddad, “Device and circuit approaches
for next-generation wireless communications,” Microwave J., vol. 42, no. 2, pp.
22-42, Feb. 1999.
23. N. O. Sokal and A. D. Sokal,
“Class E—a new class of high efficiency
tuned single-ended switching power
amplifiers,” IEEE J. Solid-State
Circuits, vol. SC-10, no. 3, pp. 168-176,
June 1975.
24. F. H. Raab, “Effects of circuit
variations on the class E tuned power
amplifier,” IEEE J. Solid State Circuits,
vol. SC-13, no. 2, pp. 239-247, April
1978.
25. F. H. Raab, “Effects of VSWR
upon the class-E RF-power amplifier,”
Proc. RF Expo East ’88, Philadelphia,
PA, pp. 299-309, Oct. 25-27, 1988.
26. J. F. Davis and D. B. Rutledge, “A
low-cost class-E power amplifier with
sine-wave drive,” Int. Microwave Symp.
Digest, vol. 2, pp. 1113-1116, Baltimore,
MD, June 7-11, 1998.
27. T. B. Mader and Z. B. Popovic,
“The transmission-line high-efficiency
class-E amplifier,” IEEE Microwave
and Guided Wave Letters, vol. 5, no. 9,
pp. 290-292, Sept. 1995.
28. D. C. Hamill, “Class DE inverters and rectifiers for DC-DC conversion,” PESC96 Record, vol. 1, pp. 854860, June 1996.
29. F. H. Raab, “Maximum efficiency
and output of class-F power amplifiers,”
IEEE Trans. Microwave Theory Tech.,
vol. 47, no. 6, pp. 1162-1166, June 2001.
30. F. H. Raab, “Class-E, -C, and -F
power amplifiers based upon a finite
number of harmonics,” IEEE Trans.
Microwave Theory Tech., vol. 47, no. 8,
pp. 1462-1468, Aug. 2001.
31. S. C. Cripps, RF Power
Amplifiers for Wireless Communication, Norwood, MA: Artech, 1999.
32. “A load pull system with harmonic tuning,” Microwave J., pp. 128-
132, March 1986.
33. B. Hughes, A. Ferrero, and A.
Cognata, “Accurate on-wafer power and
harmonic measurements of microwave
amplifiers and devices,” IEEE Int.
Microwave Symp. Digest, Albuquerque,
NM, pp. 1019-1022, June 1-5, 1992.
34. V. J. Tyler, “A new high-efficiency
Figure 17 · Internal view of a dualband (GSM/DCS) PA module for
cellular telephone handsets.
(Courtesy RF Micro Devices)
Figure 18 · Thick-film hybrid S-band
PA module. (Courtesy UltraRF)
Figure 19 · MMIC PA for X- and Kbands.
Acronyms Used in Part 2
BJT Bipolar Junction
Transistor
DSP Digital Signal
Processor
IC Integrated Circuit
IMD Intermodulation
Distortion
MOSFET Metal Oxide Silicon
FET
36 High Frequency Electronics
High Frequency Design
RF POWER AMPLIFIERS
high power amplifier,” The Marconi
Review, vol. 21, no. 130, pp. 96-109, Fall
1958.
35. A. V. Grebennikov, “Circuit
design technique for high-efficiency
class-F amplifiers,” Int. Microwave
Symp. Digest, vol. 2, pp. 771-774,
Boston, MA, June 13-15, 2000.
36. P. Colantonio, F. Giannini, G.
Leuzzi, and E. Limiti, “On the class-F
power-amplifier design,” RF and Micro-
wave Computer-Aided Engin-eering,
vol. 32, no. 2, pp. 129-149, March 1999.
37. A. N. Rudiakova and V. G.
Krizhanovski, “Driving waveforms for
class-F power amplifiers,” Int.
Microwave Symp. Digest, vol. 1, pp. 473476, Boston, MA, June 13-15, 2000.
38. A. Inoue, T. Heima, A. Ohta, R.
Hattori, and Y. Mitsui, “Analysis of
class-F and inverse class-F amplifiers,”
Int. Microwave Symp. Digest, vol. 2, pp.
775-778, Boston, MA, June 13-15, 2000.
39. F. van Rijs et al., “Influence of
output impedance on power added efficiency of Si-bipolar power transistors,”
Int. Microwave Symp. Digest, vol. 3, pp.
1945-1948, Boston, MA, June 13-15,
2000.
40. F. Huin, C. Duvanaud, V. Serru,
F. Robin, and E. Leclerc, “A single supply
very high power and efficiency integrated PHEMT amplifier for GSM applications,” Proc. 2000 RFIC Symp., Boston,
MA, CD-ROM, June 11-13, 2000.
41. B. Ingruber et al.,
“Rectangularly driven class-A harmonic-control amplifier,” IEEE Trans.
Microwave Theory Tech., pt. 1, vol. 46,
no. 11, pp. 1667-1672, Nov. 1998.
42. E. W. Bryerton, M. D. Weiss, and
Z. Popovic, “Efficiency of chip-level versus external power combining,” IEEE
Trans. Microwave Theory Tech., vol. 47,
no. 8, pp. 1482-1485, Aug. 1999.
43 S. Toyoda, “Push-pull power
amplifiers in the X band,” Int.
Microwave Symp. Digest, vol. 3, pp.
1433-1436, Denver, CO, June 8-13, 1997.
44. T. B. Mader and Z. Popovic, “The
transmission-line high-efficiency classE amplifier,” IEEE Micro-wave and
Guided Wave Lett., vol. 5, no. 9, pp. 290292, Sept. 1995.
45. A. J. Wilkinson and J. K. A.
Everard, “Transmission line load network topology for class E amplifiers,”
IEEE Trans. Microwave Theory Tech.,
vol. 47, no. 6, pp. 1202-1210, June 2001.
46. F. H. Raab, “Suboptimum operation of class-E power amplifiers,” Proc.
RF Technology Expo., Santa Clara, CA,
pp. 85-98, Feb. 1989.
47. S. Li, “UHF and X-band class-E
amplifiers,” Ph.D. Thesis, California
Institute of Technology, Pasadena, 1999.
48. F. J. Ortega-Gonzalez, J. L.
Jimenez-Martin, A. Asensio-Lopez, G.
Torregrosa-Penalva, “High-efficiency
load-pull harmonic controled class-E
power amplifier,” IEEE Microwave
Guided Wave Lett., vol. 8, no. 10, pp.
348-350, Oct. 1998.
49. E. Bryerton, “High-efficiency
switched-mode microwave circuits,”
Ph.D. dissertation, Univ. of Colorado,
Boulder, June 1999.
50. T. B. Mader, E. W. Bryerton, M.
Markovic, M. Forman, and Z. Popovic,
“Switched-mode high-efficiency microwave power amplifiers in a free-space
power-combiner array,” IEEE Trans.
Microwave Theory Tech., vol. 46, no. 10,
pt. I, pp. 1391-1398, Oct. 1998.
51. M. D. Weiss and Z. Popovic, “A 10
GHz high-efficiency active antenna,”
Int. Microwave Symp. Digest, vol. 2, pp.
663-666, Anaheim, CA, June 14-17,
1999.
52. M. Weiss, M. Crites, E. Bryerton,
J. Whitacker, and Z. Popovic, “"Time
domain optical sampling of nonlinear
microwave amplifiers and multipliers,”
IEEE Trans. Microwave Theory Tech.,
vol. 47, no.12, pp. 2599-2604, Dec. 1999.
53. J. J. Komiak, W. Kong, P. C.
Chao, and K. Nichols, “Fully monolithic
4 watt high efficiency Ka-band power
amplifier,” Int. Microwave Symp.
Digest, vol. 3, pp. 947-950, Anaheim,
CA, June 14-17, 1999.
54. S.-W. Chen et al., “A 60-GHz
high-efficiency monolithic power amplifier using 0.1-µm pHEMTs,” IEEE
Microwave and Guided Wave Lett.,
vol.5, pp. 201-203, June 1995.
55. D. L. Ingram et al., “Compact Wband solid-state MMIC high power
sources,” Int. Microwave Symp. Digest,
vol. 2, pp. 955-958, Boston, MA, June
13-15, 2000.
56. H. Granberg, “Get 300 watts
PEP linear across 2 to 30 MHz from
this push-pull amplifier,” Bulletin
EB27A, Motorola Semiconductor
Products, Phoenix, Feb. 1980.
57. H. Granberg, “A compact 1-kW
2-50 MHz solid-state linear amplifier,”
QEX, no. 101, pp. 3-8, July 1990. Also
AR347, Motorola Semiconductor
Products, Feb. Oct. 1990.
58. N. O. Sokal, “Class-E RF power
amplifiers ... ,” QEX, No. 204, pp. 9-20,
Jan./Feb. 2001.
59. M. Salib, A. Gupta, A. Ezis, M.
Lee, and M. Murphy, “A robust 3W high
efficiency 8-14 GHz GaAs/AlGaAs heterojunction bipolar transistor power
amplifier,” Int. Microwave Symp.
Digest, vol. 2, pp. 581-584, Baltimore,
MD, June 7-11, 1998.
Author Information
The authors of this series of articles are: Frederick H. Raab (lead
author), Green Mountain Radio
Research, e-mail: f.raab@ieee.org;
Peter Asbeck, University of
California at San Diego; Steve
Cripps, Hywave Associates; Peter B.
Kenington, Andrew Corporation;
Zoya B. Popovic, University of
Colorado; Nick Pothecary,
Consultant; John F. Sevic, California
Eastern Laboratories; and Nathan O.
Sokal, Design Automation. Readers
desiring more information should
contact the lead author.
Notes
1. In Part 1 of this series (May
2003 issue), the references contained
in Table 1 were not numbered correctly. The archived version has been
corrected and may be downloaded
from: www.highfrequencyelectronics.
com — click on “Archives,” select
“May 2003 — Vol. 2 No. 3” then click
on the article title.
2. This series has been extended
to five parts, to be published in succesive issues through January 2004.
Loading...