(3,25 mm x 3,25 mm)
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SW
C1
10 µF
L1
6.8 µH
R1
R2
VBAT
VOUT
FB
C2
2.2 µFC347 µF
LBO
PGND
LBI
PS
EN
GND
TPS61020
V
O
3.3 V Up To
200 mA
Low Battery
Output
R3
R4
R5
0.9-V To
6.5-V Input
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
96% EFFICIENT SYNCHRONOUS BOOST CONVERTER WITH 1.5-A SWITCH
FEATURES DESCRIPTION
• 96% Efficient Synchronous Boost Converter
– 200-mA Output Current From 0.9-V Input
– 500-mA Output Current From 1.8-V Input
• Output Voltage Remains Regulated When
Input Voltage Exceeds Nominal Output
Voltage
• Device Quiescent Current: 25 µA (Typ)
• Input Voltage Range: 0.9 V to 6.5 V
• Fixed and Adjustable Output Voltage Options
Up to 5.5 V
• Power Save Mode for Improved Efficiency at
Low Output Power
• Low Battery Comparator
• Low EMI-Converter (Integrated Antiringing
Switch)
• Load Disconnect During Shutdown
• Over-Temperature Protection
• Small 3 mm × 3 mm QFN-10 Package
APPLICATIONS
• All One-Cell, Two-Cell and Three-Cell Alkaline,
NiCd or NiMH or Single-Cell Li Battery
Powered Products
• Portable Audio Players tion mode. The device is packaged in a 10-pin QFN
• PDAs
• Cellular Phones
• Personal Medical Products
• Camera White LED Flash Light
The TPS6102x devices provide a power supply
solution for products powered by either a one-cell,
two-cell, or three-cell alkaline, NiCd or NiMH, or
one-cell Li-Ion or Li-polymer battery. Output currents
can go as high as 200 mA while using a single-cell
alkaline, and discharge it down to 0.9 V. It can also
be used for generating 5 V at 500 mA from a 3.3-V
rail or a Li-Ion battery. The boost converter is based
on a fixed frequency, pulse-width-modulation (PWM)
controller using a synchronous rectifier to obtain
maximum efficiency. At low load currents the converter enters the Power Save mode to maintain a
high efficiency over a wide load current range. The
Power Save mode can be disabled, forcing the
converter to operate at a fixed switching frequency.
The maximum peak current in the boost switch is
limited to a value of 1500 mA.
The TPS6102x devices keep the output voltage
regulated even when the input voltage exceeds the
nominal output voltage. The output voltage can be
programmed by an external resistor divider, or is
fixed internally on the chip. The converter can be
disabled to minimize battery drain. During shutdown,
the load is completely disconnected from the battery.
A low-EMI mode is implemented to reduce ringing
and, in effect, lower radiated electromagnetic energy
when the converter enters the discontinuous conduc-
PowerPAD™ package measuring 3 mm x 3 mm
(DRC).
PowerPAD is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
Copyright © 2003–2005, Texas Instruments Incorporated
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TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
These devices have limited built-in ESD protection. The leads should be shorted together or the device
placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates.
AVAILABLE OUTPUT VOLTAGE OPTIONS
T
A
-40 ° C to 85 ° C 3.0 V 1500 mA BDS 10-Pin QFN TPS61024DRC
(1) Contact the factory to check availability of other fixed output voltage versions.
(2) The DRC package is available taped and reeled. Add R suffix to device type (e.g., TPS61020DRCR) to order quantities of 3000 devices
per reel. Add a T suffix to the device type (i.e., TPS61020DRCT) to order quantities of 250 devices per reel.
OUTPUT VOLTAGE NOMINAL SWITCH CUR- PACKAGE
DC/DC RENT LIMIT MARKING
Adjustable 1500 mA BDR TPS61020DRC
Adjustable 800 mA BNE TPS61028DRC
3.3 V 1500 mA BDT TPS61025DRC
5 V 1500 mA BDU TPS61027DRC
(1)
PACKAGE PART NUMBER
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted)
Input voltage range on SW, VOUT, LBO, VBAT, PS, EN, FB, LBI –0.3 V to 7 V
Operating virtual junction temperature range, T
Storage temperature range T
(1) Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating
conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
stg
J
(1)
TPS6102x
–40 ° C to 150 ° C
–65 ° C to 150 ° C
(2)
DISSIPATION RATINGS TABLE
PACKAGE
DRC 48.7 ° C/W 2054 mW 21 mW/ ° C
THERMAL RESISTANCE POWER RATING DERATING FACTOR ABOVE
RECOMMENDED OPERATING CONDITIONS
Supply voltage at VBAT, V
Operating free air temperature range, T
Operating virtual junction temperature range, T
I
A
Θ
JA
J
TA≤ 25 ° C TA= 25 ° C
MIN NOM MAX UNIT
0.9 6.5 V
–40 85 ° C
–40 125 ° C
2
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TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
ELECTRICAL CHARACTERISTICS
over recommended free-air temperature range and over recommended input voltage range (typical at an ambient temperature
range of 25 ° C) (unless otherwise noted)
DC/DC STAGE
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
V
I
V
O
V
FB
f Oscillator frequency 480 600 720 kHz
I
SW
I
SW
Minimum input voltage for start-up RL= 120 Ω 0.9 1.2
Input voltage range, after start-up 0.9 6.5
TPS61020 and TPS61028 output
voltage range
TPS61020 and TPS61028 feedback
voltage
Switch current limit (TPS61020,
TPS61024, TPS61025, TPS61027)
VOUT= 3.3 V 1200 1500 1800 mA
1.8 5.5 V
490 500 510 mV
Switch current limit (TPS61028) VOUT= 3.3 V 800 mA
Start-up current limit 0.4 x I
SW
SWN switch on resistance VOUT= 3.3 V 260 m Ω
SWP switch on resistance VOUT= 3.3 V 290 m Ω
Total accuracy (including line and
load regulation)
± 3%
Line regulation 0.6%
Load regulation 0.6%
Quiescent current
VBAT 1 3 µA
VOUT 25 45 µA
Shutdown current V
IO= 0 mA, V
VOUT = 3.3 V, TA= 25 ° C
= 0 V, VBAT = 1.2 V, TA= 25 ° C 0.1 1 µA
EN
= VBAT = 1.2 V,
EN
V
mA
CONTROL STAGE
V
UVLO
V
IL
V
OL
V
lkg
V
IL
V
IH
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
Under voltage lockout threshold V
LBI voltage threshold V
voltage decreasing 0.8 V
LBI
voltage decreasing 490 500 510 mV
LBI
LBI input hysteresis 10 mV
LBI input current EN = VBAT or GND 0.01 0.1 µA
LBO output low voltage VO= 3.3 V, IOI= 100 µA 0.04 0.4 V
LBO output leakage current V
= 7 V 0.01 0.1 µA
LBO
EN, PS input low voltage 0.2 × VBAT V
EN, PS input high voltage 0.8 × VBAT V
EN, PS input current Clamped on GND or VBAT 0.01 0.1 µA
Overtemperature protection 140 ° C
Overtemperature hysteresis 20 ° C
3
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FB
LBO
GND
VBAT
SW
VOUT
LBI
PS
EN
PGND
DRC PACKAGE
(TOP VIEW)
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
PIN ASSIGNMENTS
Terminal Functions
TERMINAL
NAME NO.
EN 1 I Enable input. (1/VBAT enabled, 0/GND disabled)
FB 3 I Voltage feedback of adjustable versions
GND 5 Control / logic ground
LBI 7 I Low battery comparator input (comparator enabled with EN), may not be left floating, should be connected to
LBO 4 O Low battery comparator output (open drain)
PS 8 I Enable/disable power save mode (1/VBAT disabled, 0/GND enabled)
SW 9 I Boost and rectifying switch input
PGND 10 Power ground
VBAT 6 I Supply voltage
VOUT 2 O Boost converter output
PowerPAD™ Must be soldered to achieve appropriate power dissipation. Should be connected to PGND.
I/O DESCRIPTION
GND or VBAT if comparator is not used
4
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Anti-
Ringing
Gate
Control
PGND
Regulator
Error
Amplifier
PGND
Control Logic
Oscillator
Temperature
Control
VOUT
PGND
FB
SW
VBAT
EN
PS
GND
LBI
_
+
_
+
Backgate
Control
V
max
Control
VOUT
10 kΩ
20 pF
_
+
V
ref
= 0.5 V
GND
LDO
Low Battery
Comparator
_
+
V
ref
= 0.5 V
GND
FUNCTIONAL BLOCK DIAGRAM (TPS61020, TPS61028)
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
5
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SW
C1
10 µF
Power
Supply
L1
6.8 µH
R1
R2
VBAT
VOUT
FB
C2
2.2 µFC347 µF
LBO
PGND
LBI
PS
EN
GND
TPS6102x
List of Components:
U1 = TPS6102xDRC
L1 = EPCOS B82462−G4682
C1, C2 = X7R/X5R Ceramic
C3 = Low ESR Tantalum
V
CC
Boost Output
Control Output
R3
R4
R5
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
PARAMETER MEASUREMENT INFORMATION
TYPICAL CHARACTERISTICS
Table of Graphs
Maximum output current vs Input voltage 1
vs Output current (TPS61020) 2
vs Output current (TPS61025) 3
Efficiency vs Output current (TPS61027) 4
vs Input voltage (TPS61025) 5
vs Input voltage (TPS61027) 6
Output voltage
No load supply current into VBAT vs Input voltage 9
No load supply current into VOUT vs Input voltage 10
Waveforms
vs Output current (TPS61025) 7
vs Output current (TPS61027) 8
Output voltage in continuous mode (TPS61025) 11
Output voltage in continuous mode (TPS61027) 12
Output voltage in power save mode (TPS61025) 13
Output voltage in power save mode (TPS61027) 14
Load transient response (TPS61025) 15
Load transient response (TPS61027) 16
Line transient response (TPS61025) 17
Line transient response (TPS61027) 18
Start-up after enable (TPS61025) 19
Start-up after enable (TPS61027) 20
FIGURE
6
www.ti.com
0
200
400
600
800
1000
1200
1400
0.9 1.7 2.5 3.3 4.1
Maximum Output Current - mA
4.9 6.55.7
VI - Input Voltage - V
VO = 3.3 V
VO = 5 V
VO = 1.8 V
0
10
20
30
40
50
60
70
80
90
100
1 10 100 1000
VBAT = 0.9 V
VBAT = 1.8 V
VO = 1.8 V
Efficiency - %
IO - Output Current - mA
0
10
20
30
40
50
60
70
80
90
100
1 10 100 1000
Efficiency - %
IO - Output Current - mA
VBAT = 2.4 V
VBAT = 0.9 V
VO = 3.3 V
VBAT = 1.8 V
0
10
20
30
40
50
60
70
80
90
100
1 10 100 1000
Efficiency - %
IO - Output Current - mA
VBAT = 1.2 V
VBAT = 3.6 V
VBAT = 2.4 V
VO = 5 V
VBAT = 1.8 V
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
MAXIMUM OUTPUT CURRENT TPS61020
vs EFFICIENCY
INPUT VOLTAGE vs
Figure 1. Figure 2.
TPS61025 TPS61027
EFFICIENCY EFFICIENCY
vs vs
OUTPUT CURRENT OUTPUT CURRENT
OUTPUT CURRENT
Figure 3. Figure 4.
7
www.ti.com
50
55
60
65
70
75
80
85
90
95
100
0.9 1.4 1.9 2.4 2.9 3.4 3.9 4.4 4.9
Efficiency - %
VI - Input Voltage - V
IO = 100 mA
VO = 3.3 V
IO = 10 mA
IO = 250 mA
50
55
60
65
70
75
80
85
90
95
100
0.9 1.4 1.9 2.4 2.9 3.4 3.9 4.4 4.9 5.4 5.9
Efficiency - %
VI - Input Voltage - V
IO = 100 mA
VO = 5 V
IO = 250 mA
IO = 10 mA
6.4
3.20
3.25
3.30
3.35
1 10 100 1000
- Output Voltage - V
V
O
IO - Output Current - mA
VBAT = 2.4 V
VO = 3.3 V
- Output Voltage - V
V
O
IO - Output Current - mA
VBAT = 3.6 V
4.80
4.85
4.90
4.95
5
5.05
5.10
1 10 100 1000
VO = 5 V
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
TPS61025 TPS61027
EFFICIENCY EFFICIENCY
vs vs
INPUT VOLTAGE INPUT VOLTAGE
TPS61025 TPS61027
OUTPUT VOLTAGE OUTPUT VOLTAGE
OUTPUT CURRENT OUTPUT CURRENT
8
Figure 5. Figure 6.
vs vs
Figure 7. Figure 8.
www.ti.com
0.9 1.5 2 2.5 3 3.5 4
No Load Supply Current Into VOUT -
4.5 5 6 6.55.5
VI- Input Voltage - V
Aµ
-0.2
4.8
9.8
14.8
19.8
24.8
29.8
34.8
TA = 85°C
TA = -40°C
TA = 25°C
0
0.2
0.4
0.6
0.8
1.2
1.6
0.9 1.5 2 2.5 3 3.5 4
No Load Supply Current Into VBAT -
4.5 5 6 6.55.5
VI- Input Voltage - V
Aµ
TA = 85°C
TA = 25°C
1
1.4
TA = -40°C
t - Time - 1 µs/div
VI = 3.6 V ,
RL = 25 Ω,
VO = 5 V
Output Voltage
20 mV/div
Inductor Current
200 mA/div
t - Time - 1 µs/div
VI = 1.2 V ,
RL = 33 Ω,
VO = 3.3 V
Output Voltage
20 mV/div
Inductor Current
200 mA/div
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
NO LOAD SUPPLY CURRENT INTO VBAT NO LOAD SUPPLY CURRENT INTO VOUT
OUTPUT VOLTAGE IN CONTINUOUS MODE OUTPUT VOLTAGE IN CONTINUOUS MODE
vs vs
INPUT VOLTAGE INPUT VOLTAGE
Figure 9. Figure 10.
TPS61025 TPS61027
Figure 11. Figure 12.
9
www.ti.com
t - Time - 50 µs/div
VI = 1.2 V ,
RL = 330 Ω,
VO = 3.3 V
Output Voltage
20 mV/div, AC
Inductor Current
100 mA/div, DC
t - Time - 50 µs/div
VI = 3.6 V ,
RL = 250 Ω,
VO = 5 V
Output Voltage
50 mV/div, AC
Inductor Current
200 mA/div, DC
t - Time - 2 ms/div
Output Current
100 mA/div, DC
Output Voltage
20 mV/div, AC
VI = 3.6 V ,
IL = 100 mA to 200 mA,
VO = 5 V
t - Time - 2 ms/div
VI = 1.2 V ,
IL = 100 mA to 200 mA,
VO = 3.3 V
Output Current
100 mA/div, DC
Output Voltage
20 mV/div, AC
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
OUTPUT VOLTAGE IN POWER SAVE MODE OUTPUT VOLTAGE IN POWER SAVE MODE
LOAD TRANSIENT RESPONSE LOAD TRANSIENT RESPONSE
TPS61025 TPS61027
Figure 13. Figure 14.
TPS61025 TPS61027
10
Figure 15. Figure 16.
www.ti.com
t - Time - 2 ms/div
VI = 1.8 V to 2.4 V ,
RL = 33 Ω,
VO = 3.3 V
Input Voltage
500 mV/div, AC
Output Voltage
20 mV/div, AC
t - Time - 2 ms/div
VI = 3 V to 3.6 V ,
RL = 25 Ω,
VO = 5 V
Input Voltage
500 mV/div, AC
Output Voltage
20 mV/div, AC
t - Time - 1 ms/div
Output Voltage
1 V/div, DC
Inductor Current
200 mA/div, DC
Voltage At SW
2 V/div, DC
Enable
5 V/div, DC
VI = 2.4V ,
RL = 33 Ω,
VO = 3.3 V
t - Time - 500 s/divm
VI= 3.6 V,
RL= 50 W,
V
O
= 5 V
Output Voltage
2 V/div, DC
Inductor Current
200 mA/div, DC
Voltage At SW
2 V/div, DC
Enable
5 V/div, DC
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
LINE TRANSIENT RESPONSE LINE TRANSIENT RESPONSE
START-UP AFTER ENABLE START-UP AFTER ENABLE
TPS61025 TPS61027
Figure 17. Figure 18.
TPS61025 TPS61027
Figure 19. Figure 20.
11
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TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
DETAILED DESCRIPTION
CONTROLLER CIRCUIT
The controller circuit of the device is based on a fixed frequency multiple feedforward controller topology. Input
voltage, output voltage, and voltage drop on the NMOS switch are monitored and forwarded to the regulator. So
changes in the operating conditions of the converter directly affect the duty cycle and must not take the indirect
and slow way through the control loop and the error amplifier. The control loop, determined by the error amplifier,
only has to handle small signal errors. The input for it is the feedback voltage on the FB pin or, at fixed output
voltage versions, the voltage on the internal resistor divider. It is compared with the internal reference voltage to
generate an accurate and stable output voltage.
The peak current of the NMOS switch is also sensed to limit the maximum current flowing through the switch and
the inductor. The typical peak current limit is set to 1500 mA. An internal temperature sensor prevents the device
from getting overheated in case of excessive power dissipation.
Synchronous Rectifier
The device integrates an N-channel and a P-channel MOSFET transistor to realize a synchronous rectifier.
Because the commonly used discrete Schottky rectifier is replaced with a low RDS(ON) PMOS switch, the power
conversion efficiency reaches 96%. To avoid ground shift due to the high currents in the NMOS switch, two
separate ground pins are used. The reference for all control functions is the GND pin. The source of the NMOS
switch is connected to PGND. Both grounds must be connected on the PCB at only one point close to the GND
pin. A special circuit is applied to disconnect the load from the input during shutdown of the converter. In
conventional synchronous rectifier circuits, the backgate diode of the high-side PMOS is forward biased in
shutdown and allows current flowing from the battery to the output. This device however uses a special circuit
which takes the cathode of the backgate diode of the high-side PMOS and disconnects it from the source when
the regulator is not enabled (EN = low).
The benefit of this feature for the system design engineer is that the battery is not depleted during shutdown of
the converter. No additional components have to be added to the design to make sure that the battery is
disconnected from the output of the converter.
Down Regulation
In general, a boost converter only regulates output voltages which are higher than the input voltage. This device
operates differently. For example, it is able to regulate 3.0 V at the output with two fresh alkaline cells at the input
having a total cell voltage of 3.2 V. Another example is powering white LEDs with a forward voltage of 3.6 V from
a fully charged Li-Ion cell with an output voltage of 4.2 V. To control these applications properly, a down
conversion mode is implemented.
If the input voltage reaches or exceeds the output voltage, the converter changes to the conversion mode. In this
mode, the control circuit changes the behavior of the rectifying PMOS. It sets the voltage drop across the PMOS
as high as needed to regulate the output voltage. This means the power losses in the converter increase. This
has to be taken into account for thermal consideration. The down conversion mode is automatically turned off as
soon as the input voltage falls about 50 mV below the output voltage. For proper operation in down conversion
mode the output voltage should not be programmed below 50% of the maximum input voltage which can be
applied.
Device Enable
The device is put into operation when EN is set high. It is put into a shutdown mode when EN is set to GND. In
shutdown mode, the regulator stops switching, all internal control circuitry including the low-battery comparator is
switched off, and the load is isolated from the input (as described in the Synchronous Rectifier Section). This
also means that the output voltage can drop below the input voltage during shutdown. During start-up of the
converter, the duty cycle and the peak current are limited in order to avoid high peak currents drawn from the
battery.
12
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0
0.05
0.1
0.15
0.2
0.25
0.3
0.35
0
0.5
1
1.5
2
2.5
3
3.5
4
4.5
5
VBAT = 5 V
VBAT = 3.6 V
VBAT = 2.4 V
VBAT = 1.8 V
VBAT = 1.2 V
VO − Output Voltage − V
Precharge Current − A
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
DETAILED DESCRIPTION (continued)
Undervoltage Lockout
An undervoltage lockout function prevents device start-up if the supply voltage on VBAT is lower than
approximately 0.8 V. When in operation and the battery is being discharged, the device automatically enters the
shutdown mode if the voltage on VBAT drops below approximately 0.8 V. This undervoltage lockout function is
implemented in order to prevent the malfunctioning of the converter.
Softstart and Short Circuit Protection
When the device enables, the internal startup cycle starts with the first step, the precharge phase. During
precharge, the rectifying switch is turned on until the output capacitor is charged to a value close to the input
voltage. The rectifying switch is current limited during that phase. The current limit increases with the output
voltage. This circuit also limits the output current under short circuit conditions at the output. Figure 21 shows the
typical precharge current vs output voltage for specific input voltages:
Figure 21. Precharge and Short Circuit Current
After charging the output capacitor to the input voltage, the device starts switching. If the input voltage is below
1.4 V the device works with a fixed duty cycle of 50% until the output voltage reaches 1.4 V. After that the duty
cycle is set depending on the input output voltage ratio. Until the output voltage reaches its nominal value, the
boost switch current limit is set to 40% of its nominal value to avoid high peak currents at the battery during
startup. As soon as the output voltage is reached, the regulator takes control and the switch current limit is set
back to 100%.
Power Save Mode
The PS pin can be used to select different operation modes. To enable power save, PS must be set low. Power
save mode is used to improve efficiency at light load. In power save mode the converter only operates when the
output voltage trips below a set threshold voltage. It ramps up the output voltage with one or several pulses and
goes again into power save mode once the output voltage exceeds the set threshold voltage. This power save
mode can be disabled by setting the PS to VBAT. In down conversion mode, power save mode is always active
and the device cannot be forced into fixed frequency operation at light loads.
13
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TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
DETAILED DESCRIPTION (continued)
Low Battery Detector Circuit—LBI/LBO
The low-battery detector circuit is typically used to supervise the battery voltage and to generate an error flag
when the battery voltage drops below a user-set threshold voltage. The function is active only when the device is
enabled. When the device is disabled, the LBO pin is high-impedance. The switching threshold is 500 mV at LBI.
During normal operation, LBO stays at high impedance when the voltage, applied at LBI, is above the threshold.
It is active low when the voltage at LBI goes below 500 mV.
The battery voltage, at which the detection circuit switches, can be programmed with a resistive divider
connected to the LBI pin. The resistive divider scales down the battery voltage to a voltage level of 500 mV,
which is then compared to the LBI threshold voltage. The LBI pin has a built-in hysteresis of 10 mV. See the
application section for more details about the programming of the LBI threshold. If the low-battery detection
circuit is not used, the LBI pin should be connected to GND (or to VBAT) and the LBO pin can be left
unconnected. Do not let the LBI pin float.
Low-EMI Switch
The device integrates a circuit that removes the ringing that typically appears on the SW node when the
converter enters discontinuous current mode. In this case, the current through the inductor ramps to zero and the
rectifying PMOS switch is turned off to prevent a reverse current flowing from the output capacitors back to the
battery. Due to the remaining energy that is stored in parasitic components of the semiconductor and the
inductor, a ringing on the SW pin is induced. The integrated antiringing switch clamps this voltage to VBAT and
therefore dampens ringing.
14
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R3 R4
V
O
V
FB
1 180 k
V
O
500 mV
1
C
parR3
20 pF
200 k
R4
1
SW
C1Power
Supply
L1
R1
R2
VBAT
VOUT
FB
C2 C3
LBO
PGND
LBI
PS
EN
GND
TPS61020
V
CC
Boost Output
Control Output
R3
R4
R5
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
APPLICATION INFORMATION
DESIGN PROCEDURE
The TPS6102x dc/dc converters are intended for systems powered by a single up to triple cell Alkaline, NiCd,
NiMH battery with a typical terminal voltage between 0.9 V and 6.5 V. They can also be used in systems
powered by one-cell Li-Ion or Li-Polymer with a typical voltage between 2.5 V and 4.2 V. Additionally, any other
voltage source with a typical output voltage between 0.9 V and 6.5 V can power systems where the TPS6102x is
used.
Programming the Output Voltage
The output voltage of the TPS61020 dc/dc converter can be adjusted with an external resistor divider. The typical
value of the voltage at the FB pin is 500 mV. The maximum recommended value for the output voltage is 5.5 V.
The current through the resistive divider should be about 100 times greater than the current into the FB pin. The
typical current into the FB pin is 0.01 µA, and the voltage across R4 is typically 500 mV. Based on those two
values, the recommended value for R4 should be lower than 500 k Ω , in order to set the divider current at 1 µA or
higher. Because of internal compensation circuitry the value for this resistor should be in the range of 200 k Ω .
From that, the value of resistor R3, depending on the needed output voltage (V
Equation 1 :
), can be calculated using
O
If as an example, an output voltage of 3.3 V is needed, a 1.0-M Ω resistor should be chosen for R3. If for any
reason the value for R4 is chosen significantly lower than 200 k Ω additional capacitance in parallel to R3 is
recommended, in case the device shows instable regulation of the output voltage. The required capacitance
value can be easily calculated using Equation 2 :
Figure 22. Typical Application Circuit for Adjustable Output Voltage Option
(1)
(2)
Programming the LBI/LBO Threshold Voltage
The current through the resistive divider should be about 100 times greater than the current into the LBI pin. The
typical current into the LBI pin is 0.01 µA, and the voltage across R2 is equal to the LBI voltage threshold that is
generated on-chip, which has a value of 500 mV. The recommended value for R2 is therefore in the range of 500
k Ω . From that, the value of resistor R1, depending on the desired minimum battery voltage V
calculated using Equation 3 .
can be
BAT,
15
www.ti.com
R1 R2
V
BAT
V
LBIthreshold
1 390 k
V
BAT
500 mV
1
IL I
OUT
V
OUT
V
BAT
0.8
L
V
BAT
V
OUT–VBAT
IL ƒ V
OUT
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
The output of the low battery supervisor is a simple open-drain output that goes active low if the dedicated
battery voltage drops below the programmed threshold voltage on LBI. The output requires a pullup resistor with
a recommended value of 1 M Ω . If not used, the LBO pin can be left floating or tied to GND.
Inductor Selection
A boost converter normally requires two main passive components for storing energy during the conversion. A
boost inductor and a storage capacitor at the output are required. To select the boost inductor, it is
recommended to keep the possible peak inductor current below the current limit threshold of the power switch in
the chosen configuration. For example, the current limit threshold of the TPS6102xs switch is 1800 mA at an
output voltage of 5 V. The highest peak current through the inductor and the switch depends on the output load,
the input (V
using Equation 4 :
For example, for an output current of 200 mA at 3.3 V, at least 920 mA of average current flows through the
inductor at a minimum input voltage of 0.9 V.
The second parameter for choosing the inductor is the desired current ripple in the inductor. Normally, it is
advisable to work with a ripple of less than 20% of the average inductor current. A smaller ripple reduces the
magnetic hysteresis losses in the inductor, as well as output voltage ripple and EMI. But in the same way,
regulation time at load changes rises. In addition, a larger inductor increases the total system costs. With those
parameters, it is possible to calculate the value for the inductor by using Equation 5 :
), and the output voltage (V
BAT
). Estimation of the maximum average inductor current can be done
OUT
(3)
(4)
Parameter f is the switching frequency and ∆ ILis the ripple current in the inductor, i.e., 20% × IL. In this example,
the desired inductor has the value of 5.5 µH. With this calculated value and the calculated currents, it is possible
to choose a suitable inductor. In typical applications a 6.8 µH inductance is recommended. The device has been
optimized to operate with inductance values between 2.2 µH and 22 µH. Nevertheless operation with higher
inductance values may be possible in some applications. Detailed stability analysis is then recommended. Care
has to be taken that load transients and losses in the circuit can lead to higher currents as estimated in
Equation 5 . Also, the losses in the inductor caused by magnetic hysteresis losses and copper losses are a major
parameter for total circuit efficiency.
The following inductor series from different suppliers have been used with the TPS6102x converters:
Table 1. List of Inductors
VENDOR INDUCTOR SERIES
Sumida
Wurth Elektronik
EPCOS B82462-G4
Cooper Electronics Technologies
CDRH4D28
CDRH5D28
7447789
744042
SD25
SD20
(5)
16
www.ti.com
C
min
I
OUT
V
OUT
V
BAT
ƒ V V
OUT
V
ESR
I
OUT
R
ESR
A
REG
d
V
FB
4 (R3 R4)
R4 (1 i 0.9 s)
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
Capacitor Selection
Input Capacitor
At least a 10-µF input capacitor is recommended to improve transient behavior of the regulator and EMI behavior
of the total power supply circuit. A ceramic capacitor or a tantalum capacitor with a 100-nF ceramic capacitor in
parallel, placed close to the IC, is recommended.
Output Capacitor
The major parameter necessary to define the output capacitor is the maximum allowed output voltage ripple of
the converter. This ripple is determined by two parameters of the capacitor, the capacitance and the ESR. It is
possible to calculate the minimum capacitance needed for the defined ripple, supposing that the ESR is zero, by
using Equation 6 :
Parameter f is the switching frequency and ∆ V is the maximum allowed ripple.
With a chosen ripple voltage of 10 mV, a minimum capacitance of 24 µF is needed. The total ripple is larger due
to the ESR of the output capacitor. This additional component of the ripple can be calculated using Equation 7 :
An additional ripple of 16 mV is the result of using a tantalum capacitor with a low ESR of 80 m Ω . The total ripple
is the sum of the ripple caused by the capacitance and the ripple caused by the ESR of the capacitor. In this
example, the total ripple is 26 mV. Additional ripple is caused by load transients. This means that the output
capacitor has to completely supply the load during the charging phase of the inductor. A reasonable value of the
output capacitance depends on the speed of the load transients and the load current during the load change.
With the calculated minimum value of 24 µF and load transient considerations the recommended output
capacitance value is in a 47 to 100 µF range. For economical reasons, this is usually a tantalum capacitor.
Therefore, the control loop has been optimized for using output capacitors with an ESR of above 30 m Ω . The
minimum value for the output capacitor is 10 µF.
(6)
(7)
Small Signal Stability
When using output capacitors with lower ESR, like ceramics, the adjustable voltage version is recommended.
The missing ESR can be compensated in the feedback divider. Typically a capacitor in the range of 4.7 pF in
parallel to R3 helps to obtain small signal stability with lowest ESR output capacitors. For more detailed analysis,
the small signal transfer function of the error amplifier and the regulator, which is given in Equation 8 , can be
used:
Layout Considerations
As for all switching power supplies, the layout is an important step in the design, especially at high peak currents
and high switching frequencies. If the layout is not carefully done, the regulator could show stability problems as
well as EMI problems. Therefore, use wide and short traces for the main current path and for the power ground
tracks. The input capacitor, output capacitor, and the inductor should be placed as close as possible to the IC.
Use a common ground node for power ground and a different one for control ground to minimize the effects of
ground noise. Connect these ground nodes at any place close to one of the ground pins of the IC.
The feedback divider should be placed as close as possible to the control ground pin of the IC. To lay out the
control ground, it is recommended to use short traces as well, separated from the power ground traces. This
avoids ground shift problems, which can occur due to superimposition of power ground current and control
ground current.
(8)
17
www.ti.com
SW
C1
10 µF
L1
6.8 µH
R1
R2
VBAT
VOUT
FB
C2
2.2 µFC3100 µF
LBO
PGND
LBI
PS
EN
GND
TPS61027
List of Components:
U1 = TPS61027DRC
L1 = EPCOS B82462-G4682
C1, C2 = X7R,X5R Ceramic
C3 = Low ESR Tantalum
VCC 5 V
Boost Output
LBO
R5
Battery
Input
SW
C1
10 µF
L1
6.8 µH
R1
R2
VBAT
VOUT
FB
C2
2.2 µFC347 µF
LBO
PGND
LBI
PS
EN
GND
TPS61027
List of Components:
U1 = TPS61027DRC
L1 = EPCOS B82462-G4682
C1, C2 = X7R,X5R Ceramic
C3 = Low ESR Tantalum
VCC 5 V
Boost Output
LBO
R5
Battery
Input
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
APPLICATION EXAMPLES
Figure 23. Power Supply Solution for Maximum Output Power Operating From a Single Alkaline Cell
Figure 24. Power Supply Solution for Maximum Output Power Operating From a Dual/Triple Alkaline Cell
18
or Single Li-Ion Cell
www.ti.com
PS
C1
L1
TPS61020
GND
VBAT
LBI
EN
SW VOUT
FB
LBO
PGND
C2
R1
D1
C3
10 µF
4.7 µH
22 µF
10 nF
Input
0.9 V to 6.5 V
List of Components:
U1 = TPS61020DRC
L1 = Sumida CDRH2D16-4R7
C1, C2, C3 = X7R, X5R Ceramic
D1 = White LED
C1
L1
TPS61020
C2
D1
C3
Q1
PS
GND
VBAT
LBI
EN
SW VOUT
FB
LBO
PGND
10 Fm
4.7 Hm
Input
1.8 V to 6.5 V
List of Components:
U1 = TPS61020DRC
L1 = TDK VLF3010AT 4R7MR70
C1, C2, C3 = X7R, X5R Ceramic
D1 = OSRAM LWW57G
Q1 = Vishay SI1012R
22 Hm
R1
1.5 MW
R2
200 kW
2200 pF
Flashlight
Control
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
Figure 25. Power Supply Solution for Powering White LED's With LED Currents Below 150 mA in Lighting
Applications
Figure 26. Simple Power Supply Solution for Powering White LED Flashlights
19
www.ti.com
SW
C1
10 µF
L1
6.8 µH
R1
R2
VBAT
VOUT
R5
C2
2.2 µFC347 µF
LBO
PGND
LBI
PS
EN
GND
TPS61027
List of Components:
U1 = TPS61027DRC1
L1 = EPCOS B82462-G4682
C3, C5, C6, = X7R,X5R Ceramic
C3 = Low ESR Tantalum
DS1 = BAT54S
LBO
C5
0.1 µF
DS1
C6
1 µF
V
CC2
10 V
Unregulated
Auxiliary Output
Battery
Input
FB
V
CC1
5 V
Boost Main Output
SW
C1
10 µF
L1
6.8 µH
R1
R2
VBAT
VOUT
R5
C2
2.2 µFC347 µF
LBO
PGND
LBI
PS
EN
GND
TPS61027
List of Components:
U1 = TPS61027DRC
L1 = EPCOS B82462-G4682
C1, C2, C5, C6 = X7R,X5R Ceramic
C3 = Low ESR Tantalum
DS1 = BAT54S
LBO
C5
0.1 µF
DS1
C6
1 µF
V
CC2
-5 V
Unregulated
Auxiliary Output
Battery
Input
FB
V
CC1
5 V
Boost Main Output
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
Figure 27. Power Supply Solution With Auxiliary Positive Output Voltage
20
Figure 28. Power Supply Solution With Auxiliary Negative Output Voltage
www.ti.com
P
D(MAX)
T
J(MAX)
T
A
R
JA
125°C 85°C
48.7 °CW
820 mW
TPS61020, TPS61024
TPS61025, TPS61027, TPS61028
SLVS451C – SEPTEMBER 2003 – REVISED JUNE 2005
THERMAL INFORMATION
Implementation of integrated circuits in low-profile and fine-pitch surface-mount packages typically requires
special attention to power dissipation. Many system-dependent issues such as thermal coupling, airflow, added
heat sinks and convection surfaces, and the presence of other heat-generating components affect the
power-dissipation limits of a given component.
Three basic approaches for enhancing thermal performance are listed below.
• Improving the power dissipation capability of the PCB design
• Improving the thermal coupling of the component to the PCB
• Introducing airflow in the system
The maximum recommended junction temperature (T
resistance of the 10-pin QFN 3 × 3 package (DRC) is R
regulator operation is assured to a maximum ambient temperature T
dissipation is about 820 mW. More power can be dissipated if the maximum ambient temperature of the
application is lower.
) of the TPS6102x devices is 125 ° C. The thermal
J
= 48.7 ° C/W, if the PowerPAD is soldered. Specified
Θ JA
of 85 ° C. Therefore, the maximum power
A
(9)
21
PACKAGE OPTION ADDENDUM
www.ti.com
8-Aug-2005
PACKAGING INFORMATION
Orderable Device Status
(1)
Package
Type
Package
Drawing
Pins Package
Qty
Eco Plan
TPS61020DRCR ACTIVE SON DRC 10 3000 Green (RoHS &
no Sb/Br)
TPS61020DRCRG4 ACTIVE SON DRC 10 3000 Green (RoHS &
no Sb/Br)
TPS61024DRCR ACTIVE SON DRC 10 3000 Green (RoHS &
no Sb/Br)
TPS61024DRCRG4 ACTIVE SON DRC 10 3000 Green (RoHS &
no Sb/Br)
TPS61025DRCR ACTIVE SON DRC 10 3000 Green (RoHS &
no Sb/Br)
TPS61025DRCRG4 ACTIVE SON DRC 10 3000 Green (RoHS &
no Sb/Br)
TPS61027DRCR ACTIVE SON DRC 10 3000 Green (RoHS &
no Sb/Br)
TPS61027DRCRG4 ACTIVE SON DRC 10 3000 Green (RoHS &
no Sb/Br)
TPS61028DRCR ACTIVE SON DRC 10 3000 TBD CUNIPDAU Level-2-235C-1 YEAR
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Lead/Ball Finish MSL Peak Temp
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
(3)
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS) or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
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Addendum-Page 1
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