TPS6104x Low-Power DC-DC Boost Converter in SOT-23 and WSON Packages
1Features3Description
1
•1.8-V to 6-V Input Voltage Range
•Adjustable Output Voltage Range up to 28 V
•400-mA (TPS61040) and 250-mA (TPS61041)
Internal Switch Current
•Up to 1-MHz Switching Frequency
•28-μA Typical No-Load Quiescent Current
•1-μA Typical Shutdown Current
•Internal Soft Start
•Available in SOT23-5, TSOT23-5,
and 2-mm × 2-mm × 0.8-mm WSON Packages
2Applications
•LCD Bias Supply
•White-LED Supply for LCD Backlights
•Digital Still Camera
•PDAs, Organizers, and Handheld PCs
•Cellular Phones
•Internet Audio Players
•Standard 3.3-V or 5-V to 12-V Conversion
The TPS6104x is a high-frequency boost converter
dedicated for small to medium LCD bias supply and
white LED backlight supplies. The device is ideal to
generate output voltages up to 28 V from a dual-cell
NiMH/NiCd or a single-cell Li-Ion battery. The part
can also be used to generate standard 3.3-V or 5-V
to 12-V power conversions.
The TPS6104x operates with a switching frequency
up to 1 MHz. This frequency allows the use of small
external components using ceramic as well as
tantalum output capacitors. Together with the thin
WSON package, the TPS6104x gives a very small
overall solution size. The TPS61040 device has an
internal 400-mAswitch current limit, while the
TPS61041 device has a 250-mA switch current limit,
offering lower output voltage ripple and allows the
use of a smaller form factor inductor for lower power
applications. The low quiescent current (typically 28
μA) together with an optimized control scheme,
allows device operation at very high efficiencies over
the entire load current range.
Device Information
PART NUMBERPACKAGEBODY SIZE (NOM)
TPS61040SOT (5)2.90 mm ×1.60 mm
TPS61041
(1) For all available packages, see the orderable addendum at
the end of the datasheet.
TPS61040,TPS61041
(1)
SOT-23 (5)2.90 mm × 1.60 mm
WSON (6)2.00 mm × 2.00 mm
SOT-23 (5)2.90 mm ×1.60 mm
WSON (6)2.00 mm × 2.00 mm
4Typical Application Schematic
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
Changes from Revision G (December 2014) to Revision HPage
•Added 500 µs/div label to X-axis of Figure 15. ................................................................................................................... 15
Changes from Revision F (December 2010) to Revision GPage
•Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section.................................................................................................. 1
over operating free-air temperature range (unless otherwise noted)
Supply voltages on pin V
Voltages on pins EN, FB
Switch voltage on pin SW
Operating junction temperature, T
Storage temperature, T
(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
(2) All voltage values are with respect to network ground terminal.
(2)
IN
(2)
(2)
J
stg
7.2 ESD Ratings
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001
V
(ESD)
(1) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process. Manufacturing with
(2) JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process. Manufacturing with
Electrostatic dischargeV
Charged-device model (CDM), per JEDEC specification JESD22-±750
(2)
C101
less than 500-V HBM is possible with the necessary precautions. Pins listed as ±XXX V may actually have higher performance.
less than 250-V CDM is possible with the necessary precautions. Pins listed as ±YYY V may actually have higher performance.
The TPS6104x is a high-frequency boost converter dedicated for small to medium LCD bias supply and white
LED backlight supplies. The device is ideal to generate output voltages up to 28 V from a dual-cell NiMH/NiCd or
a single cell device Li-Ion battery.
8.2 Functional Block Diagram
8.3 Feature Description
8.3.1 Peak Current Control
The internal switch turns on until the inductor current reaches the typical dc current limit (I
(TPS61040) or 250 mA (TPS61041). Due to the internal propagation delay of typical 100 ns, the actual current
exceeds the dc current limit threshold by a small amount. The typical peak current limit can be calculated:
The higher the input voltage and the lower the inductor value, the greater the peak.
By selecting the TPS6104x, it is possible to tailor the design to the specific application current limit requirements.
A lower current limit supports applications requiring lower output power and allows the use of an inductor with a
lower current rating and a smaller form factor. A lower current limit usually has a lower output voltage ripple as
well.
All inductive step-up converters exhibit high inrush current during start-up if no special precaution is made. This
can cause voltage drops at the input rail during start up and may result in an unwanted or early system shut
down.
The TPS6104x limits this inrush current by increasing the current limit in two steps starting fromfor 256
cycles tofor the next 256 cycles, and then full current limit (see Figure 15).
8.3.3 Enable
Pulling the enable (EN) to ground shuts down the device reducing the shutdown current to 1 μA (typical).
Because there is a conductive path from the input to the output through the inductor and Schottky diode, the
output voltage is equal to the input voltage during shutdown. The enable pin needs to be terminated and should
not be left floating. Using a small external transistor disconnects the input from the output during shutdown as
shown in Figure 17.
8.3.4 Undervoltage Lockout
An undervoltage lockout prevents misoperation of the device at input voltages below typical 1.5 V. When the
input voltage is below the undervoltage threshold, the main switch is turned off.
8.3.5 Thermal Shutdown
An internal thermal shutdown is implemented and turns off the internal MOSFETs when the typical junction
temperature of 168°C is exceeded. The thermal shutdown has a hysteresis of typically 25°C. This data is based
on statistical means and is not tested during the regular mass production of the IC.
8.4 Device Functional Modes
8.4.1 Operation
The TPS6104x operates with an input voltage range of 1.8 V to 6 V and can generate output voltages up to 28
V. The device operates in a pulse-frequency-modulation (PFM) scheme with constant peak current control. This
control scheme maintains high efficiency over the entire load current range, and with a switching frequency up to
1 MHz, the device enables the use of very small external components.
The converter monitors the output voltage, and as soon as the feedback voltage falls below the reference voltage
of typically 1.233 V, the internal switch turns on and the current ramps up. The switch turns off as soon as the
inductor current reaches the internally set peak current of typically 400 mA (TPS61040) or 250 mA (TPS61041).
See Peak Current Control for more information. The second criteria that turns off the switch is the maximum ontime of 6 μs (typical). This is just to limit the maximum on-time of the converter to cover for extreme conditions.
As the switch is turned off the external Schottky diode is forward biased delivering the current to the output. The
switch remains off for a minimum of 400 ns (typical), or until the feedback voltage drops below the reference
voltage again. Using this PFM peak current control scheme the converter operates in discontinuous conduction
mode (DCM) where the switching frequency depends on the output current, which results in very high efficiency
over the entire load current range. This regulation scheme is inherently stable, allowing a wider selection range
for the inductor and output capacitor.
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
The TPS6104x is designed for output voltages up to 28 V with an input voltage range of 1.8 V to 6 V and a
switch peak current limit of 400 mA (250 mA for the TPS61041). The device operates in a pulse-frequencymodulation (PFM) scheme with constant peak current control. This control scheme maintains high efficiency over
the entire load current range, and with a switching frequency up to 1 MHz, the device enables the use of very
small external components. The following section provides a step-by-step design approach for configuring the
TPS61040 as a voltage regulating boost converter for LCD bias power supply, as shown in Figure 12.
9.2 Typical Application
The following section provides a step-by-step design approach for configuring the TPS611040 as a voltage
regulating boost converter for LCD bias supply, as shown in Figure 12.
Figure 12. LCD Bias Supply
9.2.1 Design Requirements
Table 2. Design Parameters
DESIGN PARAMETEREXAMPLE VALUE
Input Voltage1.8 V to 6 V
Output Voltage18 V
Output Current10 mA
9.2.2 Detailed Design Procedure
9.2.2.1 Inductor Selection, Maximum Load Current
Because the PFM peak current control scheme is inherently stable, the inductor value does not affect the stability
of the regulator. The selection of the inductor together with the nominal load current, input and output voltage of
the application determines the switching frequency of the converter. Depending on the application, inductor
values from 2.2 μH to 47 μH are recommended. The maximum inductor value is determined by the maximum on
time of the switch, typically 6 μs. The peak current limit of 400 mA/250 mA (typically) should be reached within
this 6-μs period for proper operation.
The inductor value determines the maximum switching frequency of the converter. Therefore, select the inductor
value that ensures the maximum switching frequency at the converter maximum load current is not exceeded.
The maximum switching frequency is calculated by the following formula:
where
•IP= Peak current as described in Peak Current Control
•L = Selected inductor value
•V
= The highest switching frequency occurs at the minimum input voltage(2)
IN(min)
If the selected inductor value does not exceed the maximum switching frequency of the converter, the next step
is to calculate the switching frequency at the nominal load current using the following formula:
where
•IP= Peak current as described in Peak Current Control
•L = Selected inductor value
•I
•Vd = Rectifier diode forward voltage (typically 0.3 V)(3)
= Nominal load current
load
A smaller inductor value gives a higher converter switching frequency, but lowers the efficiency.
The inductor value has less effect on the maximum available load current and is only of secondary order. The
best way to calculate the maximum available load current under certain operating conditions is to estimate the
expected converter efficiency at the maximum load current. This number can be taken out of the efficiency
graphs shown in Figure 1 through Figure 4. The maximum load current can then be estimated as follows:
where
•IP= Peak current as described in Peak Current Control
•L = Selected inductor value
•fS
•η = Expected converter efficiency. Typically 70% to 85%(4)
= Maximum switching frequency as calculated previously
max
The maximum load current of the converter is the current at the operation point where the converter starts to
enter the continuous conduction mode. Usually the converter should always operate in discontinuous conduction
mode.
Last, the selected inductor should have a saturation current that meets the maximum peak current of the
converter (as calculated in Peak Current Control). Use the maximum value for I
for this calculation.
LIM
Another important inductor parameter is the dc resistance. The lower the dc resistance, the higher the efficiency
of the converter. See Table 3 and the typical applications for the inductor selection.
Table 3. Recommended Inductor for Typical LCD Bias Supply (see Figure 23)
Table 3. Recommended Inductor for Typical LCD Bias Supply (see
Figure 23) (continued)
DEVICEINDUCTOR VALUECOMPONENT SUPPLIERCOMMENTS
TPS6104110 μHMurata LQH3C100K24
High efficiency
Small solution size
9.2.2.2 Setting the Output Voltage
The output voltage is calculated as:
(5)
For battery-powered applications, a high-impedance voltage divider should be used with a typical value for R2 of
≤200 kΩ and a maximum value for R1 of 2.2 MΩ. Smaller values might be used to reduce the noise sensitivity of
the feedback pin.
A feedforward capacitor across the upper feedback resistor R1 is required to provide sufficient overdrive for the
error comparator. Without a feedforward capacitor, or one whose value is too small, the TPS6104x shows doublepulses or a pulse burst instead of single pulses at the switch node (SW), causing higher output voltage ripple. If
this higher output voltage ripple is acceptable, the feedforward capacitor can be left out.
The lower the switching frequency of the converter, the larger the feedforward capacitor value required. A good
starting point is to use a 10-pF feedforward capacitor. As a first estimation, the required value for the feedforward
capacitor at the operation point can also be calculated using the following formula:
where
•R1 = Upper resistor of voltage divider
•fS = Switching frequency of the converter at the nominal load current (See Inductor Selection, Maximum Load
Current for calculating the switching frequency)
•CFF= Choose a value that comes closest to the result of the calculation(6)
The larger the feedforward capacitor the worse the line regulation of the device. Therefore, when concern for line
regulation is paramount, the selected feedforward capacitor should be as small as possible. See the following
section for more information about line and load regulation.
9.2.2.3 Line and Load Regulation
The line regulation of the TPS6104x depends on the voltage ripple on the feedback pin. Usually a 50 mV peakto-peak voltage ripple on the feedback pin FB gives good results.
Some applications require a very tight line regulation and can only allow a small change in output voltage over a
certain input voltage range. If no feedforward capacitor CFFis used across the upper resistor of the voltage
feedback divider, the device has the best line regulation. Without the feedforward capacitor the output voltage
ripple is higher because the TPS6104x shows output voltage bursts instead of single pulses on the switch pin
(SW), increasing the output voltage ripple. Increasing the output capacitor value reduces the output voltage
ripple.
If a larger output capacitor value is not an option, a feedforward capacitor CFFcan be used as described in the
previous section. The use of a feedforward capacitor increases the amount of voltage ripple present on the
feedback pin (FB). The greater the voltage ripple on the feedback pin (≥50 mV), the worse the line regulation.
There are two ways to improve the line regulation further:
1. Use a smaller inductor value to increase the switching frequency which will lower the output voltage ripple,
as well as the voltage ripple on the feedback pin.
2. Add a small capacitor from the feedback pin (FB) to ground to reduce the voltage ripple on the feedback pin
down to 50 mV again. As a starting point, the same capacitor value as selected for the feedforward capacitor
CFFcan be used.
For best output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low
ESR value but tantalum capacitors can be used as well, depending on the application.
Assuming the converter does not show double pulses or pulse bursts on the switch node (SW), the output
voltage ripple can be calculated as:
where
•IP= Peak current as described in Peak Current Control
•L = Selected inductor value
•I
= Nominal load current
out
•fS (I
•Vd = Rectifier diode forward voltage (typically 0.3 V)
•C
•ESR = Output capacitor ESR value(7)
) = Switching frequency at the nominal load current as calculated previously
out
= Selected output capacitor
out
See Table 4 and the Typical Application for choosing the output capacitor.
For good input voltage filtering, low ESR ceramic capacitors are recommended. A 4.7-μF ceramic input capacitor
is sufficient for most of the applications. For better input voltage filtering this value can be increased. See Table 4
and typical applications for input capacitor recommendations.
9.2.2.6 Diode Selection
To achieve high efficiency a Schottky diode should be used. The current rating of the diode should meet the
peak current rating of the converter as it is calculated in Peak Current Control. Use the maximum value for I
for this calculation. See Table 5 and the typical applications for the selection of the Schottky diode.
Table 5. Recommended Schottky Diode for Typical LCD Bias Supply (see Figure 23)
DEVICEREVERSE VOLTAGECOMPONENT SUPPLIER
30 VON Semiconductor MBR0530
TPS6104x
20 VON Semiconductor MBR0520
20 VON Semiconductor MBRM120LHigh efficiency
30 VToshiba CRS02
The device is designed to operate from an input voltage supply range between 1.8 V and 6 V. The output current
of the input power supply must be rated according to the supply voltage, output voltage and output current of
TPS6104x.
11Layout
11.1 Layout Guidelines
Typical for all switching power supplies, the layout is an important step in the design; especially at high peak
currents and switching frequencies. If the layout is not carefully done, the regulator might show noise problems
and duty cycle jitter.
The input capacitor should be placed as close as possible to the input pin for good input voltage filtering. The
inductor and diode should be placed as close as possible to the switch pin to minimize the noise coupling into
other circuits. Because the feedback pin and network is a high-impedance circuit, the feedback network should
be routed away from the inductor. The feedback pin and feedback network should be shielded with a ground
plane or trace to minimize noise coupling into this circuit.
Wide traces should be used for connections in bold as shown in Figure 23. A star ground connection or ground
plane minimizes ground shifts and noise.
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OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
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12.2 Related Links
The table below lists quick access links. Categories include technical documents, support and community
resources, tools and software, and quick access to sample or buy.
Table 6. Related Links
PARTSPRODUCT FOLDERSAMPLE & BUY
TPS61041Click hereClick hereClick hereClick hereClick here
TPS61040Click hereClick hereClick hereClick hereClick here
12.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
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Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
TECHNICALTOOLS &SUPPORT &
DOCUMENTSSOFTWARECOMMUNITY
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E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
12.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
12.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13Mechanical, Packaging, and Orderable Information
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ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
Package Type Package
(1)
Drawing
Pins Package
Qty
Eco Plan
(2)
& no Sb/Br)
& no Sb/Br)
& no Sb/Br)
& no Sb/Br)
& no Sb/Br)
& no Sb/Br)
& no Sb/Br)
& no Sb/Br)
& no Sb/Br)
& no Sb/Br)
& no Sb/Br)
& no Sb/Br)
& no Sb/Br)
Lead/Ball Finish
(6)
MSL Peak Temp
(3)
CU NIPDAULevel-1-260C-UNLIM-40 to 85PHOI
CU NIPDAULevel-1-260C-UNLIM-40 to 85PHOI
CU NIPDAULevel-2-260C-1 YEAR-40 to 85QXK
CU NIPDAULevel-2-260C-1 YEAR-40 to 85QXK
CU NIPDAULevel-1-260C-UNLIM-40 to 85CCL
CU NIPDAULevel-1-260C-UNLIM-40 to 85CCL
CU NIPDAULevel-1-260C-UNLIM-40 to 85CCL
CU NIPDAULevel-1-260C-UNLIM-40 to 85PHPI
CU NIPDAULevel-1-260C-UNLIM-40 to 85PHPI
CU NIPDAULevel-1-260C-UNLIM-40 to 85CAW
CU NIPDAULevel-1-260C-UNLIM-40 to 85CAW
CU NIPDAULevel-1-260C-UNLIM-40 to 85CAW
CU NIPDAULevel-1-260C-UNLIM-40 to 85CAW
5-Oct-2015
Op Temp (°C)Device Marking
(4/5)
Samples
Addendum-Page 1
PACKAGE OPTION ADDENDUM
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(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
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(3)
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(4)
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(5)
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OTHER QUALIFIED VERSIONS OF TPS61040, TPS61041 :
Automotive: TPS61040-Q1, TPS61041-Q1
•
5-Oct-2015
NOTE: Qualified Version Definitions:
Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects
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