Philips 74hct9046a DATASHEETS

INTEGRATED CIRCUITS

DATA SH EET

74HCT9046A

PLL with bandgap controlled VCO

Product speciﬁcation Supersedes data of March 1994 File under Integrated Circuits, IC06

1999 Jan 11

Philips Semiconductors Product speciﬁcation

PLL with bandgap controlled VCO 74HCT9046A

FEATURES

• Low power consumption

• Centre frequency up to

17 MHz (typ.) at VCC = 5.5 V

• Choice of two phase comparators

(1)

: – EXCLUSIVE-OR (PC1) – Edge-triggered JK flip-flop (PC2)

• No dead zone of PC2

• Charge pump output on PC2,

whose current is set by an external resistor R

• Centre frequency tolerance ±10%

• Excellent

voltage-controlled-oscillator (VCO) linearity

• Low frequency drift with supply voltage and temperature variations

• On chip bandgap reference

• Glitch free operation of VCO, even

at very low frequencies

• Inhibit control for ON/OFF keying and for low standby power consumption

• Operation power supply voltage range 4.5 to 5.5 V

• Zero voltage offset due to op-amp buffering

• Output capability: standard

• ICC category: MSI.

APPLICATIONS

• Tone decoding

• Data synchronization and

conditioning

• Voltage-to-frequency conversion

• Motor-speed control.

GENERAL DESCRIPTION

The 74HCT9046A is a high-speed Si-gate CMOS device. It is specified in compliance with

no. 7A”

“JEDEC standard

QUICK REFERENCE DATA

GND = 0 V; T

= 25 °C; tr = tf≤ 6 ns.

amb

SYMBOL PARAMETER CONDITIONS TYP. UNIT

VCO centre frequency C1 = 40 pF;

16 MHz R1 = 3 kΩ; VCC= 5 V

input capacitance 3.5 pF power dissipation

notes 1 and 2 20 pF

capacitance per package

Notes

1. C

is used to determine the dynamic power dissipation (PD in µW)

a) PD = CPD× V

× fi + Σ(CL× V

× fo) where:

b) fi = input frequency in MHz; CL = output load capacity in pF;

fo = output frequency in MHz; VCC = supply voltage in V; Σ(CL× V

× fo) = sum of the outputs.

2. Applies to the phase comparator section only (inhibit = HIGH). For power dissipation of the VCO and demodulator sections see Figs 26 to 28.

ORDERING INFORMATION

EXTENDED

TYPE NUMBER

PINS PIN POSITION MATERIAL CODE

PACKAGE

74HCT9046AN 16 DIL16 plastic SOT38Z 74HCT9046AD 16 SO16 plastic SOT109A

• FM modulation and demodulation where a small centre frequency tolerance is essential

• Frequency synthesis and multiplication where a low jitter is required (e.g. Video picture-in-picture)

• Frequency discrimination

(1) Rb connected between pin 15 and

ground: PC2 mode, with PCP pin 2. Pin 15 left open or connected to V PC1 mode with PC1

OUT

at pin 2.

OUT

1999 Jan 11 2

Philips Semiconductors Product speciﬁcation

PLL with bandgap controlled VCO 74HCT9046A

PINNING

SYMBOL PIN DESCRIPTION

GND 1 ground (0 V) (phase comparators) PC1

PCP COMP VCO INH 5 inhibit input C1 C1 GND 8 ground (0 V) (VCO) VCO DEM R1 11 resistor R1 connection R2 12 resistor R2 connection PC2

SIG R V

A B

OUT

IN OUT

OUT

2 phase comparator 1 output/phase

comparator pulse output

3 comparator input 4 VCO output

6 capacitor C1 connection A 7 capacitor C1 connection B

9 VCO input

10 demodulator output

13 phase comparator 2 output

GND

PC1 /

OUT

PCP

OUT

COMP

VCO

OUT

C1 C1

GND

INH

1 2 3

9046A

5 6

MBD037 - 1

SIG

PC2

OUT

DEM

VCO

OUT IN

(current source adjustable with Rb) 14 signal input 15 bias resistor (Rb) connection

Fig.1 Pin configuration.

16 supply voltage

LOGIC/FUNCTIONAL SYMBOLS AND DIAGRAMS

PC1 /

14 15

11 12

COMP

6 7

9 5

SIG R

C1 C1

R1 R2

VCO INH

A B

VCO

PCP

PC2

VCO

DEM

MBD038 - 1

OUT OUT

OUT

11 12 15

PLL

9046A

PC1 /

COMP

6 7

9 5

SIG C1 C1

R1 R2

VCO INH

IN A B

PCP PC2

DEM VCO

MBD039 - 1

OUT OUT

OUT

OUT OUT

2 13

10 4

Fig.2 Logic symbol.

1999 Jan 11 3

Fig.3 IEC logic symbol.

Philips Semiconductors Product speciﬁcation

PLL with bandgap controlled VCO 74HCT9046A

5109

VCO

DEM

VCO

OUT

COMP

314476

SIG

9046A

PC1 /

OUT

PHASE

COMPARATOR

PHASE

COMPARATOR

VCO

OUTINH

PCP

OUT

PC2

OUT

MBD040 - 1

Fig.4 Functional diagram.

1999 Jan 11 4

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1999 Jan 11 5

Philips Semiconductors Product speciﬁcation

PLL with bandgap controlled VCO 74HCT9046A

VCO

OUT

ref1

BAND

GAP

COMP

ref2

764

DEM

OUT

ref2

VCO

ref1

314

SIG

PC1 /

OUT

PCP

OUT

PC2

OUT

logic

DQ CP

down

PC1

PCP

ref2

CHARGE

PUMP

9 VCO

INH

MBD102 - 1

Fig.5 Logic diagram.

Philips Semiconductors Product speciﬁcation

PLL with bandgap controlled VCO 74HCT9046A

FUNCTIONAL DESCRIPTION

The 74HCT9046A is a phase-locked-loop circuit that comprises a linear VCO and two different phase comparators (PC1 and PC2) with a common signal input amplifier and a common comparator input (see Fig.4). The signal input can be directly coupled to large voltage signals (CMOS level), or indirectly coupled (with a series capacitor) to small voltage signals. A self-bias input circuit keeps small voltage signals within the linear region of the input amplifiers. With a passive low-pass filter, the '9046A' forms a second-order loop PLL.

The principle of this phase-locked-loop is based on the familiar HCT4046A. However extra features are built in, allowing very high performance phase-locked-loop applications. This is done, at the expense of PC3, which is skipped in this HCT9046A. The PC2 is equipped with a current source output stage here. Further a bandgap is applied for all internal references, allowing a small centre frequency tolerance. The details are summed up in the next section called: “Differences with respect to the familiar HCT4046A”. If one is familiar with the HCT4046A already, it will do to read this section only.

DIFFERENCES WITH RESPECT TO THE FAMILIAR HCT4046A

• A centre frequency tolerance of maximum ±10%.

• The on board bandgap sets the internal references resulting in a minimal frequency shift at supply voltage variations and temperature variations.

• The value of the frequency offset is determined by an internal reference voltage of 2.5 V instead

− 0.7 V. In this way the offset

of V

frequency will not shift over the supply voltage range.

• A current switch charge pump output on PC2 allows a virtually ideal performance of PC2. The gain of PC2 is independent of the voltage across the low-pass filter. Further a passive low-pass filter in the loop achieves an active performance now. The influence of the parasitic capacitance of the PC2 output plays no role here, resulting in a true correspondence of the output correction pulse and the phase difference even up to phase differences as small as a few nanoseconds.

• Because of its linear performance without dead zone, higher impedance values for the filter, hence lower C-values, can now be chosen. Correct operation will not be influenced by parasitic capacitances as in the instance with voltage source output of the 4046A.

• No PC3 on pin 15 but instead a resistor connected to GND, which sets the load/unload currents of the charge pump (PC2).

• Extra GND pin at pin 1 to allow an excellent FM demodulator performance even at 10 MHz and higher.

• Combined function of pin 2. If pin 15 is connected to V

(no bias

resistor Rb) pin 2 has its familiar function viz. output of PC1. If at pin 15 a resistor (Rb) is connected to GND it is assumed that PC2 has been chosen as phase comparator. Connection of Rb is sensed by internal circuitry and this changes the function of pin 2 into a lock detect output (PCP same characteristics as PCP

) with the

OUT

of pin 1 of the well known 74HCT4046A.

• The inhibit function differs. For the HCT4046A a HIGH level at the inhibit input (INH) disables the VCO and demodulator, while a LOW level turns both on. For the 74HCT9046A a HIGH level on the inhibit input disables the whole circuit to minimize standby power consumption.

VCO

The VCO requires one external capacitor C1 (between C1

and C1B)

and one external resistor R1 (between R1 and GND) or two external resistors R1 and R2 (between R1 and GND, and R2 and GND). Resistor R1 and capacitor C1 determine the frequency range of the VCO. Resistor R2 enables the VCO to have a frequency offset if required (see Fig.5).

The high input impedance of the VCO simplifies the design of the low-pass filters by giving the designer a wide choice of resistor/capacitor ranges. In order not to load the low-pass filter, a demodulator output of the VCO input voltage is provided at pin 10 (DEM

). The DEM

OUT

voltage equals that of the VCO input. If DEM

is used, a load resistor (Rs)

OUT

should be connected from pin 10 to GND; if unused, DEM left open. The VCO output (VCO

should be

OUT

can be connected directly to the comparator input (COMPIN), or connected via a frequency-divider. The VCO output signal has a duty factor of 50% (maximum expected deviation 1%), if the VCO input is held at a constant DC level. A LOW level at the inhibit input (INH) enables the VCO and demodulator, while a HIGH level turns both off to minimize standby power consumption.

)

1999 Jan 11 6

Philips Semiconductors Product speciﬁcation

PLL with bandgap controlled VCO 74HCT9046A

Phase comparators

The signal input (SIGIN) can be directly coupled to the self-biasing amplifier at pin 14, provided that the signal swing is between the standard HC family input logic levels. Capacitive coupling is required for signals with smaller swings.

HASE COMPARATOR 1 (PC1)

This circuit is an EXCLUSIVE-OR network. The signal and comparator input frequencies (f

) must have a

50% duty factor to obtain the maximum locking range. The transfer characteristic of PC1, assuming ripple (f

= 2fi) is suppressed, is:

DEMOUT

---------- -

–()=

SIGINΦCOMPIN

where:

DEMOUT

is the demodulator output

at pin 10. V

DEMOUT

= V

PC1OUT

(via low-pass).

The phase comparator gain is:

Vr⁄()=

---------- -

The average output voltage from PC1, fed to the VCO input via the low-pass filter and seen at the demodulator output at pin 10 (V

DEMOUT

), is the resultant of the phase differences of signals (SIGIN) and the comparator input (COMPIN) as shown in Fig.6. The average of V

DEMOUT

there is no signal or noise at SIG

is equal to1⁄2VCC when

and with this input the VCO oscillates at the centre frequency (fc). Typical waveforms for the PC1 loop locked at fc are shown in Fig.7. This figure also shows the actual waveforms across the VCO capacitor at pins 6 and 7 (V

C1A

and V

) to show the relation

C1B

between these ramps and the VCO

OUT

voltage.

The frequency capture range (2f

) is defined as the frequency range of input signals on which the PLL will lock if it was initially out-of-lock. The frequency lock range (2fL) is defined as the frequency range of the input signals on which the loop will stay locked if it was initially in lock. The capture range is smaller or equal to the lock range.

With PC1, the capture range depends on the low-pass filter characteristics and can be made as large as the lock range. This configuration remains locked even with very noisy input signals. Typical behaviour of this type of phase comparator is that it may lock to input frequencies close to the harmonics of the VCO centre frequency.

HASE COMPARATOR 2 (PC2)

This is a positive edge-triggered phase and frequency detector. When the PLL is using this comparator, the loop is controlled by positive signal transitions and the duty factors of SIGIN and COMPIN are not important. PC2 comprises two D-type flip-flops, control gating and a 3-state output stage with sink and source transistors acting as current sources, henceforth called charge pump output of PC2. The circuit functions as an up-down counter (Fig.5) where SIGIN causes an up-count and COMPIN a down count. The current switch charge pump output allows a virtually ideal performance of PC2, due to appliance of some pulse overlap of the up and down signals. See Fig.8a.

1999 Jan 11 7

Philips Semiconductors Product speciﬁcation

PLL with bandgap controlled VCO 74HCT9046A

DEMOUTVPC1OUT

PCIN

SIGINΦCOMPIN

Fig.6 Phase comparator 1; average output voltage as a function of input phase difference.

–()=

---------- -

DEMOUT(AV)

–()==

SIGINΦCOMPIN

1/2V

MBD101 - 1

PCIN

180

SIGN

COMP

VCO

OUT

PC1

OUT

VCO

C1A

C1B

Fig.7 Typical waveforms for PLL using phase comparator 1; loop-locked at fc.

1999 Jan 11 8

GND

pin 6

pin 7

MBD100

Philips Semiconductors Product speciﬁcation

PLL with bandgap controlled VCO 74HCT9046A

The pump current IP is independent from the supply voltage and is set by the internal bandgap reference of

2.5 V.

2.5

-------R

A()×=

is the external bias resistor

between pin 15 and ground. The current and voltage transfer

function of PC2 are shown in Fig.9. The phase comparator gain is:

------2π

Ar⁄()=

Typical waveforms for the PC2 loop locked at f

are shown in Fig.10.

When the frequencies of SIGIN and COMPIN are equal but the phase of SIGIN leads that of COMPIN, the up output driver at PC2

is held ‘ON’

OUT

for a time corresponding to the phase difference (Φ

). When the phase of

PCIN

SIGIN lags that of COMPIN, the down or sink driver is held ‘ON’.

When the frequency of SIGIN is higher than that of COMPIN, the source output driver is held ‘ON’ for most of the input signal cycle time and for the remainder of the cycle time both drivers are ‘OFF’ (3-state). If the SIGIN frequency is lower than the COMPIN frequency, then it is the sink driver that is held ‘ON’ for most of the cycle. Subsequently the voltage at the capacitor (C2) of the low-pass filter connected to PC2

varies until the

OUT

signal and comparator inputs are equal in both phase and frequency. At this stable point the voltage on C2 remains constant as the PC2 output is in 3-state and the VCO input at pin 9 is a high impedance. Also in this condition the signal at the phase comparator pulse output (PCP

OUT

) has a minimum output pulse width equal to the overlap time, so can be used for indicating a locked condition.

Thus for PC2 no phase difference exists between SIGIN and COMP

over the full frequency range of the VCO. Moreover, the power dissipation due to the low-pass filter is reduced because both output drivers are OFF for most of the signal input cycle. It should be noted that the PLL lock range for this type of phase comparator is equal to the capture range and is independent of the low-pass filter. With no signal present at SIGIN the VCO adjust, via PC2, to its lowest frequency.

By using current sources as charge pump output on PC2, the dead zone or backlash time could be reduced to zero. Also, the pulse widening due to the parasitic output capacitance plays no role here. This enables a linear transfer function, even in the vicinity of the zero crossing. The differences between a voltage switch charge pump and a current switch charge pump are shown in Fig.11.

The design of the low-pass filter is somewhat different when using current sources. The external resistor R3 is no longer present when using PC2 as phase comparator. The current source is set by Rb. A simple capacitor behaves as an ideal integrator now, because the capacitor is charged by a constant current. The transfer function of the voltage switch charge pump may be used. In fact it is even more valid, because the transfer function is no longer restricted for small changes only. Further the current is independent from both the supply voltage and the voltage across the filter. For one that is familiar with the low-pass filter design of the 4046A a relation may show how R

relates with a fictive series resistance, called R3'.

This relation can be derived by assuming first that a voltage controlled switch PC2 of the 4046A is

connected to the filter capacitance C2 via this fictive R3' (see Fig.8b). Then during the PC2 output pulse the charge current equals:

–

-----------------------------------

CCVC2 0()

R3'

at:

C2(0)

2.5

-------- R3'

With the initial voltage V

⁄2VCC = 2.5 V,

As shown before the charge current of the current switch of the 9046A is:

2.5

×=

------- R

Hence:

------ 17

Ω()=

R3'

Using this equivalent resistance R3' for the filter design the voltage can now be expressed as a transfer function of PC2; assuming ripple (f

) is suppressed, as:

r=fi

PC2

------ 4π

Vr⁄()=

Again this illustrates the supply voltage independent behaviour of PC2.

Examples of PC2 combined with a passive filter are shown in Figs 12 and 13. Figure 12 shows that PC2 with only a C2 filter behaves as a high-gain filter. For stability the damped version of Fig.13 with series resistance R4 is preferred.

Practical design values for R

are

between 25 and 250 kΩ with R3' = 1.5 to 15 kΩ for the filter design. Higher values for R3' require lower values for the filter capacitance which is very advantageous at low values the loop natural frequency ω

1999 Jan 11 9

Philips Semiconductors Product speciﬁcation

PLL with bandgap controlled VCO 74HCT9046A

PC2

∆ Φ = Φ

PCIN

pulse overlap of

approximately 15 ns

down

OUT

MBD046 - 1

R3'

down C2

PC2 VC2

OUT OUT

MBD099

a. At every ∆Φ, even at zero∆Φ both switches are closed simultaneously for a short period (typically 15 ns). b. Comparable voltage-controlled switch.

Fig.8 The current switch charge pump output of PC2.

MSB306 - 1

PCIN

DEMOUT(AV)

1/2V

b.a.

I x R

PCIN

SIGINΦCOMPIN

Two kinds of transfer functions may be regarded:

PCIN

------2π

PCIN

by connecting a resistor (R = 10 kΩ) between PC2

OUT

a. The current transfer:

b. The voltage transfer; this transfer can be observed at PC2

DEMOUTVPC2OUT

pump current

------ 4π

Fig.9 Phase comparator 2.

1999 Jan 11 10

and1⁄2VCC;

OUT

Philips Semiconductors Product speciﬁcation

PLL with bandgap controlled VCO 74HCT9046A

SIG

COMP

VCO

OUT

OPC

DOWN

CURRENT AT PC2

OUT

high impedance OFF state, (zero current)

PC2 /VCO

OUT IN

PCP

OUT

The pulse overlap of the up and down signals (typically 15 ns).

Fig.10 Timing diagram for PC2.

2.75

VCO

2.50

2.25

(1)

(2)

025

phase error (ns)

VCO

2.75

2.50

2.25

MBD047 - 1

025

phase error (ns)

a. Response with traditional voltage-switch charge-pump PC2 (1) Due to parasitic capacitance on PC2 (2) Backlash time (dead zone).

b. Response with current switch charge-pump PC2

OUT

as applied in the HCT9046A.

OUT

(4046A).

Fig.11 The response of a locked-loop in the vicinity of the zero crossing of the phase error.

1999 Jan 11 11

b.a.

MBD043

Philips Semiconductors Product speciﬁcation

PLL with bandgap controlled VCO 74HCT9046A

LOOP FILTER COMPONENT SELECTION

ωj()

1/OUTPUTINPUT

b. Amplitude characteristic:

c. Pole zero diagram.

C2× R3' C2×==

------ 17

a. b. c.

---------------------------- -

jω()

1A⁄ jωτ1+

≈=

----------j ωτ

Fig.12 Simple loop filter for PC2 without damping.

ωj()

OUTPUTINPUT

MBD045 - 1

a. b. c.

b. Amplitude characteristic:

c. Pole zero diagram. A = DC gain limit, due to leakage.

C2× R3' C2×==

------ 17

R4 C2×=

jω()

1jωτ2+

---------------------------- 1A⁄ jωτ

Fig.13 Simple loop filter for PC2 with damping.

1999 Jan 11 12

1 /

MBD044 - 1

+ 28 hidden pages