The TDA8752B is a triple 8-bit ADC with controllable amplifiers and clamps for the
digitizing of large bandwidth RGB signals.
The clamp level, the gain and all other settings are controlled via a serial interface
(either I2C-bus or 3-wire serial bus, selected via a logic input).
The IC also includes a PLL that can be locked to the horizontal line frequency and
generates the ADC clock. The PLL jitter is minimized for high resolution PC graphics
applications. An external clock can also be input to the ADC.
It is possible to set the TDA8752B serial bus address to four different values, when
several TDA8752B ICs are used in a system, by means of the I2C-bus interface (for
example, two ICs used in an odd/even configuration).
2.Features
■ Triple 8-bit ADC
■ Sampling rate up to 110 Msps
c
c
■ IC controllable via a serial interface, which can be either I2C-bus or 3-wire serial
bus, selected via a TTL input pin
■ IC analog voltageinputfrom0.4 to 1.2 V (p-p) to produce a full-scale ADC input of
1 V (p-p)
■ Three clamps for programming a clamping code between −63.5 and +64 in steps
of1⁄2LSB for RGB signals, and from +120 to +136 in steps of1⁄2LSB for YUV
signals
■ Three controllable amplifiers: gain controlled via the serial interface to produce a
full-scale resolution of1⁄2LSB peak-to-peak
■ Amplifier bandwidth of 250 MHz
■ Low gain variation with temperature
■ PLL controllable via the serial interface to generate the ADC clock which can be
■ Possibility to use up to four ICs in the same system when using the I2C-bus
interface, or more when using the 3-wire serial bus interface
■ 1.1 W power dissipation.
3.Applications
■ RGB high-speed digitizing
■ LCD panels drive
■ LCD projection systems
■ VGA and higher resolutions
■ Using two ICs in parallel, a higher display resolution can be obtained: 200 MHz
pixel frequency.
4.Quick reference data
Table 1:Quick reference data
SymbolParameterConditionsMinTypMaxUnit
V
CCA
V
DDD
V
CCD
V
CCO
V
CCA(PLL)
V
CCO(PLL)
I
CCA
I
DDD
I
CCD
I
CCO
I
CCA(PLL)
I
CCO(PLL)
f
clk
f
ref(PLL)
f
VCO
INLDC integral non-linearityfrom analog input to digital
DNLDC differential non-linearityfrom analog input to digital
∆G
amp
Bamplifier bandwidth−3 dB; T
t
set
9397 750 07338
Product specificationRev. 03 — 21 July 20002 of 38
analog supply voltagefor R, G and B channels4.755.05.25V
logic supply voltagefor I2C-bus and 3-wire
4.755.05.25V
serial bus
digital supply voltage4.755.05.25V
output stages supply voltagefor R, G and B channels4.755.05.25V
analog PLL supply voltage4.755.05.25V
output PLL supply voltage4.755.05.25V
analog supply current−120−mA
logic supply currentfor I2C-bus and 3-wire−1.0−mA
digital supply current−40−mA
output stages supply currentf
40logic I2C-bus/3-wire serial bus digital power supply
41logic I2C-bus/3-wire serial bus digital ground
2
C-bus/3-wire serial bus clock input
n.c.43not connected
n.c.44not connected
ROR45red channel ADC output bit out of range
GOR46green channel ADC output bit out of range
BOR47blue channel ADC output bit out of range
OGND
This triple high-speed 8-bit ADC is designed to convert RGB signals, coming from an
analog source, into digital data used by a LCD driver (pixel clockupto 200 MHz when
using 2 ICs).
8.1 IC analog video inputs
The video inputs are internally DC polarized. These inputs are AC coupled externally.
8.2 Clamps
Three independent parallel clamping circuits are used to clamp the video input
signals on the black level and to control the brightness level. The clamping code is
programmable between code −63.5 and +64 and from +120 to +136 in steps of
1
⁄2LSB. The programming of the clamp value is achieved via an 8-bit DAC. Each
clamp must be able to correct an offset from ±0.1 V to ±10 mV within 300 ns, and
correct the total offset in 10 lines.
The clamps are controlled by an external TTL positive going pulse (pin CLP). The
drop of the video signal is <1 LSB.
Normally, the circuit operates with a 0 code clamp, corresponding to the 0 ADC code.
This clamp code can be changed from −63.5 to +64 as represented in Figure 5, in
steps of1⁄2LSB. The digitizedvideo signal is alwaysbetween code 0 and code 255 of
the ADC. It is also possible to clamp from code 120 to code 136 corresponding to
120 ADC code to 136 ADC code. Then clamping on code 128 of the ADC is possible.
255
digitized
video signal
CLP
video
signal
code 64
code 0
−63.5
code
clamp
programming
FCE471
= 120 to 136
Three independent variable gain amplifiers are used to provide, to each channel, a
full-scale input range signal to the 8-bit ADC. The gain adjustment range is designed
so that for an input range varying from 0.4 to 1.2 V (p-p), the output signal
corresponds to the ADC full-scale input of 1 V (p-p).
To ensure that the gain does not vary over the whole operating temperature range, an
external supplied reference voltage V
= 2.5 V (DC), with a maximum variation of
ref
100 ppm/°C, is used to calibrate the gain at the beginning of each video line before
the clamp pulse.
9397 750 07338
Product specificationRev. 03 — 21 July 200011 of 38
) is
generated internally. During the synchronization part of the video line, the multiplexer,
controlled by the TTL synchronization signal (HSYNCI, coming from HSYNC;
see Figure 1) with a width equal to one of the video synchronization signals (e.g. the
signal coming from a synchronization separator), is switched between the two
amplifiers.
The output of the multiplexer is either the normal video signal or the 0.156 V
reference signal (during HSYNC).
The corresponding ADC outputs are then compared to a preset value loaded in a
register. Depending on the result of the comparison, the gain of the variable gain
amplifiers is adjusted (coarse gain control; see Figure 2 and 6). The three 7-bit
registers receive data via a serial interface to enable the gain to be programmed.
The preset value loaded in the 7-bit register is chosen between approximately
67 codes to ensure the full-scale input range (see Figure 6). A contrast control can be
achieved using these registers. In this case care should be taken to stay within the
allowed code range (32 to 99).
A fine correction using three 5-bit DACs, also controlled via the serial interface, is
used to fine tune the gain of the three channels (fine gain control; see Figure 2 and 7)
and to compensate the channel-to-channel gain mismatch.
With a full-scale ADC input, the resolution of the fine register corresponds to1⁄2LSB
peak-to-peak variation.
Fig 6. Coarse gain control.
To use these gain controls correctly,it is recommended to fix the coarse gain (to have
a full-scale ADC input signal) to within 4 LSB and then adjust it with the fine gain. The
gain is adjusted during HSYNC. During this time the output signal is not related to the
amplified input signal. The outputs, when the coarse gain system is stable, are
related to the programmed coarse code (see Figure 6).
0.6
ADC output
code
255
227
160
128
V
i (p-p)
2
FCE472
N
COARSE
coarse
register
value
(67 codes)
code
127
99
32
0
0.156 =
G
(max)
G
(min)
V
0.2
ref
16
9397 750 07338
Product specificationRev. 03 — 21 July 200012 of 38
The ADCs are 8-bit with a maximum clock frequency of 110 Msps. The ADCs input
range is 1 V (p-p) full-scale. One out of range bit exists per channel (ROR, GOR and
BOR). It will be at logic 1 when the signal is out of range of the full-scale of the ADCs.
Pipeline delay in the ADCs is 1 clock cycle from sampling to data output.
The ADCs reference ladders regulators are integrated.
8.5 ADC outputs
ADC outputs are straight binary. An output enable pin (OE; active LOW) enables the
output status between active and high-impedance (OE = HIGH) to be switched; it is
recommended to load the outputs with a 10 pF capacitive load. The timing must be
checked very carefully if the capacitive load is more than 10 pF.
8.6 Phase-locked loop
NCOARSE
N
FINE
.
N
FINE
= 31
= 0
V
ref
FCE473
The ADCs are clocked either by an internal PLL locked to the CKREF clock (all of the
PLL is on-chip except the loop filter capacitance) or by an external clock applied to
pin CKEXT. Selection is performed via the serial interface bus.
The reference clock (CKREF) range is between 15 and 280 kHz. Consequently, the
VCO minimum frequency is 12 MHz and the maximum frequency is 110 MHz. The
gain of the VCO part can be controlled via the serial interface, depending on the
frequency range to which the PLL is locked.
To increase the bandwidth of the PLL, the charge pump current, controlled by the
serial interface, must also be increased. The relationship between the frequency and
the current is given by the following equation:
1
f
------
n
2π
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Product specificationRev. 03 — 21 July 200013 of 38
fz= loop filter zero frequency
R = the chosen resistance for the filter
ξ = the damping factor
FO= 0 dB loop gain frequency.
Different resistances for the filter can be programmed via the serial interface. To
improve the performances, the PLL parameters should be chosen so that:
2ξ fn⋅RIP⋅⇒
==
O
F
O
0.15R I
--------- -
f
ref
2πDR×FO×
---------------------------------- -
K
O
0.3πDR×f
≤⋅⇒≤Lim=
----------------------------------------- -
P
×
K
O
ref
(3)
(4)
The values of R and IP must be chosen so that the product is the closest to Lim. In
the eventthat there are several choices, the couple forwhich the ξ value is the closest
to 1 must be chosen.
A software program called “PLL calculator’” is available on Philips Semiconductor
Internet site to calculate the best PLL parameters.
It is possible to control (independently) the phase of the ADC clock and the phase of
an additional clock output (which could be used to drive a second TDA8752B). For
this, two serial interface-controlled digital phase-shift controllers are included
(controlled by 5-bit registers, phase-shift controller steps are 11.25 deg each on the
whole PLL frequency range).
CKREF is re-synchronized, by the synchro block, on the CKAO clock. The output is
CKREFO (LOW during 8 clock periods). CKAO is the clock at the output of the phase
selector A. This clock can be used as the clocks for CKBO and CKADCO. The timing
is given in Figure 8.
Pin COAST is used to disconnect the PLL phase frequency detector during the frame
flyback or the unavailability of the CKREF signal. This signal can normally be derived
from the VSYNC signal.
The clock output is able to drive an external 10 pF load (for the on-chip ADCs).
9397 750 07338
Product specificationRev. 03 — 21 July 200014 of 38
The IC can be used as stand-alone with a sampling frequency of up to 110 MHz.
When an RGB signal is at a pixel frequency exceeding 100 to 200 MHz, it is
possible to follow one of the two possibilities given below:
– Using one TDA8752B: the sampling rate can be reduced by a factor of two, by
sampling the even pixels in the even frame and the odd pixels in the odd frame.
Pin INV is used to toggle between the frames.
– Using two TDA8752Bs: the PLL of the master TDA8752B is used to drive both
ADC clocks. The PLL of the slave TDA8752B is disconnected and the CKBO of
the master TDA8752Bis connected to pin CKEXT of the TDA8752B master and
CKAO to the slave TDA8752B. In this case, on pin CKAO CKBO will be the
output (with bit CKAB of the master at logic 1).
The master TDA8752B is used to sample the even pixels and the slave
TDA8752B forodd pixels,using a 180 deg phase shift between the clocks (both
pins CKADCO). The master chip and the slave chip have their pin INV LOW,
which guarantees the 180 deg shift ADC clock drive. It is then necessary to
adjust phase B of the master chip. Special care should be taken with the quality
of the input signal (input settling time).
If CKREFO output signal at the master chip is needed, it is possible to use one
of the two phase A values in order to avoid set-up and hold problems in the
SYNCHRO function; e.g. PHASEA = 100000 and PHASEA = 111111.
When INV is LOW, CKADCO is equal to CKEXT inverted.
CKREF
CKAO
CKREFO
t
CKAO=tCLK(buffer)+tphase selector
t
= either t
CKREFO
CKAO
Fig 8. Timing diagram.
t
CKAO
t
CKREFO
[t
CLK(buffer)
T
CLK(pixel)
−if PHASEA ≥ 01000 or t
------------------------
2
= 10 ns and t
phase selector
t
phase selector
=].
-------------------------- -
2π
T
CLK(pixel)
+ if PHASEA < 01000.
------------------------
CKAO
2
T
×
FCE470
CLK(pixel)
9397 750 07338
Product specificationRev. 03 — 21 July 200015 of 38
Slave at 180 deg phase shift with respect to pin CKADCO of the master TDA8752B.
Fig 9. Dual TDA8752B solution for pixel clock rate with a single phase adjustment (100 to 200 MHz).
8.7 I2C-bus and 3-wire serial bus interface
The I2C-bus and 3-wire serial buses control the status of the different control DACs
and registers. Control pin DIS enables or disables the full serial interface function
(disable at HIGH level). Four ICs can be used in the same system and programmed
by the same bus. Therefore, two pins (ADD1 and ADD2) are available to set each
address respectively, for use with the I2C-bus interface. All programming is described
in Section 9 “I2C-bus and 3-wire serial bus interfaces”.
All the registers are defined by a subaddress of 8 bits; bit A4 refers to the mode which
is used with the I2C-bus interface; bits Sa3 to Sa0 are the subaddresses of each
register.
Bit Mode, used only with the I2C-bus, enables two modes to be programmed:
Mode 0if bit Mode = 0, each register is programmed independently by giving its
subaddress and its content
Mode 1if bit Mode = 1, all the registers are programmed one after the other by
giving this initial condition (XXX1 1111) as the subaddress state; thus,
the registers are charged following the predefined sequence of 16 bytes
(from subaddress 0000 to 1101).
9.1.1 Offset register
This register controls the clamp levelforthe RGB channels. The relationship between
the programming code and the level of the clamp code is given in Table 5.
9397 750 07338
Product specificationRev. 03 — 21 July 200017 of 38
These two registers enable the gain control, the AGC gain with the coarse register
and the reference voltage with the fine register. The coarse register programming
equation is as follows:
GAIN
Where: V
N
COARSE
----------------------------------------------- -
V
1
⋅
ref
= 2.5 V.
ref
N
–
----------------- -
32 16×
1+
FINE
The gain correspondence is given in Table 6. The gain is linear with reference to the
programming code (N
To modulate this gain, the fine register is programmed using the above equation. With
a full-scale ADC input, the fine register resolution is a1⁄2LSB peak-to-peak
(see Table 7 for N
9397 750 07338
Product specificationRev. 03 — 21 July 200018 of 38
COAST and HSYNC signals can be inverted by setting the I2C-bus control bits
‘Vlevel’ and ‘Hlevel’ respectively. When ‘Vlevel’ and ‘Hlevel’ are set to zero
respectively, COAST and HSYNC are active HIGH.
The bit ‘Edge’ defines the rising or falling edge of CKREF to synchronize the PLL. It
will be on the rising edge if the bit is at logic 0 and on the falling edge if the bit is at
logic 1.
The bits ‘Up’ and ‘Do’ are used for the test, to force the charge pump current. These
bits have to be logic 0 during normal use.
The bits ‘Ip0’, ‘Ip1’ and ‘Ip2’ control the charge pump current, to increase the
bandwidth of the PLL, as shown in Table 8.
This register controls the PLL frequency. The bits are the LSB bits.
The default programmed value is 0011 0010 0000 = 800.
The MSB bits (Di11, Di10 and Di9) and the LSB bit (Di0) have to be programmed
before bits ‘Di8’ to ‘Di1’ are programmed, to obtain the required divider ratio. Bit ‘Di0’
is used for the parity divider number (bit ‘Di0’ = 0 means even number, while
bit ‘Di0’ = 1 means odd number). It should be noted that if the I2C-bus programmingis
done in mode 1 (bit Mode = 1) and bit ‘Di0’ has to be toggled, then the registers have
to be loaded twice to have the update divider ratio.
When the supply is completely switched off, the registers are set to their default
values; in that event they have to be reprogrammed if the required settings are
different (e.g. through an EEPROM)
When the device is in Power-down mode, the previously programmed register
values remain unaffected.
9.1.7 PHASEA and PHASEB registers
Bit ‘Cka’ is logic 0 when the used clock is the PLL clock, and logic 1 when the used
clock is the external clock.
Bit ‘Ckb’ is logic 0 when the second clock is not used.
Bits ‘Pa4’ to ‘Pa0’ and bits ‘Pb4’ to ‘Pb0’ are used to program the phase shift for the
clock, CKADCO,CKAO and CKBO (see Table 11). Concerning the PHASEB register,
bit ‘Ckab’ is used to have either CKAO or CKBO at pin CKAO (pin 81).
Table 11: Phase registers bits
Pa4 and Pb4 Pa3 and Pb3 Pa2 and Pb2 Pa1 and Pb1 Pa0 and Pb0 Phase shift (deg)
000000
0000111.25
..................
11110337.5
11111348.75
The default programmed value is as follows:
No external clock: bit ‘Cka’ is logic 0
•
No use of the second clock: bit ‘Ckb’ is logic 0
•
Phase shift for CKAO and CKADCO is 0 deg
•
Phase shift for CKBO is 0 deg
•
Clock CKAO at pin CKAO: bit ‘Ckab’ is logic 0.
•
9397 750 07338
Product specificationRev. 03 — 21 July 200021 of 38
The I2C-bus address of the circuit is 1001 1xx0.
Bits ‘A2’ and ‘A1’ are fixed by the potential on pins ADD1 and ADD2. Thus, four
TDA8752Bs can be used on the same system, using the addresses for
ADD1 and ADD2 with the I2C-bus. Bit ‘A0’ must always be equal to logic 0 because it
is not possible to read the data in the register. The timing and protocol for the I2C-bus
are standard. Two sequences are available, see Table 13 and 14.
Table 13: Address sequence for mode 0
Where: S = START condition, ACK = acknowledge and P = STOP condition.
SIC ADDRESSACKSUBADDRESS
REGISTER1
ACKDATA
REGISTER1
(see Table 4)
ACKSUBADDRESS
REGISTER2
TDA8752B
ACK...P
Table 14: Address sequence for mode 1
Where: S = START condition, ACK = acknowledge and P = STOP condition.
SIC ADDRESSACKSUBADDRESS
XXX1 1111
ACKDATA
REGISTER1
(see Table 4)
ACKDATA
REGISTER2
ACK...P
9397 750 07338
Product specificationRev. 03 — 21 July 200022 of 38
Product specificationRev. 03 — 21 July 200023 of 38
9.3 3-wire serial bus protocol
For the 3-wire serial bus the first byte refers to the register address which is programmed. The second byte refers to the
data to be sent to the chosen register (see Table 4). The acquisition is achieved via SEN.
Using the 3-wire serial bus interface, an indefinite number of ICs can operate on the same system. Pin SEN is used to
validate the circuits.
LOW-level input voltage−−0.8V
HIGH-level input voltage2.0−−V
input current pin OE with 10 kΩ resistor
input current pin PWDWN with 10 kΩ resistor
[3]
−−1.0mA
[3]
−−1.0mA
3-wire serial bus
t
rst
reset time of the chip before
−600−ns
3-wire serial bus
communication
t
su
t
h
2
C-bus
I
f
SCL
t
BUF
data set-up time−100−ns
data hold time−100−ns
[4]
clock frequency0−100kHz
time the bus must be free
4.7−−µs
before new transmission
can start
t
HD;STA
t
SU;STA
t
CKL
t
CKH
t
SU;DAT
t
HD;DAT
t
r
t
f
t
SU;STOP
C
L(bus)
start condition hold time4.0−−µs
start condition set-up timerepeated start4.7−−µs
LOW-level clock period4.7−−µs
HIGH-level clock period4.0−−µs
data set-up time250−−ns
data hold time0−−ns
SDA and SCL rise timef
SDA and SCL fall timef
= 100 kHz−−1.0µs
SCL
= 100 kHz−−300ns
SCL
stop condition set-up time4.0−−µs
bus line capacitive loading−−400pF
9397 750 07338
Product specificationRev. 03 — 21 July 200028 of 38
[1] Effective bits are obtained via a Fast Fourier Transform (FFT) treatment taking 8000 acquisition points per equivalent fundamental
period. The calculation takes into account all harmonics and noise up to half clock frequency (NYQUIST frequency).
Conversion-to-noise ratio: S/N = EB × 6.02 + 1.76 dB.
[2] Output data acquisition is available after the maximum delay time t
timings are given for a 10 pF capacitive load. A higher load can be used but the timing must then be rechecked.
[3] The input current must be limited in accordance with the limiting values.
[4] The I2C-bus timings are given for a frequency of 100kbit/s (100 kHz). This bus can be used at a frequency of 400 kbit/s (400 kHz).
, which is the time during which the data is available. All the
d(o)
CKADCO
DATA
R0 to R7, ROR
G0 to G7, GOR
B0 to B7, BOR
V
in
Fig 11. Data timing diagram.
handbook, full pagewidth
OE
output
data
output
data
V
CCD
t
dLZ
LOW
10%
n
I
n − 1
t
CPH
t
d(s)
sample N
HIGH
t
dZL
I
n
sample N + 1
t
dHZ
HIGH
50%
t
CPL
t
d(o)
I
n + 1
t
sample N + 2
90%
h(o)
LOW
TDA8752B
I
n + 2
50%
t
dZH
50 % = 1.4 V
2.4 V
1.4 V
0.4 V
FCE475
50%
3.3 kΩ
S1
V
CCD
10 pF
FCE476
fOE= 100 kHz; switch S1 connected to V
CCD
for t
dLZ
and t
OE
; switch S1 connected to GND for t
dZL
dHZ
and t
dZH
.
Fig 12. Timing diagram and test conditions of 3-state output delay time.
9397 750 07338
Product specificationRev. 03 — 21 July 200029 of 38
All supplypins haveto be decoupled,with two capacitors: one forthe high frequencies(approximately 1 nF) and one forthe low
frequencies (approximately 100 nF orhigher). If a capacitor of39 nF betweenpins CZ and CPis notavailable, use a higherone
as close as possible to this value. Resistors R1 and R2 must be connected: the recommended value is 10 kΩ.
Fig 13. Application diagram.
9397 750 07338
Product specificationRev. 03 — 21 July 200031 of 38
Inputs and outputs are protected against electrostatic discharge in normal handling.
However, to be completely safe, it is desirable to take normal precautions appropriate
to handling integrated circuits.
16. Soldering
16.1 Introduction to soldering surface mount packages
This text gives a very brief insight to a complex technology. A more in-depth account
of soldering ICs can be found in our
Packages
There is no soldering method that is ideal for all surface mount IC packages. Wave
soldering can still be used for certain surface mount ICs, but it is not suitable for fine
pitch SMDs. In these situations reflow soldering is recommended.
Reflow soldering requires solder paste (a suspension of fine solder particles, flux and
binding agent) to be applied to the printed-circuit board by screen printing, stencilling
or pressure-syringe dispensing before package placement.
Several methods exist for reflowing; for example, convection or convection/infrared
heating in a conveyor type oven. Throughput times (preheating, soldering and
cooling) vary between 100 and 200 seconds depending on heating method.
Typical reflow peak temperatures range from 215 to 250 °C. The top-surface
temperature of the packages should preferable be kept below 220 °C for thick/large
packages, and below 235 °C small/thin packages.
16.3 Wave soldering
Conventional single wave soldering is not recommended for surface mount devices
(SMDs) or printed-circuit boards with a high component density, as solder bridging
and non-wetting can present major problems.
To overcome these problems the double-wave soldering method was specifically
developed.
If wave soldering is used the following conditions must be observed for optimal
results:
Use a double-wave soldering method comprising a turbulent wave with high
•
upward pressure followed by a smooth laminar wave.
For packages with leads on two sides and a pitch (e):
•
– larger than or equal to 1.27 mm, the footprint longitudinal axis is preferred to be
parallel to the transport direction of the printed-circuit board;
– smaller than 1.27 mm, the footprint longitudinal axis must be parallel to the
transport direction of the printed-circuit board.
9397 750 07338
Product specificationRev. 03 — 21 July 200033 of 38
During placement and before soldering, the package must be fixed with a droplet of
adhesive. The adhesive can be applied by screen printing, pin transfer or syringe
dispensing. The package can be soldered after the adhesive is cured.
Typical dwell time is 4 seconds at 250 °C. A mildly-activated flux will eliminate the
need for removal of corrosive residues in most applications.
16.4 Manual soldering
Fix the component by first soldering two diagonally-opposite end leads. Use a low
voltage (24 V or less) soldering iron applied to the flat part of the lead. Contact time
must be limited to 10 seconds at up to 300 °C.
When using a dedicated tool, all other leads can be soldered in one operation within
2 to 5 seconds between 270 and 320 °C.
The footprint must incorporate solder thieves at the downstream end.
For packages with leads on four sides, the footprint must be placed at a 45° angle
to the transport direction of the printed-circuit board. The footprint must
incorporate solder thieves downstream and at the side corners.
16.5 Package related soldering information
Table 19: Suitability of surface mount IC packages for wave and reflow soldering
[1] All surface mount (SMD)packages are moisture sensitive. Depending upon the moisture content, the
[2] These packages arenot suitable for wavesoldering as a solder joint between the printed-circuit board
[3] If wave soldering is considered, then the package must be placed at a 45° angle to the solder wave
[4] Wave soldering is only suitable for LQFP, QFP and TQFP packages with a pitch (e) equal to or larger
[5] Wave soldering is onlysuitable forSSOP andTSSOP packages with a pitch(e) equal to or largerthan
, SO, SOJsuitablesuitable
maximum temperature(with respect totime) and bodysize of thepackage, there is a risk that internal
or external package cracks may occur due to vaporization of the moisture in them (the so called
popcorn effect). For details, refer to the Drypack information in the
Circuit Packages; Section: Packing Methods
and heatsink (atbottom version) can not be achieved, and as solder may stickto the heatsink (on top
version).
direction. The package footprint must incorporate solder thieves downstream and at the side corners.
than 0.8 mm; it is definitelynot suitable for packages with a pitch (e)equal to orsmaller than 0.65 mm.
0.65 mm; it is definitely not suitable for packages with a pitch (e) equal to or smaller than 0.5 mm.
not suitable
.
[2]
[3][4]
[5]
Data Handbook IC26; Integrated
suitable
suitable
suitable
[1]
9397 750 07338
Product specificationRev. 03 — 21 July 200034 of 38
Objective specificationDevelopmentThis data sheet contains the design target or goal specifications for product development. Specification may
change in any manner without notice.
Preliminary specification QualificationThis data sheet contains preliminary data, and supplementary data will be published at a later date. Philips
Semiconductors reserves the right to make changes at any time without notice in order to improve design and
supply the best possible product.
Product specificationProductionThis data sheet contains final specifications. Philips Semiconductors reserves the right to make changes at any
time without notice in order to improve design and supply the best possible product.
[1] Please consult the most recently issued data sheet before initiating or completing a design.
19. Definitions
Short-form specification — The data in a short-form specification is
extracted from a full data sheet with the same type number and title. For
detailed information see the relevant data sheet or data handbook.
Limiting values definition — Limiting values given are in accordance with
the Absolute Maximum Rating System (IEC 60134). Stress above one or
more of the limiting values may cause permanent damage to the device.
These are stress ratings only and operation of the device at these or at any
other conditions above those given in the Characteristics sections of the
specification is not implied. Exposure to limiting values for extended periods
may affect device reliability.
Application information — Applications that are described herein for any
of these products are for illustrative purposes only. Philips Semiconductors
make no representation or warranty that such applications will be suitable for
the specified use without further testing or modification.
20. Disclaimers
Life support — These products are not designed for use in life support
appliances, devices, or systems where malfunction of these products can
reasonably be expected to result in personal injury. Philips Semiconductors
[1]
customers using or selling these products for use in such applications do so
at their own risk and agree to fully indemnify Philips Semiconductors for any
damages resulting from such application.
Right to make changes — Philips Semiconductors reserves the right to
make changes, without notice, in the products, including circuits, standard
cells, and/or software, described or contained herein in order to improve
design and/or performance. Philips Semiconductors assumes no
responsibility or liability for the use of any of these products, conveys no
licence or title under any patent, copyright, or mask work right to these
products, and makes no representations or warranties that these products
are free from patent, copyright, or mask work right infringement, unless
otherwise specified.
21. Licenses
Purchase of Philips I2C components
2
Purchase of Philips I
under the Philips’ I
2
I
C system provided the system conforms to the I2C
specification defined by Philips. This specification can be
ordered using the code 9398 393 40011.
All rights are reserved. Reproduction in whole or in part is prohibited without the prior
written consent of the copyright owner.
The information presented in this document does not form part of any quotation or
contract, is believed to be accurate and reliable and may be changed without notice. No
liability will be accepted by the publisher for any consequence of its use. Publication
thereof does not convey nor imply any license under patent- or other industrial or
intellectual property rights.
Date of release: 21 July 2000Document order number: 9397 750 07338
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