Datasheet MP2106DK, MP2106DQ Datasheet (MPS)

Page 1

MP2106

The Future of Analog IC Technology

DESCRIPTION

The MP2106 is a 1.5A, 800KHz synchronous buck converter designed for low voltage applications requiring high efficiency. It is capable of providing output voltages as low as

0.9V, and integrates top and bottom switches to minimize power loss and component count. The 800KHz switching frequency reduces the size of filtering components, further reducing the solution size.

The MP2106 includes cycle-by-cycle current limiting and under voltage lockout. The internal power switches, combined with the tiny 10-pin MSOP and QFN packages, provide a solution requiring a minimum of surface area.

EVALUATION BOARD REFERENCE

Board Number Dimensions

EV2106DQ/DK-00A 2.5”X x 2.0”Y x 0.5”Z

1.5A, 15V, 800KHz

Synchronous Buck Converter

FEATURES

• 1.5A Output Current

• Synchronous Rectification

• Internal 210m and 255m Power Switches

• Input Range of 2.6V to 15V

• >90% Efficiency

• Zero Current Shutdown Mode

• Under Voltage Lockout Protection

• Soft-Start Operation

• Thermal Shutdown

• Internal Current Limit (Source & Sink)

• Tiny 10-Pin MSOP or QFN Package

APPLICATIONS

• DC/DC Regulation from Wall Adapters

• Portable Entertainment Systems

• Set Top Boxes

• Digital Video Cameras, DECT

• Networking Equipment

• Wireless Modems

TYPICAL APPLICATION

INPUT

2.6V to 15V

OFF ON

C5 C3

10nF

3.3nF

VIN BST

RUN

SS COMP

MP2106

C6 10nF

“MPS” and “The Future of Analog IC Technology” are Trademarks of Monolithic Power Systems, Inc.

Efficiency vs.

10nF

PGNDSGNDVREF

9104

OUTPUT

1.8V / 1.5A

MP2106_TAC_S01

Load Current

100

VIN=3.3V

EFFICIENCY (%)

0.01 0.1 1 10

VIN=5V

LOAD CURRENT (A)

MP2106_TAC_EC02

MP2106 Rev. 1.6 www.MonolithicPower.com 1

Page 2

MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER

PACKAGE REFERENCE

TOP VIEW

SGND

COMP

VREF

RUN

Part Number* Package Temperature

MP2106DK MSOP10

For Tape & Reel, add suffix –Z (eg. MP2106DK–Z)

For Lead Free, add suffix –LF (eg. MP2106DK–LF–Z)

ABSOLUTE MAXIMUM RATINGS

Input Supply Voltage VIN.............................. 16V

LX Voltage V

BST to LX Voltage .........................

Voltage on All Other Pins...............

Storage Temperature...............

Recommended Operating Conditions

Input Supply Voltage VIN..................2.6V to 15V

Output Voltage V

Operating Temperature..............

..................... –0.3V to VIN + 0.3V

........................0.9V to 5.5V

OUT

PGND

VIN

BST

MP2106_PD01-MSOP10

–40°C to +85°C

(1)

–0.3V to +6V –0.3V to +6V

–55°C to +150°C

(2)

–40°C to +85°C

TOP VIEW

COMP

VREF

RUN

EXPOSED PAD

ON BACKSIDE

Part Number** Package Temperature

MP2106DQ

For Tape & Reel, add suffix –Z (eg. MP2106DQ–Z)

For Lead Free, add suffix –LF (eg. MP2106DQ–LF–Z)

Thermal Resistance

QFN10

(3mm x 3mm)

(3)

MSOP10 ................................150 ..... 65... °C/W

QFN10 ....................................50 ...... 12... °C/W

Notes:

1) Exceeding these ratings may damage the device.

2) The device is not guaranteed to function outside of its operating conditions.

3) Measured on approximately 1” square of 1 oz copper.

SGND

PGND

VIN

BST

MP2106_PD02-QFN10

–40°C to +85°C

θJC

ELECTRICAL CHARACTERISTICS

VIN = 5.0V, TA = +25°C, unless otherwise noted.

Parameter Symbol Condition Min Typ Max Units

Input Voltage Range VIN 2.6 15 V Input Under Voltage Lockout 2.2 V Input Under Voltage Lockout

Hysteresis Shutdown Supply Current V Operating Supply Current V VREF Voltage V RUN Input Low Voltage VIL 0.4 V RUN Input High Voltage VHL 1.5 V RUN Hysteresis 100 mV RUN Input Bias Current 1 µA

Oscillator

Switching Frequency fSW 700 800 900 KHz Maximum Duty Cycle D Minimum On Time tON 200 ns

MP2106 Rev. 1.6 www.MonolithicPower.com 2

100 mV

 0.3V 0.5 1.0 µA

RUN

> 2V, VFB = 1.1V 1.2 1.8 mA

RUN

VIN = 2.6V to 15V 2.4 V

REF

VFB = 0.7V 85 %

MAX

Page 3

MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER

ELECTRICAL CHARACTERISTICS (continued)

VIN = 5.0V, TA = +25°C, unless otherwise noted.

Parameter Symbol Condition Min Typ Max Units Error Amplifier

Voltage Gain A Transconductance GEA 300 µA/V COMP Maximum Output Current ±30 µA FB Regulation Voltage VFB 875 895 915 mV FB Input Bias Current IFB VFB = 0.895V –100 nA

Soft-Start

Soft-Start Current ISS 2 µA

Output Switch On-Resistance

Switch On Resistance

Synchronous Rectifier On Resistance

Switch Current Limit (Source) 2.5 A Synchronous Rectifier Current Limit

(Sink) Thermal Shutdown 160

VEA

400 V/V

VIN = 5V 255 m V

= 3V 315 m

VIN = 5V 210 m V

= 3V 255 m

350 mA

°C

PIN FUNCTIONS

Pin # Name Description

Soft-Start Input. Place a capacitor from SS to SGND to set the soft-start period. The MP2106

1 SS

2 FB

3 COMP

4 VREF

5 RUN

6 BST

7 VIN

8 LX

9 PGND

10 SGND Signal Ground.

sources 2µA from SS to the soft-start capacitor at startup. As the SS voltage rises, the feedback threshold voltage increases to limit inrush current during startup.

Feedback Input. FB is the inverting input of the internal error amplifier. Connect a resistive voltage divider from the output voltage to FB to set the output voltage value.

Compensation Node. COMP is the output of the error amplifier. Connect a series RC network to compensate the regulation control loop.

Internal 2.4V Regulator Bypass. Connect a 10nF capacitor between VREF and SGND to bypass the internal regulator. Do not apply any load to VREF.

On/Off Control Input. Drive RUN high to turn on the MP2106; low to turn it off. For automatic startup, connect RUN to VIN via a pullup resistor.

Power Switch Boost. BST powers the gate of the high-side N-Channel power MOSFET switch. Connect a 10nF or greater capacitor between BST and LX.

Internal Power Input. VIN supplies the power to the MP2106 through the internal LDO regulator. Bypass VIN to PGND with a 10µF or greater capacitor. Connect VIN to the input source voltage.

Output Switching Node. LX is the source of the high-side N-Channel switch and the drain of the low-side N-Channel switch. Connect the output LC filter between LX and the output.

Power Ground. PGND is the source of the N-Channel MOSFET synchronous rectifier. Connect PGND to SGND as close to the MP2106 as possible.

MP2106 Rev. 1.6 www.MonolithicPower.com 3

Page 4

MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER

TYPICAL PERFORMANCE CHARACTERISTICS

Circuit of Figure 2, VIN = 5V, V otherwise noted.

= 1.8V, L1 = 5µH, C1 = 10µF, C2 = 22µF, TA = +25°C, unless

OUT

5V/div.

AC Coupled

10mV/div.

AC Coupled

200mV/div.

1A/div.

Steady State Operation

1.5A Load

Startup from Shutdown

1.5A Resistive Load

2V/div.

OUT

1V/div.

MP2106-TPC01

5V/div.

AC Coupled

10mV/div.

AC Coupled

20mV/div.

1A/div.

Steady State Operation

No Load

Load Transient

OUT

AC Coupled

200mV/div.

1A/div.

LOAD

1A/div.

MP2106-TPC02

Startup from Shutdown

No Load

2V/div.

OUT

1V/div.

MP2106-TPC03

1A/div.

5V/div.

1ms/div.

Short Circuit Protection

OUT

1V/div.

1A/div.

MP2106-TPC04

MP2106-TPC06

1A/div.

5V/div.

1ms/div.

Short Circuit Recovery

OUT

1V/div.

1A/div.

MP2106-TPC05

MP2106-TPC07

MP2106 Rev. 1.6 www.MonolithicPower.com 4

Page 5

OPERATION

OFF ON

RUN

ENABLE

CKT & LDO

REGULATOR

MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER

2.6V to 15V

GATE

DRIVE

REGULATOR

Vdr

CURRENT

SENSE

AMPLIFIER

REF

OSCILLATOR

2.4V

800KHz

PWM

COMPARATOR

RAMP

CURRENT

LIMIT

THRESHOLD

-CONTROL

LOGIC

CURRENT

LIMIT

COMPARATOR

UVLO &

THERMAL

SHUTDOWN

Vdr

ERROR

AMPLIFIER

BST

Vdr

OUT

-R2

-PGND

0.895V

310

SGND

COMP

MP2106_BD01

Figure 1—Functional Block Diagram

MP2106 Rev. 1.6 www.MonolithicPower.com 5

Page 6

MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER

The MP2106 measures the output voltage through an external resistive voltage divider and compares that voltage to the internal 0.9V reference in order to generate the error voltage at COMP. The current-mode regulator uses the voltage at COMP and compares it to the inductor current to regulate the output voltage. The use of current-mode regulation improves transient response and improves control loop stability.

At the beginning of each cycle, the high-side N-Channel MOSFET is turned on, forcing the inductor current to rise. The current at the drain of the high-side MOSFET is internally measured and converted to a voltage by the current sense amplifier.

That voltage is compared to the error voltage at COMP. When the inductor current rises sufficiently, the PWM comparator turns off the high-side switch and turns on the low-side

switch, forcing the inductor current to decrease. The average inductor current is controlled by the voltage at COMP, which in turn is controlled by the output voltage. Thus the output voltage controls the inductor current to satisfy the load.

Since the high-side N-Channel MOSFET requires voltages above V

to drive its gate, a

bootstrap capacitor from LX to BST is required to drive the high-side MOSFET gate. When LX is driven low (through the low-side MOSFET), the BST capacitor is internally charged. The voltage at BST is applied to the high-side MOSFET gate to turn it on, and maintains that voltage until the high-side MOSFET is turned off and the low-side MOSFET is turned on, and the cycle repeats. Connect a 10nF or greater capacitor from BST to SW to drive the high-side MOSFET gate.

APPLICATION INFORMATION

INPUT

2.6V to 15V

RUN

10nF

OPEN

Figure 2—Typical Application Circuit

Internal Low-Dropout Regulator

The internal power to the MP2106 is supplied from the input voltage (VIN) through an internal

2.4V low-dropout linear regulator, whose output is VREF. Bypass VREF to SGND with a 10nF or greater capacitor for proper operation. The internal regulator can not supply more current than is required to operate the MP2106. Therefore, do not apply any external load to VREF.

3.3nF C6

10nF

VIN BST

MP2106

PGNDSGNDVREF

FBCOMP

9104

OUTPUT

1.8V / 1.5A

MP2106_TAC_F02

Soft-Start

The MP2106 includes a soft-start timer that slowly ramps the output voltage at startup to prevent excessive current at the input.

When power is applied to the MP2106, and RUN is asserted, a 2µA internal current source charges the external capacitor at SS. As the capacitor charges, the voltage at SS rises. The MP2106 internally limits the feedback threshold voltage at FB to that of the voltage at SS. This forces the output voltage to rise at the same rate as the voltage at SS, forcing the output

MP2106 Rev. 1.6 www.MonolithicPower.com 6

Page 7

MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER

voltage to ramp linearly from 0V to the desired regulation voltage during soft-start.

The soft-start period is determined by the equation:

5C45.0t

Where C5

(in nF) is the soft-start capacitor from

SS to GND, and t

×=

(in ms) is the soft-start period. Determine the capacitor required for a given soft-start period by the equation:

t22.25C ×=

Use values between 10nF and 22nF for C5 to set the soft-start period (between 4ms and 10ms).

Setting the Output Voltage (see Figure 2)

Set the output voltage by selecting the resistive voltage divider ratio. The voltage divider drops the output voltage to the 0.895V feedback voltage. Use 10k for the low-side resistor of the voltage divider. Determine the high-side resistor by the equation:

⎛

OUT

⎜

⎜ ⎝

Where R2 is the high-side resistor, V

⎞ ⎟

1R1

−=

⎟

V895.0

⎠

is the

OUT

output voltage and R1 is the low-side resistor.

Selecting the Input Capacitor

The input current to the step-down converter is discontinuous, and so a capacitor is required to supply the AC current to the step-down converter while maintaining the DC input voltage. A low ESR capacitor is required to keep the noise at the IC to a minimum. Ceramic capacitors are preferred, but tantalum or low ESR electrolytic capacitors may also suffice.

The capacitor can be electrolytic, tantalum or ceramic. Because it absorbs the input switching current it must have an adequate ripple current rating. Use a capacitor with RMS current rating greater than 1/2 of the DC load current.

For stable operation, place the input capacitor as close to the IC as possible. A smaller high quality 0.1µF ceramic capacitor may be placed closer to the IC with the larger capacitor placed further away. If using this technique, it is recommended that the larger capacitor be a tantalum or electrolytic type. All ceramic

capacitors should be placed close to the MP2106. For most applications, a 10µF ceramic capacitor will work.

Selecting the Output Capacitor

The output capacitor (C2) is required to maintain the DC output voltage. Low ESR capacitors are preferred to keep the output voltage ripple to a minimum. The characteristics of the output capacitor also affect the stability of the regulation control system. Ceramic, tantalum, or low ESR electrolytic capacitors are recommended.

The output voltage ripple is:

RIPPLE

OUT

⎛

OUT

⎞

⎜

⎟ ⎟

⎠

+×

ESR

⎜ ⎝

is the input voltage,

is the switching

⎛ ⎜

−×

⎜

⎝

is the output voltage ripple, fSW is

RIPPLE

⎞ ⎟

⎟

××

2Cf8

⎠

Where V the switching frequency, V R

is the equivalent series resistance of the

ESR

output capacitors and f frequency.

Choose an output capacitor to satisfy the output ripple requirements of the design. A 22µF ceramic capacitor is suitable for most applications.

Selecting the Inductor

The inductor is required to supply constant current to the output load while being driven by the switched input voltage. A larger value inductor results in less ripple current that will result in lower output ripple voltage. However, the larger value inductor is likely to have a larger physical size and higher series resistance. Choose an inductor that does not saturate under the worst-case load conditions. A good rule for determining the inductance is to allow the peak-to-peak ripple current to be approximately 30% to 40% of the maximum load current. Make sure that the peak inductor current (the load current plus half the peak-topeak inductor ripple current) is below 2.5A to prevent loss of regulation due to the current limit.

MP2106 Rev. 1.6 www.MonolithicPower.com 7

Page 8

MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER

Calculate the required inductance value by the equation:

()

VVV

−×

OUTINOUT

SWIN

IfV

∆××

Where I is the peak-to-peak inductor ripple current. It is recommended to choose I to be 30%~40% of the maximum load current.

Compensation

The system stability is controlled through the COMP pin. COMP is the output of the internal transconductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the characteristics of the control system.

The DC loop gain is:

A ×××

Where V

⎛ ⎜

VDC

⎜ ⎝

is the feedback voltage, A

⎞

⎟ ⎟

OUT

⎠

RGA

LOADCSVEA

is the

VEA

transconductance error amplifier voltage gain, G

is the current sense transconductance

(roughly the output current divided by the voltage at COMP) and R

is the load

LOAD

resistance:

OUT

Where I

R =

LOAD

is the output load current.

OUT

The system has 2 poles of importance, one is due to the compensation capacitor (C3), and the other is due to the load resistance and the output capacitor (C2), where:

P1 is the first pole, and G

VEA

is the error amplifier

3CA2

××π

transconductance (300µA/V) and

LOAD

2CR2

××π

The system has one zero of importance, due to the compensation capacitor (C3) and the

compensation resistor (R3). The zero is:

3C3R2

××π

If large value capacitors with relatively high equivalent-series-resistance (ESR) are used, the zero due to the capacitance and ESR of the output capacitor can be compensated by a third pole set by R3 and C4. The pole is:

4C3R2

××π

The system crossover frequency (the frequency where the loop gain drops to 1, or 0dB, is important. Set the crossover frequency to below one tenth of the switching frequency to insure stable operation. Lower crossover frequencies result in slower response and worse transient load recovery. Higher crossover frequencies degrade the phase and/or gain margins and can result in instability.

Table 1—Compensation Values for Typical

Output Voltage/Capacitor Combinations

OUT

1.8V 22µF Ceramic 6.8k 3.3nF None

2.5V 22µF Ceramic 9.1k 2.2nF None

3.3V 22µF Ceramic 12k 1.8nF None

47µF Tantalum

1.8V

(300m)

47µF Tantalum

2.5V

(300m)

47µF Tantalum

3.3V

(300m)

C2 R3 C3 C4

13k 2nF 1nF

18k 1.2nF 750pF

24k 1nF 560pF

Choosing the Compensation Components

The values of the compensation components given in Table 1 yield a stable control loop for the given output voltage and capacitor. To optimize the compensation components for conditions not listed in Table 1, use the following procedure.

MP2106 Rev. 1.6 www.MonolithicPower.com 8

Page 9

MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER

Choose the compensation resistor to set the desired crossover frequency. Determine the value by the following equation:

fV2C2

×××π

COUT

VGG

FBCSEA

Where f

is the desired crossover frequency

××

(preferably 33KHz).

Choose the compensation capacitor to set the zero below one fourth of the crossover frequency. Determine the value by the following equation:

××π

f3R

Determine if the second compensation capacitor, C4 is required. It is required if the ESR zero of the output capacitor happens at less than half of the switching frequency. Or:

1fR2C

>×××π

SWESR

If this is the case, then add the second compensation capacitor.

External Boost Diode

For input voltages less than or equal to 5V, it is recommended that an external boost diode be added. This will help improve the regulator efficiency. The diode can be a low cost diode such as an IN4148 or BAT54.

BOOST DIODE

10nF

MP2106_F03

MP2106

BST

Figure 3—External Boost Diode

This diode is also recommended for high duty

cycle operation (when

output voltage (V

OUT

>65%) and high

>12V) applications.

OUT

However, do not exceed the absolute maximum voltage for these pins.

Determine the value by the equation:

R2C

(max)ESR

Where R

ESR(MAX)

is the maximum ESR of the

output capacitor.

MP2106 Rev. 1.6 www.MonolithicPower.com 9

Page 10

MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER

PACKAGE INFORMATION

MSOP10

MP2106 Rev. 1.6 www.MonolithicPower.com 10

Page 11

MP2106 – 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER

QFN10 (3mm x 3mm)

NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications.

Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not assume any legal responsibility for any said applications.

Page 12

Datasheet MP2106DK, MP2106DQ Datasheet (MPS)

Specifications and Main Features

Frequently Asked Questions

User Manual