Datasheet MP1593DNLF Specification

MP1593

3A, 28V, 385kHz

Step-Down Converter

The Future of Analog IC Technology

DESCRIPTION

The MP1593 is a step-down regulator with an internal Power MOSFET. It achieves 3A of continuous output current over a wide input supply range with excellent load and line regulation.

Current mode operation provides fast transient response and eases loop stabilization.

Fault condition protection includes cycle-by-cycle current limiting and thermal shutdown. An adjustable soft-start reduces the stress on the input source at startup. In shutdown mode the regulator draws 20µA of supply current.

The MP1593 requires a minimum number of readily available external components, providing a compact solution.

EVALUATION BOARD REFERENCE

Board Number Dimensions

EV1593DN-00A 2.1”X x 1.3”Y x 0.4”Z

FEATURES

 3A Output Current  Programmable Soft-Start  100mΩ Internal Power MOSFET Switch  Stable with Low ESR Output Ceramic

Capacitors

 Up to 95% Efficiency  20μA Shutdown Mode  Fixed 385kHz Frequency  Thermal Shutdown  Cycle-by-Cycle Over Current Protection  Wide 4.75V to 28V Operating Input Range  Output Adjustable from 1.22V  Under-Voltage Lockout  Available in 8-Pin SOIC Package

APPLICATIONS

 Distributed Power Systems  Battery Chargers  Pre-Regulator for Linear Regulators  Flat Panel TVs  Set-Top Boxes  Cigarette Lighter Powered Devices  DVD/PVR Devices

All MPS parts are lead-free and adhere to the RoHS directive. For MPS green status, please visit MPS website under Products, Quality Assurance page.

“MPS” and “The Future of Analog IC Technology” are registered trademarks of Monolithic Power Systems, Inc.

TYPICAL APPLICATION

C5

1

SW

FB

6

3

5

C3

8.2nF

R3

5.6kΩ

10nF

D1 B340A

L1

10μ H

4A

R1

16.9kΩ

1%

R2 10kΩ 1%

OUTPUT

3.3V 3A

C2 22μ F/6.3V CERAMIC x2

INPUT

4.75V to 28V

OFF ON

10μ F/35V CERAMIC

C1

2

7

EN

MP1593

8

SS

GND COMP

4

C4

x2

0.1μ F

C6

(optional)

BSIN

Efficiency vs Load Current

100

95

90

85

80

75

70

65

EFFICIENCY (%)

60

55

50

0 0.5 1.0 1.5 2.0 2.5 3.0 3.5

VIN = 9V

VIN = 24V

VIN = 12V

LOAD CURRENT (A)

MP1593 – 3A, 28V, 385kHz STEP-DOWN CONVERTER

ORDERING INFORMATION

Part Number* Package Top Marking Free Air Temperature (TA)

MP1593DN SOIC8E MP1593DN

* For Tape & Reel, add suffix –Z (e.g. MP1593DN–Z).

For RoHS Compliant packaging, add suffix –LF (e.g. MP1593DN–LF–Z)

PACKAGE REFERENCE

TOP VIEW

BS

1

IN

2

SW

3

GND

4

EXPOSED PAD

ON BACKSIDE

CONNECT TO PIN 4

(2)

(3)

(1)

ABSOLUTE MAXIMUM RATINGS

Supply Voltage VIN........................-0.3V to +30V

Switch Voltage V Boost Voltage V

All Other Pins..................................-0.3V to +6V

Continuous Power Dissipation (T

………………………………………………....2.5W

Junction Temperature...............................150C

Lead Temperature ....................................260C

Storage Temperature .............. -65°C to +150C

Recommended Operating Conditions

Input Voltage VIN............................4.75V to 28V

Operating Junct. Temp (T

...............-0.5V to VIN + 0.3V

SW

..........VSW – 0.3V to VSW + 6V

BS

= +25°C)

A

)........-40C to +125C

J

-40C to +85C

SS

8

EN

7

COMP

6

FB

5

Thermal Resistance

(4)

θ

JA

θJC

SOIC8E (Exposed Pad) ..........50 ...... 10 ... C/W

Notes:

1) Exceeding these ratings may damage the device.

2) The maximum allowable power dissipation is a function of the maximum junction temperature T ambient thermal resistance θ

. The maximum allowable continuous power dissipation at

T

A

any ambient temperature is calculated by P (MAX)-TA)/θJA. Exceeding the maximum allowable power dissipation will cause excessive die temperature, and the regulator will go into thermal shutdown. Internal thermal shutdown circuitry protects the device from permanent damage.

3) The device is not guaranteed to function outside of its operating conditions.

4) Measured on JESD51-7, 4-layer PCB.

(MAX), the junction-to-

J

, and the ambient temperature

JA

(MAX) = (T

D

J

MP1593 – 3A, 28V, 385kHz STEP-DOWN CONVERTER

ELECTRICAL CHARACTERISTICS

VIN = 12V, TA = +25C, unless otherwise noted.

Parameter Symbol Condition Min Typ Max Units

Shutdown Supply Current VEN = 0V 20 30 μA

Supply Current VEN =3V, VFB = 1.4V 1.0 1.2 mA

Feedback Voltage VFB

4.75V  V V

COMP

Error Amplifier Voltage Gain AEA 400 V/V

Error Amplifier Transconductance

High-Side Switch On-Resistance

Low-Side Switch On-Resistance

High-Side Switch Leakage Current

G

EA

R

DS(ON)1

R

DS(ON)2

V

I

COMP

100 140 mΩ

10 Ω

= 0V, VSW = 0V 0 10 μA

EN

Current Limit 4.8 6.2 7.6 A

Current Sense to COMP Transconductance

Oscillation Frequency f

Short Circuit Oscillation Frequency

Maximum Duty Cycle D

Minimum Duty Cycle D

5.4 A/V

G

CS

335 385 435 kHz

OSC1

f

VFB = 0V 25 45 60 kHz

OSC2

VFB = 1.0V 90 %

MAX

VFB = 1.5V 0 %

MIN

EN Rising Threshold 2.05 2.5 2.95 V

EN Threshold Hysteresis 150 mV

Enable Pull Up Current VEN = 0V 1.0 1.7 2.5 μA

Under-Voltage Lockout Threshold

Under-Voltage Lockout Threshold Hysteresis

Rising 3.75 4.05 4.35 V

V

IN

210 mV

Soft-Start Period CSS = 0.1μF 10 ms

Thermal Shutdown

160

 28V

IN

< 2V

= 10μA

1.194 1.222 1.250 V

500 800 1120 μA/V

C

MP1593 – 3A, 28V, 385kHz STEP-DOWN CONVERTER

TYPICAL PERFORMANCE CHARACTERISTICS

Refer to Typical Application Schematic on Page 1

Feedback Voltage vs Temperature

1.245

1.235

1.225

1.215

1.205

FEEDBACK VOLTAGE (V)

1.195

-60 -40 -20 0 20 40 60 80 100 120140

TEMPERATURE (°C)

Soft-Start Waveforms

V

OUT

1V/Div.

I

L

1A/Div.

Peak Current Limit vs Temperature

5.0

4.9

4.8

4.7

4.6

4.5

4.4

4.3

4.2

PEAK CURRENT LIMIT (A)

4.1

4.0

-50 -25 -0 25 50 75 100 125 150 -60 -40 -20 0 20 40 60 80 100 120140

TEMPERATURE (°C)

Turn Off Waveforms

V

Oscillation Frequency vs Temperature

420

410

400

390

380

370

360

350

OSCILLATION FREQUENCY (KHz)

340

TEMPERATURE (°C)

Load Transient Waveforms

OUT

1V/Div.

I

L

1A/Div.

V

OUT

100mV/Div.

I

L

1A/Div.

Switching Waveforms

4ms/Div.

I

L

1A/Div.

V

OUT

10mV/Div.

V

IN

100mV/Div.

V

SW

10V/Div.

Efficiency vs Load Current

100

95

90

85

80

75

70

65

EFFICIENCY (%)

60

55

50

0 500 1000 1500 2000 2500 3000 3500

VIN = 5V

VIN = 24V

VIN = 12V

LOAD CURRENT (mA)

V

= 12V, V

IN

= 3.3V, 1A - 2A STEP

OUT

Efficiency vs Load Current

100

95

90

85

80

75

70

65

EFFICIENCY (%)

60

55

50

0 500 1000 1500 2000 2500 3000 3500

VIN = 9V

VIN = 24V

VIN = 12V

LOAD CURRENT (mA)

MP1593 – 3A, 28V, 385kHz STEP-DOWN CONVERTER

PIN FUNCTIONS

Pin # Name Description

1 BS

2 IN

3 SW

4 GND Ground. Note: Connect the exposed pad to Pin 4.

5 FB

6 COMP

7 EN

8 SS

High-Side Gate Drive Boost Input. BS supplies the drive for the high-side N-Channel MOSFET switch. Connect a 10nF or greater capacitor from SW to BS to power the high-side switch.

Power Input. IN supplies power to the IC. Drive IN with a 4.75V to 28V power source. Bypass IN

to GND with a suitably large capacitor to eliminate noise on the input to the IC. See Input Capacitor.

Power Switching Output. SW is the switching node that supplies power to the output. Connect the output LC filter from SW to the output load. Note that a capacitor is required from SW to BS to power the high-side switch.

Feedback Input. FB senses the output voltage and regulates it. Drive FB with a resistive voltage

divider from the output voltage to ground. The feedback threshold is 1.222V. See Setting the Output Voltage.

Compensation Node. COMP is used to compensate the regulation control loop. Connect a series RC network from COMP to GND. In some cases, an additional capacitor from COMP to GND is

required. See Compensation.

Enable Input. EN is a digital input that turns the regulator on or off. Drive EN high to turn on the regulator; low to turn it off. An Under-Voltage Lockout (UVLO) function can be implemented by the addition of a resistor divider from V

to GND. For complete low current shutdown the EN pin

IN

voltage needs to be less than 1.5V. For automatic startup leave EN disconnected.

Soft-Start Control Input. SS controls the soft-start period. Connect a capacitor from SS to GND to set the soft-start period. A 0.1μF capacitor sets the soft-start period to 10ms. To disable the soft-start feature, leave SS disconnected.

OPERATION

2

IN

1.2V

7

EN

+

SHUTDOWN

-- COMPARATOR

COMPARATOR

--

INTERNAL

REGULATORS

OSCILLATOR

LOCKOUT

45/385KHz

MP1593 – 3A, 28V, 385kHz STEP-DOWN CONVERTER

CURRENT

SENSE

SLOPE

COMP

CLK

AMPLIFIER

+

CURRENT

-- COMPARATOR

+

--

SRQ

5V

1

M1

Q

1.8V

3

M2

BS

SW

2.60V/

2.39V

FREQUENCY

FOLDBACK

COMPARATOR

+

--

1.22V0.7V

5

FB

--

+

AMPLIFIER

ERROR

Figure 1—Functional Block Diagram

The MP1593 is a current-mode step-down regulator. It regulates input voltages from 4.75V to 28V down to an output voltage as low as 1.22V, and is able to supply up to 3A of continuous load current.

The MP1593 uses current-mode control to regulate the output voltage. The output voltage is measured at FB through a resistive voltage divider and amplified through the internal error amplifier. The output current of the transconductance error amplifier is presented at COMP where a network compensates the regulation control system. The voltage at COMP is compared to the internally measured switch current to control the output voltage.

6

COMP

8

SS

The converter uses an internal N-Channel MOSFET switch to step-down the input voltage to the regulated output voltage. Since the MOSFET requires a gate voltage greater than the input voltage, a boost capacitor connected between SW and BS drives the gate. The capacitor is internally charged when SW is low.

An internal 10Ω switch from SW to GND is used to insure that SW is pulled to GND when it is low to fully charge the BS capacitor.

4

GND

MP1593 – 3A, 28V, 385kHz STEP-DOWN CONVERTER

APPLICATION INFORMATION

COMPONENT SELECTION

Setting the Output Voltage

The output voltage is set using a resistive voltage divider from the output voltage to the FB pin. The voltage divider divides the output voltage down to the feedback voltage by the ratio:

2R

VV



OUTFB

Where V

is the feedback voltage and V

FB

the output voltage.

Thus the output voltage is:



OUT

22.1V

R2 can be as high as 100kΩ, but a typical value is 10kΩ. Using that value, R1 is determined by:

OUT

For a 3.3V output voltage, R2 is 10kΩ and R1 is 17kΩ.

Inductor

The inductor is required to supply constant current to the output load while being driven by the switched input voltage. A larger value inductor will result in less ripple current that will result in lower output ripple voltage. However, larger value inductors will have larger physical size, higher series resistance and/or lower saturation current. A good standard for determining the inductance to use is to allow the inductor peak-to-peak ripple current to be approximately 30% of the maximum switch current limit. Also, make sure that the peak inductor current is below the maximum switch current limit. The inductance value can be calculated by:

2R1R



is

OUT



2R1R

2R

)k)(22.1V(18.81R



Choose an inductor that will not saturate under the maximum inductor peak current. The peak inductor current can be calculated by:

V

OUT

V

 



IN



V

II

LOADLP

OUT



S

 

1





Lf2





Where I

is the load current.

LOAD

Table 1 lists a number of suitable inductors from various manufacturers. The choice of which inductor to use mainly depends on the price vs. size requirements and any EMI requirement.

Table 1—Inductor Selection Guide

Package

Vendor/

Model

Sumida

CR75 Open Ferrite 7.0 7.8 5.5

CDH74 Open Ferrite 7.3 8.0 5.2

CDRH5D28 Shielded Ferrite 5.5 5.7 5.5

CDRH6D28 Shielded Ferrite 6.7 6.7 3.0

CDRH104R Shielded Ferrite 10.1 10.0 3.0

Toko

D53LC

Type A

D75C Shielded Ferrite 7.6 7.6 5.1

D104C Shielded Ferrite 10.0 10.0 4.3

D10FL Open Ferrite 9.7 1.5 4.0

Coilcraft

DO3308 Open Ferrite 9.4 13.0 3.0

DO3316 Open Ferrite 9.4 13.0 5.1

Core Type

Shielded Ferrite 5.0 5.0 3.0

Core

Material

Dimensions

(mm)

WL H

 



ΔIf

LS



Where V

V

OUT



L



is the input voltage, fS is the

IN

switching frequency and ΔI



1

L



V

OUT

 

V

IN



is the peak-to-peak

inductor ripple current.

MP1593 – 3A, 28V, 385kHz STEP-DOWN CONVERTER

V

Output Rectifier Diode

The output rectifier diode supplies current to the inductor when the high-side switch is off. Use a Schottky diode to reduce losses due to diode forward voltage and recovery times.

Choose a diode whose maximum reverse voltage rating is greater than the maximum input voltage, and whose current rating is greater than the maximum load current. Table 2 lists example Schottky diodes and manufacturers.

Table 2—Diode Selection Guide

Diode

SK33 30V, 3A Diodes Inc.

SK34 40V, 3A Diodes Inc.

B330 30V, 3A Diodes Inc.

B340 40V, 3A Diodes Inc.

MBRS330 30V, 3A On Semiconductor

MBRS340 40V, 3A On Semiconductor

oltage/Current

Rating

Manufacture

Input Capacitor

The input current to the step-down converter is discontinuous, therefore a capacitor is required to supply the AC current to the step-down converter while maintaining the DC input voltage. Use low ESR capacitors for the best performance. Ceramic capacitors are preferred, but tantalum or low-ESR electrolytic capacitors will also suffice.

Since the input capacitor (C1) absorbs the input switching current it requires an adequate ripple current rating. The RMS current in the input capacitor can be estimated by:

V

OUT

V

 



IN



= 2V

IN

OUT

,



V

OUT





II

LOAD1C

1





V

IN



The worst-case condition occurs at V where:

The input capacitor can be electrolytic, tantalum or ceramic. When using electrolytic or tantalum capacitors, a small, high quality ceramic capacitor (i.e. 0.1μF) should be placed as close to the IC as possible. When using ceramic capacitors, make sure that they have enough capacitance to provide sufficient charge to prevent excessive voltage ripple at the input. The input voltage ripple caused by the capacitance can be estimated by:

OUT

V

 

1





IN



I

LOAD

V



IN

S

V

1Cf



V

OUT

V

IN

 

 

Output Capacitor

The output capacitor is required to maintain the DC output voltage. Ceramic, tantalum or low ESR electrolytic capacitors are recommended.

Low ESR capacitors are preferred to keep the output voltage ripple low. The output voltage ripple can be estimated by:



OUT



 



1



Lf



V



V

OUT

S



V

OUT

V



 



IN







R

ESR



1

 



2Cf8

S



Where L is the inductor value, C2 is the output capacitance value and R

is the equivalent

ESR

series resistance (ESR) value of the output capacitor.

In the case of ceramic capacitors, the impedance at the switching frequency is dominated by the capacitance, which is the main cause of the output voltage ripple. For simplification, the output voltage ripple can be estimated by:

ΔV

OUT

V



OUT

2



S

 



1



2CLf8



V

OUT

V

 



IN



In the case of tantalum or electrolytic capacitors, the ESR dominates the impedance at the switching frequency. For simplification, the output ripple can be approximated to:

I

LOAD



I

1C

2

For simplification, choose the input capacitor whose RMS current rating is greater than half of the maximum load current.

V

OUT





Lf

S

ΔV 

The characteristics of the output capacitor also affect the stability of the regulation system. The MP1593 can be optimized for a wide range of

V

 



1

 

OUT

V



IN



R

ESR

capacitance and ESR values.

MP1593 – 3A, 28V, 385kHz STEP-DOWN CONVERTER

Compensation Components

The MP1593 employs current mode control for easy compensation and fast transient response. The system stability and transient response are controlled through the COMP pin. COMP is the output of the internal transconductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the characteristics of the control system.

The DC gain of the voltage feedback loop is given by:

V

AGRA 

VEACSLOADVDC

Where A G

is the current sense transconductance and

CS

R

is the load resistor value.

LOAD

is the error amplifier voltage gain,

VEA

V

FB

OUT

The system has two poles of importance. One is due to the compensation capacitor (C3) and the output resistor of error amplifier, while the other is due to the output capacitor and the load resistor. These poles are located at:

G

Where G

f



1P

f



2P

is the error amplifier

EA



1



A3C2

R2C2

VEA

LOAD

transconductance.

In this case (as shown in Figure 3), a third pole set by the compensation capacitor (C6) and the compensation resistor (R3) is used to compensate the effect of the ESR zero on the loop gain. This pole is located at:

f



3P

1

3R6C2



The goal of compensation design is to shape the converter transfer function to get a desired loop gain. The system crossover frequency (where the feedback loop has unity gain) is important.

Lower crossover frequencies result in slower line and load transient responses, while higher crossover frequencies could cause system instability. A good standard is to set the crossover frequency to approximately one-tenth of the switching frequency. The switching frequency for the MP1593 is 385KHz, so the desired crossover frequency is around 38KHz.

Table 3 lists the typical values of compensation components for some standard output voltages with various output capacitors and inductors. The values of the compensation components have been optimized for fast transient responses and good stability at given conditions.

The system has one zero of importance, due to the compensation capacitor (C3) and the compensation resistor (R3). This zero is located at:

f



1Z

1

3R3C2



The system may have another zero of importance, if the output capacitor has a large capacitance and/or a high ESR value. The zero, due to the ESR and capacitance of the output capacitor, is located at:

f



ESR

1



R2C2

ESR

MP1593 – 3A, 28V, 385kHz STEP-DOWN CONVERTER

Table 3—Compensation Values for Typical

Output Voltage/Capacitor Combinations

V

L C2 R3 C3 C6

OUT

1.8V 4.7μH

2.5V

3.3V

5V

12V

1.8 4.7μH

2.5V

3.3V

5V

2.5V

3.3V

5V

12V

4.7-

6.8μH

6.8-

10μH

10-

15μH

15-

22μH

4.7-

6.8μH

6.8-

10μH

10-

15μH

4.7-

6.8μH

6.8-

10μH

10-

15μH

15-

22μH

100μF

Ceramic

47μF

Ceramic

22μFx2

Ceramic

22μFx2

Ceramic

22μFx2

Ceramic

100μF

SP-CAP

47μF

SP-CAP

47μF

SP-CAP

47μF

SP CAP

560μF Al.

30mΩ ESR

560μF Al

30mΩ ESR

470μF Al.

30mΩ ESR

220μF Al.

30mΩ ESR

5.6kΩ 3.3nF None

3.9kΩ 5.6nF None

5.6kΩ 8.2nF None

7.5kΩ 10nF None

10kΩ 3.3nF None

5.6kΩ 3.3nF 100pF

4.7kΩ 5.6nF None

6.8kΩ 10nF None

10kΩ 10nF None

10kΩ 5.6nF 1.5nF

10kΩ 8.2nF 1.5nF

15kΩ 5.6nF 1nF

15kΩ 4.7nF 390pF

To optimize the compensation components for conditions not listed in Table 3, the following procedure can be used.

1. Choose the compensation resistor (R3) to set the desired crossover frequency. Determine R3 by the following equation:

V

f2C2

Where f

3R 



is the desired crossover frequency

C



GG



CSEA

OUT

C

V

FB

(which typically has a value no higher than 38KHz).

2. Choose the compensation capacitor (C3) to achieve the desired phase margin. For applications with typical inductor values, setting the compensation zero, f

, below one forth of

Z1

the crossover frequency provides sufficient phase margin.

Determine C3 by the following equation:

3C



4

f3R2



C

Where R3 is the compensation resistor value.

3. Determine if the second compensation capacitor (C6) is required. It is required if the ESR zero of the output capacitor is located at less than half of the 385kHz switching frequency, or the following relationship is valid:

ESR

f

S



2

is

ESR

is the

S

1



R2C2

Where C2 is the output capacitance value, R the ESR value of the output capacitor and f switching frequency. If this is the case, then add the second compensation capacitor (C6) to set the pole f

at the location of the ESR zero.

P3

Determine C6 by the equation:

R2C



6C



Where C2 is the output capacitance value, R

3R

ESR

is

ESR

the ESR value of the output capacitor and R3 is the compensation resistor.

PCB Layout Guide

PCB layout is very important to achieve stable operation. It is highly recommended to duplicate EVB layout for optimum performance.

If change is necessary, please follow these guidelines and take Figure2 and 3 for references.

1) Keep the path of switching current short and minimize the loop area formed by Input cap, high-side MOSFET and low-side MOSFET/schottky diode.

2) Keep the connection of low-side MOSFET/schottky diode between SW pin and input power ground as short and wide as possible.

3) Bypass ceramic capacitors are suggested to be put close to the V

and VCC Pin.

IN

4) Ensure all feedback connections are short and direct. Place the feedback resistors and compensation components as close to the chip as possible.

MP1593 – 3A, 28V, 385kHz STEP-DOWN CONVERTER

5) Route SW away from sensitive analog areas such as FB.

6) Connect IN, SW, and especially GND respectively to a large copper area to cool the chip to improve thermal performance and long-term reliability. For single layer, do not solder exposed pad of the IC.

C4

C5

R4

8 SSBS

1

7

EN

IN 2

C3 R3 C6

6

5FB

COMP

SW

GND

3

4

R1

R2

SGND

L1

C1

PGND

D1

C2

TOP Layer

SGND

Vout

Feeback

5 FB

GND 4

R1 R2

C6 C3

R3

SGND

C5

C4

R4

8 SSBS

1

7 EN

IN 2

6

COMP

SW 3

L1

C1

PGND

D1

C2

Figure 3―PCB Layout (Single Layer)

External Bootstrap Diode

An external bootstrap diode may enhance the efficiency of the regulator, the applicable conditions of external BST diode are:

 V

 Duty cycle is high: D=

=5V or 3.3V; and

OUT

V

OUT

>65%

V

IN

In these cases, an external BST diode is recommended from the output of the voltage regulator to BST pin, as shown in Fig.4

External BST Diode

BST

MP1593

SW

IN4148

C

BST

L

C

OUT

5V or 3.3V

Figure 4—Add Optional External Bootstrap

Diode to Enhance Efficiency

The recommended external BST diode is IN4148, and the BST cap is 0.1~1µF.

Bottom Layer

Figure 2―PCB Layout (Double Layer)

TYPICAL APPLICATION CIRCUITS

INPUT

4.75V to 28V

MP1593 – 3A, 28V, 385kHz STEP-DOWN CONVERTER

C5

10nF

1

BSIN

3

SW

OUTPUT

2.5V 3A

OFF ON

2

7

EN

MP1593

8

SS

GND COMP

C6

(optional)

FB

64

C3

3.3nF

5

D1 B340A

Figure 5—MP1593 with AVX 47μF, 6.3V Ceramic Output Capacitor

C5

3

5

C3

3.3nF

10nF

D1 B340A

OUTPUT

2.5V 3A

INPUT

4.75V to 28V

OFF ON

2

7

EN

MP1593

8

SS

GND COMP

C6

(optional)

1

BSIN

SW

FB

64

Figure 6—MP1593 with Panasonic 47μF, 6.3V Special Polymer Output Capacitor

PACKAGE INFORMATION

0.189(4.80)

0.197(5.00)

85

MP1593 – 3A, 28V, 385kHz STEP-DOWN CONVERTER

SOIC8E (EXPOSED PAD)

0.124(3.15)

0.136(3.45)

PIN 1 ID

0.013(0.33)

0.020(0.51)

0.024(0.61)

0.063(1.60)

0.150(3.80)

0.157(4.00)

14

TOP VIEW

0.051(1.30)

0.067(1.70) SEATING PLANE

0.000(0.00)

0.006(0.15)

0.050(1.27) BSC

FRONT VIEW

0.050(1.27)

0.228(5.80)

0.244(6.20)

SEE DETAIL "A"

GAUGE PLANE

0.010(0.25) BSC

o

0o-8

BOTTOM VIEW

SIDE VIEW

0.010(0.25)

0.020(0.50)

0.016(0.41)

0.050(1.27)

DETAIL "A"

0.089(2.26)

0.101(2.56)

o

x 45

0.0075(0.19)

0.0098(0.25)

0.103(2.62)

0.213(5.40)

NOTE:

1) CONTROL DIMENSION IS IN INCHES. DIMENSION IN BRACKET IS IN MILLIMETERS.

2) PACKAGE LENGTH DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.

3) PACKAGE WIDTH DOES NOT INCLUDE INTERLEAD FLASH

0.138(3.51)

RECOMMENDED LAND PATTERN

OR PROTRUSIONS.

4) LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.004" INCHES MAX.

5) DRAWING CONFORMS TO JEDEC MS-012, VARIATION BA.

6) DRAWING IS NOT TO SCALE.

NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third

party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not assume any legal responsibility for any said applications.