Datasheet MP1583DNLF Specification

Page 1
MP1583
3A, 23V, 385KHz
Step-Down Converter
The Future of Analog IC Technology
DESCRIPTION
The MP1583 is a step-down regulator with a built-in internal Power MOSFET. It achieves 3A of continuous output current over a wide input supply range with excellent load and line regulation.
Current mode operation provides fast transient response and eases loop stabilization.
Fault condition protection includes cycle-by-cycle current limiting and thermal shutdown. An adjustable soft-start reduces the stress on the input source at startup. In shutdown mode the regulator draws 20A of supply current.
The MP1583 requires a minimum number of external components, providing a compact solution.
FEATURES
3A Output Current
Programmable Soft-Start
100m Internal Power MOSFET Switch
Stable with Low ESR Output Ceramic Capacitors
Up to 95% Efficiency
20A Shutdown Mode
Fixed 385KHz Frequency
Thermal Shutdown
Cycle-by-Cycle Over Current Protection
Wide 4.75V to 23V Operating Input Range
Output Adjustable from 1.22V to 21V
Under-Voltage Lockout
APPLICATIONS
Distributed Power Systems
Battery Chargers
Pre-Regulator for Linear Regulators
TYPICAL APPLICATION
INPUT
4.75V to 23V
OPEN =
AUTOMATIC
STARTUP
CERAMIC
10μ F
2
7
EN
8
10nF
MP1583
SS
GND COMP
BSIN
1
SW
FB
64
B330A
5.6nF
3.9kΩ
3
5
10nF
15μ H
10.5kΩ
10kΩ
“MPS” and “The Future of Analog IC Technology” are Registered Trademarks of Monolithic Power Systems, Inc.
Efficiency Curve
V
= 10V
IN
OUTPUT
2.5V 3A
22μ F CERAMIC
100
90
V
80
70
EFFICIENCY (%)
60
50
OUT
0 0.5 1.0 1.5 2.0 2.5 3.0 3.5
LOAD CURRENT (A)
=2.5V
V
V
OUT
OUT
=5.0V
=3.3V
MP1583_EC01
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Page 2
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
ORDERING INFORMATION
Part Number* Package Top Marking Free Air Temperature (TA)
MP1583DN SOIC8E MP1583DN MP1583DP PDIP8 MP1583DP
–40°C to +85°C –40°C to +85°C
* For Tape & Reel, add suffix –Z (e.g. MP8736DL–Z)
For RoHS compliant packaging, add suffix –LF (e.g. MP8736DL–LF–Z)
PACKAGE REFERENCE
TOP VIEW
SOIC8N/PDIP8
BS
1
IN
2
SW
3
GND
4
EXPOSED PAD
(SOIC8N ONLY)
CONNECT TO PIN 4
ABSOLUTE MAXIMUM RATINGS
(1)
Supply Voltage VIN.......................–0.3V to +28V
Switch Voltage V Bootstrap Voltage V
................. –1V to VIN + 0.3V
SW
....VSW – 0.3V to VSW + 6V
BS
FB, COMP and SS Pins.................–0.3V to +6V
Continuous Power Dissipation (T
= +25°C)
A
(2)
SOIC8E...................................................... 2.5W
PDIP8 ........................................................ 1.2W
Junction Temperature...............................150°C
Lead Temperature ....................................260°C
Storage Temperature ............. –65°C to +150°C
Recommended Operating Conditions
(3)
Input Voltage VIN............................4.75V to 23V
Operating Junct. Temp (T
)...... -40°C to +125°C
J
SS
8
EN
7
COMP
6
FB
5
MP1583_PD01
Thermal Resistance
(4)
θ
JA
θJC
SOIC8E .................................. 50...... 10... °C/W
PDIP8 .................................... 104 ..... 45... °C/W
Notes:
1) Exceeding these ratings may damage the device.
2) The maximum allowable power dissipation is a function of the maximum junction temperature T ambient thermal resistance
. The maximum allowable continuous power dissipation at
T
A
any ambient temperature is calculated by P T
)/JA. Exceeding the maximum allowable power dissipation
A
will cause excessive die temperature, and the regulator will go into thermal shutdown. Internal thermal shutdown circuitry protects the device from permanent damage.
3) The device is not guaranteed to function outside of its operating conditions.
4) Measured on JESD51-7, 4-layer PCB
(MAX), the junction-to-
J
, and the ambient temperature
JA
(MAX)=(TJ(MAX)-
D
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Page 3
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS
VIN = 12V, TA = +25°C, unless otherwise noted.
Parameters Symbol Condition Min Typ Max Units
Shutdown Supply Current VEN = 0V 20 30 µA
Supply Current VEN = 2.8V, VFB =1.4V 1.0 1.2 mA
Feedback Voltage VFB
Error Amplifier Voltage Gain A
VEA
Error Amplifier Transconductance GEA
High-Side Switch On-Resistance R
Low-Side Switch On-Resistance R
DS(ON)1
DS(ON)2
4.75V V
400 V/V
ΔI
COMP
0.1
10
23V
IN
= ±10A
High-Side Switch Leakage Current VEN = 0V, VSW = 0V 0 10 µA
Current Limit 4.0 4.9 6.0 A
Current Sense to COMP Transconductance GCS 3.8 A/V
Oscillation Frequency fS 335 385 435 KHz
Short Circuit Oscillation Frequency VFB = 0V 25 40 55 KHz
Maximum Duty Cycle D
VFB = 1.0V 90 %
MAX
Minimum Duty Cycle VFB = 1.5V 0 %
EN Shutdown Threshold Voltage 0.9 1.2 1.5 V
Enable Pull Up Current VEN = 0V 1.1 1.8 2.5 µA
EN UVLO Threshold VEN Rising 2.37 2.54 2.71 V
EN UVLO Threshold Hysteresis 210 mV
Soft-Start Period CSS = 0.1µF 10 ms
Thermal Shutdown 160
1.194 1.222 1.250 V
500 800 1120 µA/V
°C
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Page 4
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency Curve
VIN = 7V
100
90
V
=2.5V
80
70
EFFICIENCY (%)
60
50
0 0.5 1.0 1.5 2.0 2.5 3.0 3.5
V
EN
5V/div.
V
OUT
2V/div.
OUT
V
LOAD CURRENT (A)
V
OUT
OUT
=5.0V
=3.3V
MP1583-TPC01
Soft-Start
CSS Open, V
1.5A Resistive Load
V
EN
5V/div.
V
OUT
2V/div.
I
L
1A/div.
V
EN
5V/div.
V
OUT
2V/div.
= 10V, V
IN
OUT
= 3.3V,
MP1583-TPC02
I
1A/div.
L
MP1583-TPC03
1A/div.
I
L
1ms/div.
PIN FUNCTIONS
Pin # Name Description
1 BS
2 IN
3 SW
4 GND Ground. (Note: For the SOIC8E package, connect the exposed pad on backside to Pin 4).
5 FB
6 COMP
High-Side Gate Drive Bootstrap Input. BS supplies the drive for the high-side N-Channel MOSFET switch. Connect a 4.7nF or greater capacitor from SW to BS to power the high-side switch.
Power Input. IN supplies the power to the IC. Drive IN with a 4.75V to 23V power source. Bypass
IN to GND with a suitably large capacitor to eliminate noise on the input to the IC. See Input Capacitor
Power Switching Output. SW is the switching node that supplies power to the output. Connect the output LC filter from SW to the output load. Note that a capacitor is required from SW to BS to power the high-side switch.
Feedback Input. FB senses the output voltage and regulates it. Drive FB with a resistive voltage
divider from the output voltage. The feedback threshold is 1.222V. See Setting the Output Voltage
Compensation Node. COMP is used to compensate the regulation control loop. Connect a series
RC network from COMP to GND to compensate the regulation control loop. See Compensation
MP1583-TPC04
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Page 5
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
PIN FUNCTIONS (continued)
Pin # Name Description
Enable/UVLO. A voltage greater than 2.71V enables operation. For complete low current shutdown the EN pin voltage needs to be at less than 900mV. When the
7 EN
voltage on EN exceeds 1.2V, the internal regulator will be enabled and the soft-start capacitor will begin to charge. The MP1583 will start switching after the EN pin voltage reaches 2.71V. There is 7V zener connected between EN and GND. If EN is driven by external signal, the voltage should never exceed 7V.
8 SS
Soft-Start Control Input. SS controls the soft-start period. Connect a capacitor from SS to GND to set the soft-start period. To disable the soft-start feature, leave SS unconnected.
OPERATION
The MP1583 is a current-mode step-down regulator. It regulates input voltages from 4.75V to 23V down to an output voltage as low as 1.222V, and is able to supply up to 3A of load current.
The MP1583 uses current-mode control to regulate the output voltage. The output voltage is measured at FB through a resistive voltage divider and amplified through the internal error amplifier. The output current of the transconductance error amplifier is presented at COMP where a RC network compensates the regulation control system.
The voltage at COMP is compared to the internally measured switch current to control the output voltage.
The converter uses an internal N-Channel MOSFET switch to step-down the input voltage to the regulated output voltage. Since the MOSFET requires a gate voltage greater than the input voltage, a boost capacitor connected between SW and BS drives the gate. The capacitor is internally charged when SW is low.
An internal 10 switch from SW to GND is used to insure that SW is pulled to GND when SW is low in order to fully charge the BS capacitor.
IN
EN
2
1.2V
7
2.54V
FREQUENCY
FOLDBACK
COMPARATOR
+
SHUTDOWN
-- COMPARATOR
COMPARATOR
--
+
INTERNAL
REGULATORS
OSCILLATOR
LOCKOUT
+
--
5
CURRENT
SENSE
40/385KHz
7V
1.222V0.7V
FB
SLOPE
COMP
CLK
1μ A
--
+
ERROR
AMPLIFIER
GM = 800μ A/V
AMPLIFIER
Σ
+
--
6
COMP
CURRENT
COMPARATOR
+
--
SRQ
Q
1.8V
8
SS
Figure 1—Functional Block Diagram
5V
1
BS
3
SW
4
GND
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MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
APPLICATION INFORMATION
COMPONENT SELECTION
Setting the Output Voltage
The output voltage is set using a resistive voltage divider from the output voltage to the FB pin. The voltage divider divides the output voltage down to the feedback voltage by the ratio:
2
R
VV
=
OUTFB
Where V
is the feedback voltage and V
FB
the output voltage.
Thus the output voltage is:
VV
22.1
OUT
×=
A typical value for R2 can be as high as 100k, but a typical value is 10k. Using that value, R1 is determined by:
OUT
For example, for a 3.3V output voltage, R2 is 10k, and R1 is 17k.
Inductor
The inductor is required to supply constant current to the output load while being driven by the switched input voltage. A larger value inductor will result in less ripple current and lower output ripple voltage. However, larger value inductors have a larger physical size, higher series resistance, and/or lower saturation current. A good rule for determining the inductance to use is to allow the inductor peak-to-peak ripple current to be approximately 30% of the maximum switch current limit. Also, make sure that the peak inductor current is below the maximum switch current limit. The inductance value can be calculated by:
21
RR
+
is
OUT
RR
21
+
R
2
))(22.1(18.81 Ω×= kVVR
Choose an inductor that will not saturate under the maximum inductor peak current.
The peak inductor current can be calculated by:
V
OUT
V
IN
Where I
II
LOADLP
is the load current.
LOAD
V
+=
OUT
S
1
×
Lf2
××
Table 1 lists a number of suitable inductors from various manufacturers. The choice of which inductor to use mainly depends on the price vs. size requirements and any EMI requirements.
Table 1—Inductor Selection Guide
Package
Dimensions
Vendor/
Model
Sumida
CR75 Open Ferrite 7.0 7.8 5.5
CDH74 Open Ferrite 7.3 8.0 5.2
CDRH5D28 Shielded Ferrite 5.5 5.7 5.5
CDRH5D28 Shielded Ferrite 5.5 5.7 5.5
CDRH6D28 Shielded Ferrite 6.7 6.7 3.0
CDRH104R Shielded Ferrite 10.1 10.0 3.0
Toko
D53LC
Type A
D75C Shielded Ferrite 7.6 7.6 5.1
D104C Shielded Ferrite 10.0 10.0 4.3
D10FL Open Ferrite 9.7 1.5 4.0
Coilcraft
DO3308 Open Ferrite 9.4 13.0 3.0
DO3316 Open Ferrite 9.4 13.0 5.1
Core Type
Shielded Ferrite 5.0 5.0 3.0
Core
Material
(mm)
WL H
V
OUT
=
ΔIf
×
LS
Where V
L
is the input voltage, fS is the 385KHz
IN
switching frequency and I
1
×
is the peak-to-peak
L
V
OUT
V
IN
inductor ripple current.
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MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
V
Output Rectifier Diode
The output rectifier diode supplies the current to the inductor when the high-side switch is off. Use a Schottky diode to reduce losses due to the diode forward voltage and recovery times.
Choose a diode whose maximum reverse voltage rating is greater than the maximum input voltage, and whose current rating is greater than the maximum load current. Table 2 lists example Schottky diodes and manufacturers.
Table 2—Diode Selection Guide
Diode
SK33 30V, 3A Diodes Inc.
SK34 40V, 3A Diodes Inc.
B330 30V, 3A Diodes Inc.
B340 40V, 3A Diodes Inc.
MBRS330 30V, 3A On Semiconductor
MBRS340 40V, 3A On Semiconductor
oltage/Current
Rating
Manufacture
Input Capacitor
The input current to the step-down converter is discontinuous, therefore a capacitor is required to supply the AC current to the step-down converter while maintaining the DC input voltage. Use low ESR capacitors for the best performance. Ceramic capacitors are preferred, but tantalum or low-ESR electrolytic capacitors will also suffice.
Since the input capacitor absorbs the input switching current it requires an adequate ripple current rating. The RMS current in the input capacitor can be estimated by:
V
OUT
×
×=
II
1
LOADC
1
V
IN
V
OUT
V
IN
The input capacitor can be electrolytic, tantalum or ceramic. When using electrolytic or tantalum capacitors, a small, high quality ceramic capacitor (i.e. 0.1F) should be placed as close to the IC as possible.
When using ceramic capacitors, make sure that they have enough capacitance to provide sufficient charge to prevent excessive voltage ripple at the input. The input voltage ripple caused by capacitance can be estimated by:
V
OUT
V
IN
V
OUT
IN
1
××
I
V
IN
LOAD
=Δ
×
S
V
Cf
1
Where C1 is the input capacitance value.
Output Capacitor
The output capacitor is required to maintain the DC output voltage. Ceramic, tantalum or low ESR electrolytic capacitors are recommended. Low ESR capacitors are preferred so as to keep the output voltage ripple low. The output voltage ripple can be estimated by:
V
V
OUT
OUT
=Δ
S
1
×
Lf
×
V
OUT
V
IN
+×
R
ESR
1
××
Cf
28
S
Where L is the inductor value, C2 is the output capacitance value and R
is the equivalent
ESR
series resistance (ESR) value of the output capacitor.
In the case of ceramic capacitors, the impedance at the switching frequency is dominated by the capacitance, which is the main cause for the output voltage ripple. For simplification, the output voltage ripple can be estimated by:
The worst-case condition occurs at V where:
I
I =
LOAD
1
C
2
For simplification, choose an input capacitor whose RMS current rating is greater than half of
IN
= 2V
OUT
,
ΔV
OUT
=
In the case of tantalum or electrolytic capacitors, the ESR dominates the impedance at the switching frequency. For simplification, the output ripple can be approximated to:
the maximum load current.
ΔV ×
OUT
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V
OUT
2
×××
S
V
=
S
OUT
×
1
Lf
×
1
CLf
28
V
OUT
×
IN
V
V
OUT
V
IN
R
ESR
Page 8
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
The MP1583 can be optimized for a wide range of capacitance and ESR values.
Compensation Components
The MP1583 employs current mode control for easy compensation and fast transient response. The system stability and transient response are controlled through the COMP pin. COMP is the output of the internal transconductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the characteristics of the control system.
The DC gain of the voltage feedback loop is:
V
AGRA ×××=
VEACSLOADVDC
Where A G
is the current sense transconductance and
CS
R
is the load resistor value.
LOAD
is the error amplifier voltage gain,
VEA
FB
V
OUT
The system has two poles of importance. One is due to the compensation capacitor (C3) and the output resistor of error amplifier while the other is due to the output capacitor and the load resistor. These poles are located at:
G
f××=
1
P
EA
32
π
AC
VEA
In this case, a third pole set by the compensation capacitor (C6) and the compensation resistor (R3) is used to compensate the effect of the ESR zero on the loop gain. This pole is located at:
f
=
3
P
1
π
362
RC
××
The goal of compensation design is to shape the converter transfer function to get a desired loop gain. The system crossover frequency (where the feedback loop has unity gain) is important.
Lower crossover frequencies result in slower line and load transient responses, while higher crossover frequencies could cause system instability. A good standard is to set the crossover frequency to approximately one-tenth of the switching frequency. The switching frequency for the MP1583 is 385KHz, so the desired crossover frequency is around 38KHz.
Table 3 lists the typical values of compensation components for some standard output voltages with various output capacitors and inductors. The values of the compensation components have been optimized for fast transient responses and good stability at given conditions.
Where G
=
2
P
is the error amplifier
EA
1
RCf××
22
π
LOAD
transconductance.
The system has one zero of importance, due to the compensation capacitor (C3) and the compensation resistor (R3). This zero is located at:
f
=
1
Z
1
π
332
RC
××
The system may have another zero of importance, if the output capacitor has a large capacitance and/or a high ESR value. The zero due to the ESR and capacitance of the output capacitor is located at:
ESR
=
1 22
π
RCf××
ESR
Table 3—Compensation Values for Typical
Output Voltage/Capacitor Combinations
(Please reference Fig. 3 and Fig. 4)
V
C2 R3 C3 C6
OUT
2.5V
3.3V
5V
12V
2.5V
3.3V
5V
12V
22F
Ceramic
22F
Ceramic
22F
Ceramic
22F
Ceramic
560F Al.
30m ESR
560F Al
30m ESR
470F Al.
30m ESR
220F Al.
30m ESR
3.9k 5.6nF None
4.7k 4.7nF None
7.5k 4.7nF None
16.9k 1.5nF None
91k 1nF 150pF
120k 1nF 120pF
100k 1nF 120pF
169k 1nF 39pF
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Page 9
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
To optimize the compensation components for conditions not listed in Table 2, the following procedure can be used.
1. Choose the compensation resistor (R3) to set the desired crossover frequency. Determine R3 by the following equation:
V
f2C2
Where f
3R ×
=
is the desired crossover frequency
C
××π
GG
×
CSEA
OUT
C
V
FB
(which typically has a value no higher than 38KHz).
2. Choose the compensation capacitor (C3) to achieve the desired phase margin. For applications with typical inductor values, setting the compensation zero, f
, below one forth of
Z1
the crossover frequency provides sufficient phase margin. Determine C3 by the following equation:
3C
>
4
f3R2
××π
C
Where R3 is the compensation resistor value.
3. Determine if the second compensation capacitor (C6) is required. It is required if the ESR zero of the output capacitor is located at less than half of the 385KHz switching frequency, or if the following relationship is valid:
PCB Layout Guide
PCB layout is very important to achieve stable operation. Please follow these guidelines and take Figure2 and 3 for references.
1) Keep the path of switching current short and minimize the loop area formed by Input cap, high-side and low-side MOSFETs.
2) Keep the connection of low-side MOSFET between SW pin and input power ground as short and wide as possible.
3) Ensure all feedback connections are short and direct. Place the feedback resistors and compensation components as close to the chip as possible.
4) Route SW away from sensitive analog areas such as FB.
5) Connect IN, SW, and especially GND respectively to a large copper area to cool the chip to improve thermal performance and long-term reliability. For single layer, do not solder exposed pad of the IC
5
FB
R1 R2 C6
C3
R3
SGND
SGND
C4
R4
8
SS/REFBS
7
EN
6
COMP
1
××π
R2C2
ESR
If this is the case, then add the second compensation capacitor (C6) to set the pole f at the location of the ESR zero. Determine C6 by the equation:
6C
×
=
f
S
<
2
C5
IN
SW
GND
1
2
3
4
L1
P3
C1
R2C
ESR
3R
PGND
D1
C2
Figure 2PCB Layout (Single Layer)
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MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
C6
R4
8
SS/REFBS
1
C2
7 EN
IN 2
C4 R3 C3
6
5FB
COMP
SW
GND
3
4
R1
R2
SGND
L1
C1
PGND
D1
C5
Top Layer
External Bootstrap Diode
An external bootstrap diode may enhance the efficiency of the regulator, the applicable conditions of external BST diode are:
z V
z Duty cycle is high: D=
=5V or 3.3V; and
OUT
V
OUT
>65%
V
IN
In these cases, an external BST diode is recommended from the output of the voltage regulator to BST pin, as shown in Fig.4
External BST Diode
BST
MP1583
SW
IN4148
C
BST
5V or 3.3V
+
L
C
OUT
Figure 4—Add Optional External Bootstrap
Diode to Enhance Efficiency
The recommended external BST diode is IN4148, and the BST cap is 0.1~1µF.
Bottom Layer
Figure 3PCB Layout (Double Layer)
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MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
TYPICAL APPLICATION CIRCUITS
INPUT
4.75V to 23V
OPEN = AUTOMATIC
STARTUP
10μ F/25V CERAMIC
C1
Figure 5—3.3V output 3A solution with Murata 22µF, 10V Ceramic Output Capacitor
2
7
EN
8
SS
GND COMP
C4 10nF
MP1583
C6
NS
BSIN
1
SW
FB
64
C3
4.7nF R3
4.7kΩ
C5
10nF
L1
3
5
D1
15μ H
R1
16.9kΩ
R2 10kΩ
OUTPUT
3.3V 3A
C2 22μ F/10V Murata
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Page 12
PACKAGE INFORMATION
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
SOIC8E (EXPOSED PAD)
PIN 1 ID
0.013(0.33)
0.020(0.51)
0.189(4.80)
0.197(5.00)
85
0.150(3.80)
0.157(4.00)
14
TOP VIEW
0.051(1.30)
0.067(1.70) SEATING PLANE
0.000(0.00)
0.006(0.15)
0.050(1.27) BSC
FRONT VIEW
0.228(5.80)
0.244(6.20)
BOTTOM VIEW
SEE DETAIL "A"
GAUGE PLANE
0.010(0.25) BSC
0.124(3.15)
0.136(3.45)
SIDE VIEW
0.010(0.25)
0.020(0.50)
0.089(2.26)
0.101(2.56)
0.0075(0.19)
0.0098(0.25)
o
x 45
0.024(0.61)
0.063(1.60)
0.050(1.27)
0o-8
o
0.016(0.41)
0.050(1.27)
DETAIL "A"
0.103(2.62)
0.213(5.40)
NOTE:
1) CONTROL DIMENSION IS IN INCHES. DIMENSION IN BRACKET IS IN MILLIMETERS.
2) PACKAGE LENGTH DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
3) PACKAGE WIDTH DOES NOT INCLUDE INTERLEAD FLASH
0.138(3.51)
RECOMMENDED LAND PATTERN
OR PROTRUSIONS.
4) LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.004" INCHES MAX.
5) DRAWING CONFORMS TO JEDEC MS-012, VARIATION BA.
6) DRAWING IS NOT TO SCALE.
MP1583 Rev. 3.1 www.MonolithicPower.com 12 6/20/2011 MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2011 MPS. All Rights Reserved.
Page 13
MP1583 – 3A, 23V, 385KHz STEP-DOWN CONVERTER
PDIP8
0.387 (9.830)
0.367 (9.322)
0.040 (1.016)
0.020 (0.508)
PIN 1 IDENT.
0.260 (6.604)
0.240 (6.096)
0.065 (1.650)
0.050 (1.270)
0.035 (0.889)
0.015 (0.381)
0.140(3.556)
0.120(3.048)
0.100 BSC(2.540)
0.021(0.533)
0.015(0.381)
0.325(8.255)
0.300(7.620)
0.014 (0.356)
0.008 (0.200)
0.392(9.957)
0.332(8.433)
0.145(3.683)
0.134(3.404)
3°~11° Lead Bend
NOTE:
1) Control dimension is in inches. Dimension in bracket is millimeters.
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications.
Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not assume any legal responsibility for any said applications.
MP1583 Rev. 3.1 www.MonolithicPower.com 13 6/20/2011 MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2011 MPS. All Rights Reserved.
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