Analog Devices TMP01FS, TMP01FP, TMP01FJ, TMP01ES Datasheet

REV. D

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.

a

TMP01

*

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] www.analog.com
Fax: [phone] © Analog Devices, Inc., 2002

Low Power Programmable

Temperature Controller

FUNCTIONAL BLOCK DIAGRAM

VPTAT

V+

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

2.5V

SENSOR

1

2

3

4

8

7

6

5

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

R1

TMP01

VREF

SET

HIGH

SET

LOW

GND

R2

R3

UNDER

OVER

FEATURES
–558C to +1258C (–678F to +2578F) Operation

61.08C Accuracy Over Temperature (typ)
Temperature-Proportional Voltage Output
User-Programmable Temperature Trip Points
User-Programmable Hysteresis
20 mA Open Collector Trip Point Outputs
TTL/CMOS Compatible
Single-Supply Operation (4.5 V to 13.2 V)
Low-Cost 8-Pin DIP and SO Packages

APPLICATIONS
Over/Under Temperature Sensor and Alarm
Board Level Temperature Sensing
Temperature Controllers
Electronic Thermostats
Thermal Protection
HVAC Systems
Industrial Process Control
Remote Sensors

GENERAL DESCRIPTION

The TMP01 is a temperature sensor that generates a voltage
output proportional to absolute temperature and a control signal
from one of two outputs when the device is either above or below
a specific temperature range. Both the high/low temperature trip
points and hysteresis (overshoot) band are determined by user­selected external resistors. For high volume production, these
resistors are available on-board.

The TMP01 consists of a band gap voltage reference combined
with a pair of matched comparators. The reference provides
both a constant 2.5 V output and a voltage proportional to
absolute temperature (VPTAT) which has a precise temperature
coefficient of 5 mV/K and is 1.49 V (nominal) at 25°C. The
comparators compare VPTAT with the externally set tempera­ture trip points and generate an open-collector output signal
when one of their respective thresholds has been exceeded.

*Protected by U.S. Patent No. 5,195,827.

Hysteresis is also programmed by the external resistor chain and
is determined by the total current drawn out of the 2.5 V refer­ence. This current is mirrored and used to generate a hysteresis
offset voltage of the appropriate polarity after a comparator has
been tripped. The comparators are connected in parallel, which
guarantees that there is no hysteresis overlap and eliminates
erratic transitions between adjacent trip zones.

The TMP01 utilizes proprietary thin-film resistors in conjunc­tion with production laser trimming to maintain a temperature
accuracy of ±1°C (typical) over the rated temperature range,
with excellent linearity. The open-collector outputs are capable
of sinking 20 mA, enabling the TMP01 to drive control relays
directly. Operating from a 5 V supply, quiescent current is only
500 µA (max).

The TMP01 is available in low-cost 8-pin epoxy mini-DIP and
SO (small outline) packages.

REV. D

–2–

TMP01FP, TMP01ES/TMP01FS–SPECIFICATIONS

Plastic DIP and Surface Mount

Packages (V+ = 5 V, GND = O V, –40ⴗC ≤ TA ≤ +85ⴗC, unless otherwise noted.)

Parameter Symbol Conditions Min Typ Max Unit

INPUTS SET HIGH, SET LOW

Offset Voltage V

OS

0.25 mV

Offset Voltage Drift TCV

OS

3 µV/°C

Input Bias Current, “E” I

B

25 50 nA

Input Bias Current, “F” I

B

25 100 nA

OUTPUT VPTAT

1

Output Voltage VPTAT TA = 25°C, No Load 1.49 V
Scale Factor TC

VPTAT

5 mV/K

Temperature Accuracy, “E” T

A

= 25°C, No Load –1.5 ±0.5 1.5 °C

Temperature Accuracy, “F” T

A

= 25°C, No Load –3 ±1.0 3 °C

Temperature Accuracy, “E” 10°C < T

A

< 40°C, No Load ±0.75 °C

Temperature Accuracy, “F” 10°C < T

A

< 40°C, No Load ±1.5 °C

Temperature Accuracy, “E” –40°C < T

A

< 85°C, No Load –3.0 ± 1 3.0 °C

Temperature Accuracy, “F” –40°C < T

A

< 85°C, No Load –5.0 ± 2 5.0 °C

Temperature Accuracy, “E” –55°C < T

A

< 125°C, No Load ±1.5 °C

Temperature Accuracy, “F” –55°C < T

A

< 125°C, No Load ±2.5 °C

Repeatability Error

4

∆VPTAT 0.25 Degree

Long-Term Drift Error

2,6

0.25 0.5 Degree

Power Supply Rejection Ratio PSRR TA = 25°C, 4.5 V ≤ V+ ≤ 13.2 V ±0.02 ± 0.1 %/V

OUTPUT VREF

Output Voltage, “E” VREF T

A

= 25°C, No Load 2.495 2.500 2.505 V

Output Voltage, “F” VREF T

A

= 25°C, No Load 2.490 2.500 2.510 V

Output Voltage, “E” VREF –40°C < T

A

< 85°C, No Load 2.490 2.500 2.510 V

Output Voltage, “F” VREF –40°C < T

A

< 85°C, No Load 2.485 2.500 2.515 V

Output Voltage, “E” VREF –55°C < T

A

< 125°C, No Load 2.5 ± 0.01 V

Output Voltage, “F” VREF –55°C < T

A

< 125°C, No Load 2.5 ± 0.015 V

Drift TC

VREF

–10 ppm/°C
Line Regulation 4.5 V ≤ V+ ≤ 13.2 V ±0.01 ± 0.05 %/V
Load Regulation 10 µA ≤ I

VREF

≤ 500 µA ±0.1 ±0.25 %/mA

Output Current, Zero Hysteresis I

VREF

7 µA
Hysteresis Current Scale Factor

1

SF

HYS

5.0 µA/°C

Turn-On Settling Time To Rated Accuracy 25 µs

OPEN-COLLECTOR OUTPUTS OVER, UNDER

Output Low Voltage V

OL

I

SINK

= 1.6 mA 0.25 0.4 V

V

OL

I

SINK

= 20 mA 0.6 V

Output Leakage Current I

OH

V+ = 12 V 1 100 µA

Fall Time t

HL

See Test Load 40 ns

POWER SUPPLY

Supply Range V+ 4.5 13.2 V
Supply Current I

SY

Unloaded, +V = 5 V 400 500 µA

I

SY

Unloaded, +V = 13.2 V 450 800 µA

Power Dissipation P

DISS

+V = 5 V 2.0 2.5 mW

NOTES

1

K = °C + 273.15.

2

Guaranteed but not tested.

3

Does not consider errors caused by heating due to dissipation of output load currents.

4

Maximum deviation between 25°C readings after temperature cycling between –55 °C and +125°C.

5

Typical values indicate performance measured at TA = 25°C.

6

Observed in a group sample over an accelerated life test of 500 hours at 150°C.

Specifications subject to change without notice.

Test Load

20pF

1k⍀

V+

REV. D

–3–

TMP01

TMP01

TMP01FJ–SPECIFICATIONS

TO-99 Metal Can Package (V+ = 5 V, GND = O V, –40ⴗC ≤ TA ≤ +85ⴗC,
unless otherwise noted.)

Parameter Symbol Conditions Min Typ Max Unit

INPUTS SET HIGH, SET LOW

Offset Voltage V

OS

0.25 mV

Offset Voltage Drift TCV

OS

3 µV/°C

Input Bias Current, “F” I

B

25 100 nA

OUTPUT VPTAT

1

Output Voltage VPTAT TA = 25°C, No Load 1.49 V
Scale Factor TC

VPTAT

5 mV/K

Temperature Accuracy, “F” T

A

= 25°C, No Load –3 ± 1.0 3 °C

10°C < T

A

< 40°C, No Load ±1.5 °C

–40°C < T

A

< 85°C, No Load –5.0 ± 2 5.0 °C

–55°C < T

A

< 125°C, No Load ±2.5 °C

Repeatability Error

4

∆VPTAT 0.25 Degree

Long-Term Drift Error

2,6

0.25 0.5 Degree

Power Supply Rejection Ratio PSRR TA = 25°C, 4.5 V ≤ V+ ≤ 13.2 V ±0.02 ±0.1 %/V

OUTPUT VREF

Output Voltage, “F” VREF T

A

= 25°C, No Load 2.490 2.500 2.510 V

VREF –40°C < T

A

< 85°C, No Load 2.480 2.500 2.520 V

VREF –55°C < T

A

< 125°C, No Load 2.5 ± 0.015 V

Drift TC

VREF

–10 ppm/°C
Line Regulation 4.5 V ≤ V+ ≤ 13.2 V ±0.01 ± 0.05 %/V
Load Regulation 10 µA ≤ I

VREF

≤ 500 µA ±0.1 ±0.25 %/mA

Output Current, Zero Hysteresis I

VREF

7 µA
Hysteresis Current Scale Factor

1

SF

HYS

5.0 µA/°C

Turn-On Settling Time To Rated Accuracy 25 µs

OPEN-COLLECTOR OUTPUTS OVER, UNDER

Output Low Voltage V

OL

I

SINK

= 1.6 mA 0.25 0.4 V

V

OL

I

SINK

= 20 mA 0.6 V

Output Leakage Current I

OH

V+ = 12 V 1 100 µA

Fall Time

2

t

HL

See Test Load 40 ns

POWER SUPPLY

Supply Range V+ 4.5 13.2 V
Supply Current I

SY

Unloaded, +V = 5 V 400 500 µA

I

SY

Unloaded, +V = 13.2 V 450 800 µA

Power Dissipation P

DISS

+V = 5 V 2.0 2.5 mW

NOTES

1

K = °C + 273.15.

2

Guaranteed but not tested.

3

Does not consider errors caused by heating due to dissipation of output load currents.

4

Maximum deviation between 25°C readings after temperature cycling between –55 °C and +125°C.

5

Typical values indicate performance measured at TA = 25°C.

6

Observed in a group sample over an accelerated life test of 500 hours at 150°C.

Specifications subject to change without notice.

REV. D

TMP01

–4–

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the TMP01 features proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.

WARNING!

ESD SENSITIVE DEVICE

ABSOLUTE MAXIMUM RATINGS

1

Maximum Supply Voltage . . . . . . . . . . . . . . . . –0.3 V to +15 V

Maximum Input Voltage

(SETHIGH, SETLOW) . . . . . . . . . –0.3 V to [(V+) +0.3 V]

Maximum Output Current (VREF, VPTAT) . . . . . . . . . 2 mA

Maximum Output Current (Open-Collector Outputs) . . 50 mA

Maximum Output Voltage (Open-Collector Outputs) . . . . 15 V

Operating Temperature Range . . . . . . . . . . . –55°C to +150°C

Dice Junction Temperature . . . . . . . . . . . . . . . . . . . . . . 150°C

Storage Temperature Range . . . . . . . . . . . . – 65°C to +150°C

Lead Temperature (Soldering 60 sec) . . . . . . . . . . . . . . 300°C

NOTES

1

Stresses above those listed under Absolute Maximum Ratings may cause perma-

nent damage to the device. This is a stress rating; functional operation at or above
this specification is not implied. Exposure to the above maximum rating conditions
for extended periods may affect device reliability.

2

Digital inputs and outputs are protected, however, permanent damage may occur

on unprotected units from high energy electrostatic fields. Keep units in conduc­tive foam or packaging at all times until ready to use. Use proper antistatic handling
procedures.

3

Remove power before inserting or removing units from their sockets.

Package Type θ

JA

θ

JC

Unit

8-Pin Plastic DIP (P) 103

1

43 °C/W

8-Lead SOIC (S) 158

2

43 °C/W

8-Lead TO-99 Can (J) 150

1

18 °C/W

NOTES

1

θJA is specified for device in socket (worst-case conditions).

2

θJA is specified for device mounted on PCB.

ORDERING GUIDE

Temperature Package Package

Model/Grade Range

l

Description Option

TMP01FP XIND Plastic DIP N-8
TMP01ES XIND SOIC SO-8
TMP01FS XIND SOIC SO-8
TMP01FJ

2

XIND TO-99 Can H-08A

NOTES

1

XIND = –40°C to +85°C.

2

Consult factory for availability of MIL/883 version in TO-99 can.

GENERAL DESCRIPTION

The TMP01 is a linear voltage-output temperature sensor,
with a window comparator that can be programmed by the user
to activate one of two open-collector outputs when a predeter­mined temperature setpoint voltage has been exceeded. A low
drift voltage reference is available for setpoint programming.

The temperature sensor is basically a very accurate, temperature
compensated, band gap-type voltage reference with a buffered
output voltage proportional to absolute temperature (VPTAT),
accurately trimmed to a scale factor of 5 mV/K. See the Applica­tions Information following.

The low drift 2.5 V reference output VREF is easily divided
externally with fixed resistors or potentiometers to accurately
establish the programmed heat/cool setpoints, independent of
temperature. Alternatively, the setpoint voltages can be supplied
by other ground referenced voltage sources such as user­programmed DACs or controllers. The high and low setpoint
voltages are compared to the temperature sensor voltage, thus
creating a two-temperature thermostat function. In addition,
the total output current of the reference (I

VREF

) determines the
magnitude of the temperature hysteresis band. The open collector
outputs of the comparators can be used to control a wide variety
of devices.

VPTAT

V+

ENABLE

TMP01

VREF

SET

HIGH

SET

LOW

GND

7

8

1

2

3

4

6

HYSTERESIS

CURRENT

CURRENT

MIRROR

I

HYS

VOLTAGE

REFERENCE

AND

SENSOR

1k⍀

HYSTERESIS
VOLTAGE

5

TEMPERATURE

OUTPUT

WINDOW

COMPARATOR

UNDER

OVER

Figure 2. Detailed Block Diagram

REV. D

–5–

Typical Performance Characteristics–TMP01

20501510

SUPPLY VOLTAGE – V

SUPPLY CURRENT – ␮A

550

350

400

375

450

425

475

500

525

+25ⴗC

+125ⴗC

+85ⴗC

–55ⴗC

–40ⴗC

TPC 1. Supply Current vs. Supply Voltage

5.0

3.0

4.5

3.5

4.0

–75 125–50 1007550250–25

TEMPERATURE – ⴗC

MINIMUM SUPPLY VOLTAGE – V

TPC 2. Minimum Supply Voltage vs. Temperature

–75 125–50 1007550250–25

+2.0

–3.0

+1.0

–1.0

0

+1.5

+0.5

–0.5

–1.5

TEMPERATURE – ⴗC

VPTAT ERROR – ⴗC

V+ = 5V

TPC 3. VPTAT Accuracy vs. Temperature

2.508

2.496

2.500

2.504

2.506

2.502

2.498

–75 125–50 1007550250–25

TEMPERATURE – ⴗC

VREF – V

V+ = 5V

TPC 4. VREF Accuracy vs. Temperature

6.0

0

3.0

1.0

2.0

5.0

4.0

50100403020

VC = 15V
V+ = 5V

T

A

= 25ⴗC

I

C

– mA

V

CE

– V

TPC 5. Open-Collector Output (

OVER, UNDER

) Saturation

Voltage vs. Output Current

2.510

2.490

2.496

2.492

2.494

2.502

2.498

2.500

2.504

2.506

2.508

10002000 800400 600

X + 3␴

X

X – 3␴

CURVES NOT NORMALIZED
EXTRAPOLATED FROM OPERATING LIFE DATA

T = HOURS OF OPERATION AT 125ⴗC; V+ = 5V

VREF – V

TPC 6. VREF Long Term Drift Accelerated by Burn-In

REV. D

TMP01

–6–

100

1k 1M100k10k

–20

100

40

20

0

60

80

FREQUENCY – Hz

PSRR – dB

V+ = 5V
I

VREF

= 10␮A

TPC 7. VREF Power Supply Rejection vs. Frequency

1.0

0.1

0.01

–75 –50 1251007550250–25

OFFSET VOLTAGE – mV

V+ = 5V
I

VREF

= 7.5␮A

TPC 8. Set High, Set Low Input Offset Voltage
vs. Temperature

8

0

2

1

4

3

5

6

7

–0.4 –0.24

–0.32

0–0.08–0.16 0.160.08

OFFSET – mV

NUMBER OF DEVICES

V+ = 5V
T

A

= 25ⴗC

I

VREF

= 5␮A

TPC 9. Comparator Input Offset Distribution

7.26.2 7

6.8

6.66.4 87.87.67.4
REFERENCE CURRENT – ␮A

NUMBER OF DEVICES

10

0

2

1

4

3

5

6

7

8

9

V+ = 5V
T

A

= 25ⴗC

TPC 10. Zero Hysteresis Current Distribution

REV. D

TMP01

–7–

Temperature Hysteresis

The temperature hysteresis is the number of degrees beyond the
original setpoint temperature that must be sensed by the TMP01
before the setpoint comparator will be reset and the output
disabled. Figure 2 shows the hysteresis profile. The hysteresis is
programmed by the user by setting a specific load on the refer­ence voltage output VREF. This output current I

VREF

is also
called the hysteresis current, which is mirrored internally and
fed to a buffer with an analog switch.

LO

HI

OUTPUT

VOLTAGE

OVER, UNDER

TEMPERATURE

HYSTERESIS

LOW

HYSTERESIS HIGH =
HYSTERESIS LOW

T

SETLOW

T

SETHIGH

HYSTERESIS

HIGH

Figure 3. TMP01 Hysteresis Profile

After a temperature setpoint has been exceeded and a comparator
tripped, the buffer output is enabled. The output is a current of
the appropriate polarity which generates a hysteresis offset volt­age across an internal 1000 Ω resistor at the comparator input.
The comparator output remains “on” until the voltage at the
comparator input, now equal to the temperature sensor voltage
VPTAT summed with the hysteresis offset, has returned to the
programmed setpoint voltage. The comparator then returns
LOW, deactivating the open-collector output and disabling the
hysteresis current buffer output. The scale factor for the pro­grammed hysteresis current is:

I

HYS

= I

VREF

= 5 µA/°C + 7 µA

Thus since VREF = 2.5 V, with a reference load resistance of
357 kΩ or greater (output current 7 µA or less), the temperature
setpoint hysteresis will be zero degrees. See the temperature
programming discussion below. Larger values of load resistance
will only decrease the output current below 7 µA and will have
no effect on the operation of the device. The amount of hyster­esis is determined by selecting a value of load resistance for
VREF, as shown below.

Programming the TMP01

In the basic fixed-setpoint application utilizing a simple resistor
ladder voltage divider, the desired temperature setpoints are
programmed in the following sequence:

1. Select the desired hysteresis temperature.

2. Calculate the hysteresis current I

VREF

.

3. Select the desired setpoint temperatures.

4. Calculate the individual resistor divider ladder values needed

to develop the desired comparator setpoint voltages at
SETHIGH and SETLOW.

The hysteresis current is readily calculated, as shown. For
example, for 2 degrees of hysteresis, I

VREF

= 17 µA. Next, the

setpoint voltages V

SETHIGH

and V

SETLOW

are determined using

the VPTAT scale factor of 5 mV/K = 5 mV/(°C + 273.15),
which is 1.49 V for 25°C. Then calculate the divider resistors,
based on those setpoints. The equations used to calculate the
resistors are:

V

SETHIGH

= (T

SETHIGH

+ 273.15)(5 mV/°C)

V

SETLOW

= (T

SETLOW

+ 273.15) (5 mV/°C)

R1 (kΩ) = (V

VREF

– V

SETHIGH

)/I

VREF

=

= (2.5 V – V

SETHIGH

)/I

VREF

R2 (kΩ) = (V

SETHIGH

– V

SETLOW

)/I

VREF

R3 (kΩ) = V

SETLOW/IVREF

1

2

3

4

8

7

6

5

(V

VREF

– V

SETHIGH

)/I

VREF

= R1

TMP01

(V

SETHIGH

– V

SETLOW

)/I

VREF

= R2

V

SETLOW

/I

VREF

= R3

V

SETHIGH

V

SETLOW

V

VREF

= 2.5V

I

VREF

GND

V+

VPTAT

UNDER

OVER

Figure 4. TMP01 Setpoint Programming

The total R1 + R2 + R3 is equal to the load resistance needed
to draw the desired hysteresis current from the reference, or I

VREF

.

The formulas shown above are also helpful in understanding
the calculation of temperature setpoint voltages in circuits other
than the standard two-temperature thermostat. If a setpoint
function is not needed, the appropriate comparator should be
disabled. SETHIGH can be disabled by tying it to V+, SETLOW
by tying it to GND. Either output can be left unconnected.

218 248 273 298 323 348 373 398

–55 –25 –18 0 25 50 75 100 125

–67 –25 0 32 50 77 100 150 200 212 257

VPTAT

K

ⴗ

C

ⴗ

F

1.09 1.24 1.365 1.49 1.615 1.74 1.865 1.99

Figure 5. Temperature—VPTAT Scale

REV. D

TMP01

–8–

Understanding Error Sources

The accuracy of the VPTAT sensor output is well characterized
and specified; however, preserving this accuracy in a heating or
cooling control system requires some attention to minimizing
the various potential error sources. The internal sources of
setpoint programming error include the initial tolerances and
temperature drifts of the reference voltage VREF, the setpoint
comparator input offset voltage and bias current, and the hyster­esis current scale factor. When evaluating setpoint programming
errors, remember that any VREF error contribution at the com­parator inputs is reduced by the resistor divider ratios. The
comparator input bias current (inputs SETHIGH, SETLOW)
drops to less than 1 nA (typ) when the comparator is tripped.
This can account for some setpoint voltage error, equal to the
change in bias current times the effective setpoint divider ladder
resistance to ground.

The thermal mass of the TMP01 package and the degree of ther­mal coupling to the surrounding circuitry are the largest factors
in determining the rate of thermal settling, which ultimately
determines the rate at which the desired temperature measure­ment accuracy may be reached. Thus, allow sufficient time for
the device to reach the final temperature. The typical thermal
time constant for the plastic package is approximately 140 sec­onds in still air. Therefore, to reach the final temperature accuracy
within 1%, for a temperature change of 60 degrees, a settling
time of 5 time constants, or 12 minutes, is necessary.

The setpoint comparator input offset voltage and zero hyster­esis current affect setpoint error. While the 7 µA zero hysteresis
current allows the user to program the TMP01 with moderate
resistor divider values, it does vary somewhat from device to
device, causing slight variations in the actual hysteresis obtained
in practice. Comparator input offset directly impacts the pro­grammed setpoint voltage and thus the resulting hysteresis band,
and must be included in error calculations.

External error sources to consider are the accuracy of the pro­gramming resistors, grounding error voltages, and the overall
problem of thermal gradients. The accuracy of the external
programming resistors directly impacts the resulting setpoint
accuracy. Thus, in fixed-temperature applications, the user should
select resistor tolerances appropriate to the desired program­ming accuracy. Resistor temperature drift must be taken into
account also. This effect can be minimized by selecting good
quality components, and by keeping all components in close
thermal proximity. Applications requiring high measurement
accuracy require great attention to detail regarding thermal
gradients. Careful circuit board layout, component placement,
and protection from stray air currents are necessary to minimize
common thermal error sources.

Also, the user should take care to keep the bottom of the setpoint
programming divider ladder as close to GND (Pin 4) as possible
to minimize errors due to IR voltage drops and coupling of exter­nal noise sources. In any case, a 0.1 µF capacitor for power
supply bypassing is always recommended at the chip.

Safety Considerations in Heating and Cooling System Design

Designers should anticipate potential system fault conditions
which may result in significant safety hazards which are outside
the control of and cannot be corrected by the TMP01-based
circuit. Governmental and industrial regulations regarding
safety requirements and standards for such designs should be
observed where applicable.

APPLICATIONS INFORMATION
Self-Heating Effects

In some applications, the user should consider the effects of
self-heating due to the power dissipated by the open-collector
outputs, which are capable of sinking 20 mA continuously. Under
full load, the TMP01 open-collector output device is dissipating

P

DISS

= 0.6 V × .020A = 12 mW

which in a surface-mount SO package accounts for a tempera­ture increase due to self-heating of

∆T = P

DISS

× θJA = .012 W × 158°C/W = 1.9°C.

This will, of course, directly affect the accuracy of the TMP01
and will, for example, cause the device to switch the heating
output “OFF” 2 degrees early. Alternatively, bonding the same
package to a moderate heatsink limits the self-heating effect to
approximately

∆T = P

DISS

× θJC = .012 W × 43°C/W = 0.52°C.

which is a much more tolerable error in most systems. The VREF
and VPTAT outputs are also capable of delivering sufficient
current to contribute heating effects and should not be ignored.

Buffering the Voltage Reference

As mentioned before, the reference output VREF is used to
generate the temperature setpoint programming voltages for the
TMP01 and also is used to determine the hysteresis tempera­ture band by the reference load current I

VREF

. The on-board

output buffer amplifier is typically capable of 500 µA output
drive into as much as 50 pF load (max). Exceeding this load will
affect the accuracy of the reference voltage, could cause thermal
sensing errors due to dissipation, and may induce oscillations.
Selection of a low drift buffer functioning as a voltage follower
with high input impedance will ensure optimal reference accuracy,
and will not affect the programmed hysteresis current. Amplifi­ers which offer the low drift, low power consumption, and low
cost appropriate to this application include the OP295, and
members of the OP90, OP97, OP177 families, and others as
shown in the following applications circuits.

With excellent drift and noise characteristics, VREF offers a good
voltage reference for data acquisition and transducer excitation
applications as well. Output drift is typically better than –10 ppm/
°C, with 315 nV/√

Hz (typ) noise spectral density at 1 kHz.

REV. D

TMP01

–9–

Preserving Accuracy Over Wide Temperature Range
Operation

The TMP01 is unique in offering both a wide range temperature
sensor and the associated detection circuitry needed to imple­ment a complete thermostatic control function in one monolithic
device. While the voltage reference, setpoint comparators, and
output buffer amplifiers have been carefully compensated to
maintain accuracy over the specified temperature range, the
user has an additional task in maintaining the accuracy over
wide operating temperature ranges in this application. Since the
TMP01 is both sensor and control circuit, in many applications,
it is possible that the external components used to program and
interface the device may be subjected to the same temperature
extremes. Thus, it may be necessary to locate components in
close thermal proximity to minimize large temperature differen­tials, and to account for thermal drift errors such as resistor
matching tempcos, amplifier error drift, and the like where
appropriate. Circuit design with the TMP01 requires a slightly
different perspective regarding the thermal behavior of elec­tronic components.

Thermal Response Time

The time required for a temperature sensor to settle to a speci­fied accuracy is a function of the thermal mass of the sensor,
and the thermal conductivity between the sensor and the object
being sensed. Thermal mass is often considered equivalent to
capacitance. Thermal conductivity is commonly specified using
the symbol Q, and can be thought of as the reciprocal of thermal
resistance. It is commonly specified in units of degrees per watt
of power transferred across the thermal joint. Thus, the time
required for the TMP01 to settle to the desired accuracy is
dependent on the package selected, the thermal contact estab­lished in that particular application, and the equivalent power of
the heat source. In most applications, the settling time is prob­ably best determined empirically.

Switching Loads with the Open-Collector Outputs

In many temperature sensing and control applications, some
type of switching is required. Whether it be to turn on a heater
when the temperature goes below a minimum value or to turn
off a motor that is overheating, the open-collector outputs Over
and Under can be used. For the majority of applications, the
switches used need to handle large currents on the order of 1 A
and above. Because the TMP01 is accurately measuring tempera­ture, the open-collector outputs should handle less than 20 mA
of current to minimize self-heating. Clearly, the Overtemp and
Undertemp outputs should not drive the equipment directly.
Instead, an external switching device is required to handle
the large currents. Some examples of these are relays, power
MOSFETs, thyristors, IGBTs, and Darlingtons.

Figures 5a through 5e show a variety of circuits where the
TMP01 controls a switch. The main consideration in these
circuits, such as the relay in Figure 5a, is the current required
to activate the switch.

MOTOR
SHUTDOWN

2604-12-311

COTO

IN4001

OR EQUIV.

12V

R1

R2

R3

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

1

2

3

4

7

HYSTERESIS

GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

8

5

6

Figure 6a. Reed Relay Drive

It is important to check the particular relay to ensure that the
current needed to activate the coil does not exceed the TMP01’s
recommended output current of 20 mA. This is easily deter­mined by dividing the relay coil voltage by the specified coil
resistance. Keep in mind that the inductance of the relay will create
large voltage spikes that can damage the TMP01 output unless
protected by a commutation diode across the coil, as shown.
The relay shown has a contact rating of 10 W maximum. If a
relay capable of handling more power is desired, the larger con­tacts will probably require a commensurately larger coil, with
lower coil resistance and thus higher trigger current. As the
contact power handling capability increases, so does the current
needed for the coil. In some cases, an external driving transistor
should be used to remove the current load on the TMP01 as
explained in the next section.

Power FETs are popular for handling a variety of high current
dc loads. Figure 5b shows the TMP01 driving a p-channel
MOSFET transistor for a simple heater circuit. When the out­put transistor turns on, the gate of the MOSFET is pulled down
to approximately 0.6 V, turning it on. For most MOSFETs, a
gate-to-source voltage or Vgs on the order of –2 V to –5 V is
sufficient to turn the device on. Figure 5c shows a similar circuit
for turning on an n-channel MOSFET, except that now the gate
to source voltage is positive. For this reason, an external transis­tor must be used as an inverter so that the MOSFET will turn
on when the “Under Temp” output pulls down.

NC = NO CONNECT

NC

NC

IRFR9024
OR EQUIV.

HEATING
ELEMENT

2.4k⍀ (12V)

1.2k⍀ (6V)
5%

V+

R1

R2

R3

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

1

2

3

4

7

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

TMP01

VPTATVREF

8

5

6

Figure 6b. Driving a P-Channel MOSFET

REV. D

TMP01

–10–

IRF130

NC = NO CONNECT

NC

NC

2N1711

HEATING
ELEMENT

V+

R1

R2

R3

4.7k⍀ 4.7k⍀

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

7

8

5

6

Figure 6c. Driving a N-Channel MOSFET

Isolated Gate Bipolar Transistors (IGBT) combine many of the
benefits of power MOSFETs with bipolar transistors, and are
used for a variety of high power applications. Because IGBTs
have a gate similar to MOSFETs, turning on and off the devices
is relatively simple as shown in Figure 5d. The turn-on voltage
for the IGBT shown (IRGBC40S) is between 3.0 V and 5.5 V.
This part has a continuous collector current rating of 50 A and a
maximum collector to emitter voltage of 600 V, enabling it to
work in very demanding applications.

IRGBC40S

NC = NO CONNECT

NC

NC

2N1711

V+

R1

R2

R3

4.7k⍀

4.7k⍀

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS

GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

7

8

5

6

MOTOR
CONTROL

Figure 6d. Driving an IGBT

The last class of high power devices discussed here are thyristors,
which includes SCRs and Triacs. Triacs are a useful alternative
to relays for switching ac line voltages. The 2N6073A shown in
Figure 5e is rated to handle 4A (rms). The optoisolated MOC3011.
Triac shown features excellent electrical isolation from the noisy
ac line and complete control over the high power Triac with
only a few additional components.

NC = NO CONNECT

NC

NC

V+ = 5V

R1

R2

R3

300⍀

150⍀

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

TMP01

VPTATVREF

7

8

5

6

MOC3011

1

2

34

5

6

LOAD

AC

2N6073A

Figure 6e. Controlling the 2N6073A Triac

High Current Switching

As mentioned above, internal dissipation due to large loads on
the TMP01 outputs will cause some temperature error due to
self-heating. External transistors remove the load from the TMP01,
so that virtually no power is dissipated in the internal transistors
and no self-heating occurs. Figures 6a through 6c show a few
examples using external transistors. The simplest case, using a
single transistor on the output to invert the output signal is shown
in Figure 6a. When the open collector of the TMP01 turns “ON”
and pulls the output down, the external transistor Q1’s base will
be pulled low, turning off the transistor. Another transistor can
be added to reinvert the signal as shown in Figure 6b. Now,
when the output of the TMP01 is pulled down, the first transis­tor, Q1, turns off and its collector goes high, which turns Q2 on,
pulling its collector low. Thus, the output taken from the collector
of Q2 is identical to the output of the TMP01. By picking a
transistor that can accommodate large amounts of current, many
high power devices can be switched.

2N1711

V+

R1

R2

R3

4.7k⍀

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

7

8

5

6

I

C

Q1

Figure 7a. An External Resistor Minimizes Self-Heating

Q1

Q2

2N1711

V+

R1

R2

R3

4.7k⍀

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

7

8

5

6

I

C

4.7k⍀
2N1711

Figure 7b. Second Transistor Maintains Polarity of
TMP01 Output

An example of a higher power transistor is a standard Darlington
configuration as shown in Figure 6c. The part chosen, TIP-110,
can handle 2 A continuous which is more than enough to con­trol many high power relays. In fact, the Darlington itself can be
used as the switch, similar to MOSFETs and IGBTs.

REV. D

TMP01

–11–

MOTOR
SWITCH

RELAY

12V

2N1711

V+

R1

R2

R3

4.7k⍀

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

7

8

5

6

I

C

4.7k⍀

TIP-110

Figure 7c. Darlington Transistor Can Handle Large Currents

Buffering the Temperature Output Pin

The VPTAT sensor output is a low impedance dc output voltage
with a 5 mV/K temperature coefficient, and is useful in a number
of measurement and control applications. In many applications,
this voltage needs to be transmitted to a central location for
processing. The buffered VPTAT voltage output is capable of
500 µA drive into 50 pF (max). As mentioned in the discussion
above regarding buffering circuits for the VREF output, it is
useful to consider external amplifiers for interfacing VPTAT to
external circuitry to ensure accuracy, and to minimize loading
which could create dissipation-induced temperature sensing
errors. An excellent general purpose buffer circuit using the
OP177 is shown in Figure 7. It is capable of driving over 10 mA,
and will remain stable under capacitive loads of up to 0.1 µF.
Other interfacing ideas are shown below.

Differential Transmitter

In noisy industrial environments, it is difficult to send an accurate
analog signal over a significant distance. However, by sending
the signal differentially on a wire pair, these errors can be signifi­cantly reduced. Since the noise will be picked up equally on
both wires, a receiver with high common-mode input rejection
can be used to cancel out the noise very effectively at the receiving

0.1␮F

V+

V–

C

L

V

OUT

VPTAT

OP177

V+

R1

R2

R3

100⍀

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

7

8

5

6

10k⍀

Figure 8. Buffer VPTAT to Handle Difficult Loads

end. Figure 8 shows two amplifiers being used to send the signal
differentially, and an excellent differential receiver, the AMP03,
which features a common-mode rejection ratio of 95 dB at dc
and very low input and drift errors.

V

OUT

VPTAT

1/2

OP297

V+

R1

R2

R3

50⍀

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

7

8

5

6

10k⍀

V+

V–

10k⍀

1/2

OP297

10k⍀

AMP03

50⍀

Figure 9. Send the Signal Differentially for Noise Immunity

REV. D

TMP01

–12–

4 mA to 20 mA Current Loop

Another very common method of transmitting a signal over long
distances is to use a 4 mA to 20 mA loop, as shown in Figure 9.
An advantage of using a 4 mA to 20 mA loop is that the accuracy
of a current loop is not compromised by voltage drops across
the line. One requirement of 4 mA to 20 mA circuits is that the
remote end must receive all of its power from the loop, mean­ing that the circuit must consume less than 4 mA. Operating from
5 V, the quiescent current of the TMP01 is 500 µA max, and
the OP90s is 20 µA max, totaling less than 4 mA. Although not
shown, the open collector outputs and temperature setting pins
can be connected to do any local control of switching.

The current is proportional to the voltage on the VPTAT output,
and is calibrated to 4 mA at a temperature of –40°C, to 20 mA
for +85°C. The main equation governing the operation of this
circuit gives the current as a function of VPTAT:

I

OUT

=

1

R6

VPTAT × R5

R2

–

VREF × R3

R3 + R1

1 +

R5

R2





















The resulting temperature coefficient of the output current is
128 µA/°C.

5

8

1

4

R

L

2N1711

VREF

GNDV+VPTAT

TMP01

R5

100k⍀

R2

39.2k⍀

7

6

4

3

2

OP90

R1

243k⍀

R3

100k⍀

R6

100⍀

4–20mA

5V TO 13.2V

Figure 10. 4mA to 20 mA Current Loop

To determine the resistor values in this circuit, first note that
VREF remains constant over temperature. Thus the ratio of R5
over R2 must give a variation of I

OUT

from 4 mA to 20 mA as

VPTAT varies from 1.165 V at –40°C to 1.79 V at +85°C. The
absolute value of the resistors is not important, only the ratio.
For convenience, 100 kΩ is chosen for R5. Once R2 is calcu­lated, the value of R3 and R1 is determined by substituting
4 mA for I

OUT

and 1.165 V for VPTAT and solving. The final

values are shown in the circuit. The OP90 is chosen for this
circuit because of its ability to operate on a single supply and its
high accuracy. For initial accuracy, a 10 kΩ trim potentiometer
can be included in series with R3, and the value of R3 lowered
to 95 kΩ. The potentiometer should be adjusted to produce an
output current of 12.3 mA at 25°C.

Temperature-to-Frequency Converter

Another common method of transmitting analog information
is to convert a voltage to the frequency domain. This is easily
done with any of the low cost monolithic voltage-to-frequency
converters (VFCs) available, which feature a robust, open­collector digital output. A digital signal is very immune to noise
and voltage drops because the only important information is the
frequency. As long as the conversions between temperature and
frequency are done accurately, the temperature data can be
successfully transmitted.

A simple circuit to do this combines the TMP01 with an AD654
VFC, as shown in Figure 10. The AD654 outputs a square wave
that is proportional to the dc input voltage according to the
following equation:

F

OUT

=

V

IN

10 (R1+ R2) C

T

By simply connecting the VPTAT output to the input of the
AD654, the 5 mV/°C temperature coefficient gives a sensitivity
of 25 Hz/°C, centered around 7.5 kHz at 25°C. The trimming
resistor R2 is needed to calibrate the absolute accuracy of the
AD654. For more information on that part, consult the AD654
data sheet. Finally, the AD650 can be used to accurately convert
the frequency back to a dc voltage on the receiving end.

4

3

7

6

8

1

2

5

AD654

VPTAT

V+

R1

R2

R3

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

7

8

5

6

R1

1.8k⍀

OSC

V+

F

OUT

C

T

0.1␮F
5k⍀

V+

R2

500⍀

Figure 11. Temperature-to-Frequency Converter

REV. D

TMP01

–13–

IN4148

I

1

I

2

6

5

3

4

1

2

2.5V

V+

REF43

4

6

2

1.16V TO 1.7V

ISOLATION

BARRIER

OP290

V+

V+

R1

R2

R3

100⍀

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

7

8

5

6

7

6

4

3

2

OP290

680pF

R1

470k⍀

V+

7

6

4

3

2

OP90

604k⍀

100k⍀

680pF

IL300XC

Figure 12. Isolation Amplifier

Isolation Amplifier

In many industrial applications, the sensor is located in an envi­ronment that needs to be electrically isolated from the central
processing area. Figure 11 shows a simple circuit that uses an
8-pin optoisolator (IL300XC) that can operate across a 5,000 V
barrier. IC1 (an OP290 single-supply amplifier) is used to drive
the LED connected between Pins 1 to 2. The feedback actually
comes from the photodiode connected from Pins 3 to 4. The
OP290 drives the LED such that there is enough current gener­ated in the photodiode to exactly equal the current derived from
the VPTAT voltage across the 470 kΩ resistor. On the receiving
end, an OP90 converts the current from the second photodiode
to a voltage through its feedback resistor R2. Note that the other
amplifier in the dual OP290 is used to buffer the 2.5 V reference
voltage of the TMP01 for an accurate, low drift LED bias level
without affecting the programmed hysteresis current. A REF43
(a precision 2.5 V reference) provides an accurate bias level at
the receiving end.

To understand this circuit, it helps to examine the overall
equation for the output voltage. First, the current (I1) in the
photodiode is set by:

I

V VPTAT

k

1

25

470

=

−.
Ω

Note that the IL300XC has a gain of 0.73 (typical) with a min
and max of 0.693 and 0.769 respectively. Since this is less than

1.0, R2 must be larger than R1 to achieve overall unity gain. To
show this, the full equation is:

VVIRV

V VPTAT

k

k VPTAT

OUT

=−=−

−











=25 25 07

25

470

644

22

...

.

Ω

Ω

A trim is included for R2 to correct for the initial gain accuracy
of the IL300XC. To perform this trim, simply adjust for an
output voltage equal to VPTAT at any particular temperature.
For example, at room temperature, VPTAT = 1.49 V, so adjust
R2 until V

OUT

= 1.49 V as well. Both the REF43 and the OP90
operate from a single supply, and contribute no significant error
due to drift.

In order to avoid the accuracy trim, and to reduce board space,
complete isolation amplifiers are available, such as the high
accuracy AD202.

Out-of-Range Warning

By connecting the two open collector outputs of the TMP01
together into a “wired-OR” configuration, a temperature “out­of-range” warning signal is generated. This can be useful in
sensitive equipment calibrated to work over a limited temperature
range. R1, R2, and R3 in Figure 11 are chosen to give a tempera­ture range of 10°C around room temperature (25°C). Thus, if
the temperature in the equipment falls below 15°C or rises above
35°C, the Undertemp output or Overtemp output respectively
will go low and turn the LED on. The LED may be replaced
with a simple pull-up resistor to give a logic output for control­ling the instrument, or any of the switching devices discussed
above can be used.

LED

VPTAT

V+

R1

47.5k⍀

R2

4.99k⍀

R3

71.5k⍀

200⍀

TEMPERATURE

SENSOR AND

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

7

8

5

6

Figure 13. Out-of-Range Warning

REV. D

TMP01

–14–

Translating 5 mV/K to 10 mV/ⴗC

A useful circuit shown in Figure 13 translates the VPTAT output
voltage, which is calibrated in Kelvins, into an output that can
be read directly in degrees Celsius on a voltmeter display. To
accomplish this, an external amplifier is configured as a differen­tial amplifier. The resistors are scaled so the VREF voltage will
exactly cancel the VPTAT voltage at 0.0°C.

5

1

+15V

–15V

10pF

V

OUT

(10mV/ⴗC)

(V

OUT

= 0.0V @ T = 0.0ⴗC)

487⍀

7

6

4

3

2

OP177

100k⍀

100k⍀

4.12k⍀

VPTAT

VREF

TMP01

4.22k⍀

105k⍀

Figure 14. Translating 5 mV/K to 10 mV/°C

However, the gain from VPTAT to the output is two, so that
5 mV/K becomes 10 mV/°C. Thus, for a temperature of 80°C,
the output voltage is 800 mV. Circuit errors will be due prima­rily to the inaccuracies of the resistor values. Using 1% resistors,
the observed error was less than 10 mV, or 1°C. The 10 pF
feedback capacitor helps to ensure against oscillations. For
better accuracy, a adjustment potentiometer can be added in
series with either 100 kΩ resistor.

Translating VPTAT to the Fahrenheit Scale

A very similar circuit to the one shown in Figure 13 can be used
to translate VPTAT into an output that can be read directly in
degrees Fahrenheit, with a scaling of 10 mV/°F. Only unity gain
or less is available from the first stage differentiating circuit, so
the second amplifier provides a gain of two to complete the
conversion to the Fahrenheit scale. Using the circuit in Figure
14, a temperature of 0.0°F gives an output of 0.00 V. At room
temperature (70°F), the output voltage is 700 mV. A –40°C to
+85°C operating range translates into –40°F to +185°F. The
errors are essentially the same as for the circuit in Figure 13.

V

OUT

= 0.0V @ T = 0.0ⴗF

(10mV/ⴗF)

5

1

+15V

–15V

10pF

121⍀

7

6

4

3

2

1/2

OP297

100k⍀

100k⍀

6.49k⍀

VPTAT

VREF

TMP01

1.0k⍀

90.9k⍀

100k⍀

7

6

5

1/2

OP297

100k⍀

Figure 15. Translating 5 mV/K to 10 mV/°F

REV. D

TMP01

–15–

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm).

8-Lead Epoxy DIP

(N-8)

0.160 (4.06)

0.115 (2.93)

0.130

(3.30)
MIN

0.210
(5.33)

MAX

0.015
(0.381) TYP

0.430 (10.92)

0.348 (8.84)

0.280 (7.11)

0.240 (6.10)

4

5

8

1

0.070 (1.77)

0.045 (1.15)

0.022 (0.558)

0.014 (0.356)

0.325 (8.25)

0.300 (7.62)

0°- 15°

0.100

(2.54)

BSC

0.015 (0.381)

0.008 (0.204)

SEATING

PLANE

0.195 (4.95)

0.115 (2.93)

8-Lead SOIC

(R-8)

SEATING

PLANE

0.0500 (1.27) BSC

4

58

1

0.2440 (6.20)

0.2284 (5.80)

0.1574 (4.00)

0.1497 (3.80)

0.1968 (5.00)

0.1890 (4.80)

0.0500 (1.27)

0.0160 (0.41)

0°-8°

× 45°

0.0196 (0.50)

0.0099 (0.25)

0.0098 (0.25)

0.0075 (0.19)

0.102 (2.59)

0.094 (2.39)

0.0192 (0.49)

0.0138 (0.35)

0.0098 (0.25)

0.0040 (0.10)

8-Lead TO-99

(TO-99)

45

°

BSC

0.115

(2.92)

BSC

0.034 (0.86)

0.027 (0.69)

0.045 (1.14)

0.027 (0.69)

0.160 (4.06)

0.110 (2.79)

0.115
(2.92)
BSC

0.230
(5.84)

BSC

6

8

5

7

1

4

2

3

REFERENCE PLANE

BASE & SEATING PLANE

0.335 (8.51)

0.305 (7.75)

0.370 (9.40)

0.335 (8.51)

0.750 (19.05)

0.500 (12.70)

0.045 (1.14)

0.010 (0.25)

0.050

(1.27)

MAX

0.040 (1.02) MAX

0.019 (0.48)

0.016 (0.41)

0.021 (0.53)

0.016 (0.41)

0.185 (4.70)

0.165 (4.19)

0.250 (6.35)
MIN

REV. D

TMP01

–16–

C00333–0–3/02 (D)

PRINTED IN U.S.A.

Revision History

Location Page

1/02—Data Sheet changed from REV. C to REV. D.

Edits to GENERAL DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

Edits to SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

Edits to WAFER TEST LIMITS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

Edits to DICE CHARACTERISTICS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

Edits to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5