Fast, Voltage-Out DC–440 MHz,
FEATURES
Multistage demodulating logarithmic amplifier
Voltage output, rise time <15 ns
High current capacity: 25 mA into grounded RL
95 dB dynamic range: −91 dBV to +4 dBV
Single supply of 2.7 V min at 8 mA typ
DC–440 MHz operation, ±0.4 dB linearity
Slope of +24 mV/dB, intercept of −108 dBV
Highly stable scaling over temperature
Fully differential dc-coupled signal path
100 ns power-up time, 1 mA sleep current
APPLICATIONS
Conversion of signal level to decibel form
Transmitter antenna power measurement
Receiver signal strength indication (RSSI)
Low cost radar and sonar signal processing
Network and spectrum analyzers
Signal-level determination down to 20 Hz
True-decibel ac mode for multimeters
GENERAL DESCRIPTION
The AD8310 is a complete, dc–440 MHz demodulating
logarithmic amplifier (log amp) with a very fast voltage mode
output, capable of driving up to 25 mA into a grounded load in
under 15 ns. It uses the progressive compression (successive
detection) technique to provide a dynamic range of up to 95 dB
to ±3 dB law conformance or 90 dB to a ±1 dB error bound up
to 100 MHz. It is extremely stable and easy to use, requiring no
significant external components. A single-supply voltage of
2.7 V to 5.5 V at 8 mA is needed, corresponding to a power
consumption of only 24 mW at 3 V. A fast-acting CMOScompatible enable pin is provided.
Each of the six cascaded amplifier/limiter cells has a smallsignal gain of 14.3 dB, with a −3 dB bandwidth of 900 MHz.
A total of nine detector cells are used to provide a dynamic
range that extends from −91 dBV (where 0 dBV is defined as
the amplitude of a 1 V rms sine wave), an amplitude of about
±40 µV, up to +4 dBV (or ±2.2 V). The demodulated output
is accurately scaled, with a log slope of 24 mV/dB and an
intercept of –108 dBV. The scaling parameters are supplyand temperature-independent.
Rev. D
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
95 dB Logarithmic Amplifier
AD8310
FUNCTIONAL BLOCK DIAGRAM
MIRROR
2µA
dB
/
2
3kΩ
AD8310
COMM
COMM
+
–
1kΩ
COMM
3kΩ
SUPPLY
+INPUT
–INPUT
COMMON
5
8
1
2
VPOS
INHI
1.0kΩ
INLO
COMM
BAND GAP REFERENCE
8mA
3
NINE DETECTOR CELLS
AND BIASING
SIX 14.3dB 900MHz
AMPLIFIER STAGES
SPACED 14.3dB
INPUT-OFFSET
COMPENSATION LOOP
Figure 1.
The fully differential input offers a moderately high impedance
(1 kΩ in parallel with about 1 pF). A simple network can match
the input to 50 Ω and provide a power sensitivity of −78 dBm to
+17 dBm. The logarithmic linearity is typically within ±0.4 dB
up to 100 MHz over the central portion of the range, but it is
somewhat greater at 440 MHz. There is no minimum frequency
limit; the AD8310 can be used down to low audio frequencies.
Special filtering features are provided to support this wide
range.
The output voltage runs from a noise-limited lower boundary
of 400 mV to an upper limit within 200 mV of the supply
voltage for light loads. The slope and intercept can be readily
altered using external resistors. The output is tolerant of a wide
variety of load conditions and is stable with capacitive loads of
100 pF.
The AD8310 provides a unique combination of low cost, small
size, low power consumption, high accuracy and stability, high
dynamic range, a frequency range encompassing audio to UHF,
fast response time, and good load-driving capabilities, making
this product useful in numerous applications that require the
reduction of a signal to its decibel equivalent.
The AD8310 is available in the industrial temperature range of
–40°C to +85°C in an 8-lead MSOP package.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
Fax: 781.326.8703 © 2004 Analog Devices, Inc. All rights reserved.
www.analog.com
ENBL
BFIN
VOUT
OFLT
33pF
ENABLE
7
BUFFER
6
INPUT
4
OUTPUT
OFFSET
3
FILTER
01084-001
AD8310
TABLE OF CONTENTS
Specifications..................................................................................... 3
dBV vs. dBm................................................................................ 15
Absolute Maximum Ratings............................................................ 4
ESD Caution.................................................................................. 4
Pin Configuration and Function Descriptions............................. 5
Typical Performance Characteristics............................................. 6
Theory of Operation ........................................................................ 9
Progressive Compression ............................................................ 9
Slope and Intercept Calibration................................................ 10
Offset Control............................................................................. 10
Product Overview........................................................................... 11
Enable Interface.......................................................................... 11
Input Interface............................................................................. 11
Offset Interface ...........................................................................12
Output Interface .........................................................................12
Using the AD8310........................................................................... 14
Basic Connections ...................................................................... 14
Input Matching ........................................................................... 15
Narrow-Band Matching ............................................................ 16
General Matching Procedure.................................................... 16
Slope and Intercept Adjustments ............................................. 17
Increasing the Slope to a Fixed Value ...................................... 17
Output Filtering.......................................................................... 18
Lowering the High-Pass Corner Frequency of the Offset
Compensation Loop
Applications..................................................................................... 19
Cable-Driving ............................................................................. 19
DC-Coupled Input..................................................................... 19
Evaluation Board ............................................................................ 20
Outline Dimensions....................................................................... 22
Ordering Guide .......................................................................... 22
.................................................................. 18
Transfer Function in Terms of Slope and Intercept............... 15
REVISION HISTORY
10/04—Data Sheet Changed from Rev. C to Rev. D
Format Updated.......................................................... Universal
Typical Performance Characteristics Reordered ......................... 6
Changes to Figures 41 and 42....................................................... 20
7/03—Data Sheet Changed from Rev. B to Rev. C
Replaced TPC 12............................................................................... 5
Change to DC-Coupled Input Section ........................................ 14
Replaced Figure 20 .........................................................................15
Updated Outline Dimensions....................................................... 16
2/03—Data Sheet Changed from Rev. A to Rev. B
Change to Evaluation Board Section ........................................... 15
Change to Table III......................................................................... 16
Updated Outline Dimensions....................................................... 16
1/00—Data Sheet Changed from Rev. 0 to Rev. A
10/99—Revision 0: Initial Version
Rev. D | Page 2 of 24
AD8310
SPECIFICATIONS
TA = 25°C, VS = 5 V, unless otherwise noted.
Table 1.
Parameter Conditions Min Typ Max Unit
INPUT STAGE Inputs INHI, INLO
Maximum Input1 Single-ended, p-p ±2.0 ±2.2 V
4 dBV
Equivalent Power in 50 Ω Termination resistor of 52.3 Ω 17 dBm
Differential drive, p-p 20 dBm
Noise Floor Terminated 50 Ω source 1.28 nV/√Hz
Equivalent Power in 50 Ω 440 MHz bandwidth −78 dBm
Input Resistance From INHI to INLO 800 1000 1200 Ω
Input Capacitance From INHI to INLO 1.4 pF
DC Bias Voltage Either input 3.2 V
LOGARITHMIC AMPLIFIER Output VOUT
±3 dB Error Dynamic Range From noise floor to maximum input 95 dB
Transfer Slope 10 MHz ≤ f ≤ 200 MHz 22 24 26 mV/dB
Overtemperature, –40°C < TA < +85°C 20 26 mV/dB
Intercept (Log Offset)
Equivalent dBm (re 50 Ω) −102 −95 −86 dBm
Overtemperature, −40°C ≤ TA ≤ +85°C −120 −96 dBV
Equivalent dBm (re 50 Ω) −107 −83 dBm
Temperature sensitivity −0.04 dB/°C
Linearity Error (Ripple) Input from –88 dBV (–75 dBm) to +2 dBV (+15 dBm) ±0.4 dB
Output Voltage Input = –91 dBV (–78 dBm) 0.4 V
Input = 9 dBV (22 dBm) 2.6 V
Minimum Load Resistance, RL 100 Ω
Maximum Sink Current 0.5 mA
Output Resistance 0.05 Ω
Video Bandwidth 25 MHz
Rise Time (10% to 90%) Input Level = −43 dBV (−30 dBm), RL ≥ 402 Ω, CL ≤ 68 pF 15 ns
Input Level = −3 dBV (+10 dBm), RL ≥ 402 Ω, CL ≤ 68 pF 20 ns
Fall Time (90% to 10%) Input Level = −43 dBV (−30 dBm), RL ≥ 402 Ω, CL ≤ 68 pF 30 ns
Input Level = −3 dBV (+10 dBm), RL ≥ 402 Ω, CL ≤ 68 pF 40 ns
Output Settling Time to 1% Input Level = −13 dBV (0 dBm), RL ≥ 402 Ω, CL ≤ 68 pF 40 ns
POWER INTERFACES
Supply Voltage, VPOS 2.7 5.5 V
Quiescent Current Zero-signal 6.5 8.0 9.5 mA
Overtemperature −40°C < TA< +85°C 5.5 8.5 10 mA
Disable Current 0.05 µA
Logic Level to Enable Power High condition, −40°C < TA< +85°C 2.3 V
Input Current when High 3 V at ENBL 35 µA
Logic Level to Disable Power Low condition, −40°C < TA< +85°C 0.8 V
1
The input level is specified in dBV, because logarithmic amplifiers respond strictly to voltage, not power. 0 dBV corresponds to a sinusoidal single-frequency input of
1 V rms. A power level of 0 dBm (1 mW) in a 50 Ω termination corresponds to an input of 0.2236 V rms. Therefore, the relationship between dBV and dBm is a fixed
offset of 13 dBm in the special case of a 50 Ω termination.
2
Guaranteed but not tested; limits are specified at six sigma levels.
2
10 MHz ≤ f ≤ 200 MHz −115 −108 −99 dBV
Rev. D | Page 3 of 24
AD8310
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Value
Supply Voltage, VS 7.5 V
Input Power (re 50 Ω), Single-Ended 18 dBm
Differential Drive 22 dBm
Internal Power Dissipation 200 mW
θ
JA
Maximum Junction Temperature 125°C
Operating Temperature Range −40°C to +85°C
Storage Temperature Range –65°C to +150°C
Lead Temperature Range (Soldering 60 s) 300°C
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
200°C/W
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may effect
device reliability.
Rev. D | Page 4 of 24
AD8310
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
INLO
COMM
OFLT
VOUT
1
AD8310
2
TOP VIEW
3
(Not to Scale)
4
8
7
6
5
INHI
ENBL
BFIN
VPOS
01084-002
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin No. Mnemonic Function
1 INLO One of Two Balanced Inputs, Biased Roughly to VPOS/2.
2 COMM Common Pin (Usually Grounded).
3 OFLT Offset Filter Access, Nominally at about 1.75 V.
4 VOUT Low Impedance Output Voltage, 25 mA Maximum Load.
5 VPOS Positive Supply, 2.7 V to 5.5 V at 8 mA Quiescent Current.
6 BFIN Buffer Input; Used to Lower Postdetection Bandwidth.
7 ENBL CMOS Compatible Chip Enable (Active when High).
8 INHI Second of Two Balanced Inputs.
Rev. D | Page 5 of 24
AD8310
TYPICAL PERFORMANCE CHARACTERISTICS
3.0
2.4
4.0
2.5
2.0
1.5
= –40°C
T
RSSI OUTPUT (V)
1.0
0.5
A
T
= +25°C
A
0
–120 20–100
(–87dBm)
T
= +85°C
A
–80 –60 –40 –20 0
INPUT LEVEL (dBV)
(+13dBm)
Figure 3. RSSI Output vs. Input Level, 100 MHz Sine Input at T
and +85°C, Single-Ended Input
3.0
2.5
2.0
1.5
RSSI OUTPUT (V)
1.0
0.5
0
–120 –100
(–87dBm)
–80 –60 –40 –20 0
INPUT LEVEL (dBV)
Figure 4. RSSI Output vs. Input Level at T
10MHz
(+13dBm)
= 25°C for Frequencies
A
of 10 MHz, 50 MHz, and 100 MHz
3.0
2.5
2.0
1.5
RSSI OUTPUT (V)
1.0
0.5
0
–120 20–100
(–87dBm)
–80 –60 –40 –20 0
INPUT LEVEL (dBV)
300MHz
440MHz
(+13dBm)
01084-011
= −40°C, +25°C,
A
50MHz
100MHz
20
01084-012
200MHz
01084-013
2.1
1.8
1.5
(V)
1.2
OUT
V
0.9
0.6
0.3
0
–90 –20–80
–40°C
+85°C
+85°C
+25°C
–70 –60 –50 –40 –30
+25°C
–40°C
–10 0 10
PIN (dBm)
3.0
2.0
1.0
0
–1.0
–2.0
–3.0
–4.0
ERROR (dB)
Figure 6. Log Linearity of RSSI Output vs. Input Level, 100 MHz Sine Input
= −40°C, +25°C, and +85°C
at T
A
5
4
3
2
1
0
–1
ERROR (dB)
–2
–3
–4
–5
–120 20–100
(–87dBm)
–80 –60 –40 –20 0
INPUT LEVEL (dBV)
Figure 7. Log Linearity of RSSI Output vs. Input Level, at T
50MHz
100MHz
(+13dBm)
10MHz
= 25°C,
A
01084-015
for Frequencies of 10 MHz, 50 MHz, and 100 MHz
5
4
3
2
1
0
–1
ERROR (dB)
–2
–3
–4
–5
–120 20–100
(–87dBm)
–80 –60 –40 –20 0
INPUT LEVEL (dBV)
300MHz
440MHz
200MHz
(+13dBm)
01084-016
01084-014
Figure 5. RSSI Output vs. Input Level at T
of 200 MHz, 300 MHz, and 440 MHz
= 25°C for Frequencies
A
Rev. D | Page 6 of 24
Figure 8. Log Linearity of RSSI Output vs. Input Level at T
for Frequencies of 200 MHz, 300 MHz, and 440 MHz
= 25°C
A
AD8310
100pF
3300pF
50µs PER
HORIZONTAL
DIVISION
V
OUT
0.01µF
GROUND REFERENCE
Figure 9. Small-Signal AC Response of RSSI Output with External BFIN
Capacitance of 100 pF, 3300 pF, and 0.01 µF
500mV PER
VERTICAL
DIVISION
01084-009
500mV PER
VERTICAL
DIVISION
GROUND REFERENCE
10mV PER
VERTICAL
DIVISION
INPUT
Figure 12. Small-Signal RSSI Pulse Response
= 402 Ω and CL = 68 pF
with R
L
V
OUT
25ns PER
HORIZONTAL
DIVISION
01084-010
V
OUT
100Ω
500mV PER
VERTICAL
DIVISION
GND REFERENCE
INPUT
500mV PER
VERTICAL
DIVISION
154Ω
Figure 10. Large-Signal RSSI Pulse Response with C
and R
= 100 Ω, 154 Ω, and 200 Ω
L
V
OUT
500mV PER
VERTICAL
DIVISION
GND REFERENCE
INPUT
–3dBV INPUT
LEVEL SHOWN
HERE
Figure 11. RSSI Pulse Response with R
= 402 Ω and CL = 68 pF, for Inputs
L
200Ω
100ns PER
HORIZONTAL
DIVISION
L
100ns PER
HORIZONTAL
DIVISION
500mV PER
VERTICAL
DIVISION
Stepped from 0 dBV to −33 dBV, −23 dBV, −13 dBV, and −3 dBV
= 100 pF
01084-005
01084-006
V
OUT
500mV PER
VERTICAL
DIVISION
GND REFERENCE
INPUT
500mV PER
VERTICAL
DIVISION
CURVES
OVERLAP
100ns PER
HORIZONTAL
DIVISION
Figure 13. Large-Signal RSSI Pulse Response with R
and C
= 33 pF, 68 pF, and 100 pF
L
V
OUT
200mV PER
VERTICAL
DIVISION
GND REFERENCE
INPUT
100ns PER
HORIZONTAL
DIVISION
20mV PER
VERTICAL
DIVISION
Figure 14. Small-Signal RSSI Pulse Response with R
and Back Termination of 50 Ω (Total Load = 100 Ω)
= 100 Ω
L
= 50 Ω
L
01084-007
01084-008
Rev. D | Page 7 of 24
AD8310
100
10
1
0.1
0.01
0.001
SUPPLY CURRENT (mA)
T
= +85°C
A
T
A
= +25°C
V
OUT
500mV PER
VERTICAL
DIVISION
–3dBV
–23dBV
–43dBV
–63dBV
–83dBV
0.0001
0.00001
0.5 2.50.7
= –40°C
T
A
0.9 1.1 1.3 1.5 1.7 1.9 2.1 2.3
ENABLE VOLTAGE (V)
Figure 15. Supply Current vs. Enable Voltage at T
30
29
28
27
26
25
24
RSSI SLOPE (mV/dB)
23
22
21
20
1 100010
FREQUENCY (MHz)
Figure 16. RSSI S lope vs. Frequency
40
35
30
25
20
COUNT
15
10
5
0
21.5 22.0
NORMAL
(23.6584,
0.308728)
22.5 23.0 23.5 24.0 24.5
SLOPE (mV/dB)
= −40°C, +25°C, and +85°C
A
100
01084-019
01084-003
01084-017
5V PER
VERTICAL
DIVISION
200ns PER HORIZONTAL DIVISION
ENABLE
01084-004
Figure 18. Power-On/Off Response Time with RF Input of −83 dBV to −3 dBV
–99
–101
–103
–105
–107
–109
–111
–113
RSSI INTERCEPT (dBV)
–115
–117
–119
1 100010
FREQUENCY (MHz)
100
01084-018
Figure 19. RSSI Intercept vs. Frequency
24
22
20
18
16
14
12
COUNT
10
8
6
4
2
0
–115 –113
–111 –109 –107 –105 –103 –101 –99 –97
INTERCEPT (dBV)
NORMAL
(–107.6338,
2.36064)
01084-020
Figure 17. Transfer Slope Distribution, V
= 5 V, Frequency = 100 MHz, 25°C
S
Rev. D | Page 8 of 24
Figure 20. Intercept Distribution V
= 5 V, Frequency = 100 MHz, 25°C
S
AD8310
(
)
=
THEORY OF OPERATION
Logarithmic amplifiers perform a more complex operation than
classical linear amplifiers, and their circuitry is significantly
different. A good grasp of what log amps do and how they do it
can help users avoid many pitfalls in their applications. For a
complete discussion of the theory, see the AD8307 data sheet.
The essential purpose of a log amp is not to amplify (though
amplification is needed internally), but to compress a signal of
wide dynamic range to its decibel equivalent. It is, therefore, a
measurement device. An even better term might be logarithmic
converter, because the function is to convert a signal from one
domain of representation to another via a precise nonlinear
transformation:
⎛
⎞
V
IN
⎜
=
VV log (1)
OUT
Y
⎟
⎜
⎟
V
X
⎝
⎠
where:
V
is the output voltage.
OUT
V
is the slope voltage. The logarithm is usually taken to
Y
base ten, in which case
V
is the input voltage.
IN
V
is the intercept voltage.
X
Log amps implicitly require two references (here
V
is also the volts-per-decade.
Y
V
and VY)
X
that determine the scaling of the circuit. The accuracy of a log
amp cannot be any better than the accuracy of its scaling
reference
s. In the AD8310, these are provided by a band gap
reference.
V
OUT
5V
Y
4V
V
OUT
Y
3V
Y
2V
Y
V
Y
=0
VIN=10–2V
–40dBc
–2V
Y
LOWER INTERCEPT
VIN =V
X
Figure 21. General Form of the Logarithmic Function
0dBc
X
V
V
IN
+40dBc
SHIFT
=102V
LOG V
IN
=104V
V
IN
+80dBc
X
X
While Equation 1, plotted in Figure 21, is fundamentally correct,
a different formula is appropriate for specifying the calibration
attributes or demodulating log amps like the AD8310, operating
in RF applications with a sine wave input.
01084-021
SLOPE
e demodulated and filtered baseband (video or RSSI)
where:
V
is th
OUT
OUT
output.
V
is the logarithmic slope, now expressed in V/dB
SLOPE
(25 mV/dB for the AD8310).
P
is the input power, expressed in dB relative to some
IN
reference power level.
P
is the logarithmic intercept, expressed in dB relative t
O
same reference level.
A widely used referenc
50 Ω, a level of 0 dBm. Note that the quantity (
The logarithmic function disappears from the formula, becaus
the conversion has already been implicitly performed in stating
the input in decibels. This is strictly a concession to popular
convention. Log amps manifestly do not respond to power
(tacitly, power absorbed at the input), but rather to input
voltage. The input is specified in dBV (decibels with respect t
1 V rms) throughout this data sheet. This is more precise,
although still incomplete, because the signal waveform is also
involved. Many users specify RF signals in terms of power
(usually in dBm/50 Ω) and this convention is used in this data
sheet when specifying the performance of the AD8310.
PROGRESSIVE COMPRESSION
High speed, high dynamic-range log am
nonlinear amplifier cells to generate the logarithmic functio
a series of contiguous segments, a type of piecewise linear
technique. The AD8310 employs six cells in its main signal path,
each having a small-signal gain of 14.3 dB (×5.2) and a −3 dB
bandwidth of about 900 MHz. The overall gain is about 20,000
(86 dB) and the overall bandwidth of the chain is approximatel
500 MHz, resulting in a gain-bandwidth product (GBW) of
10,000 GHz, about a million times that of a typical op amp. This
very high GBW is essential to accurate operation under smallsignal conditions and at high frequencies. The AD8310 exhibits
a logarithmic response down to inputs as small as 40 µV
at 440 MHz.
Progressive co
video response or accept an RF input and demodulate this
signal to develop an output that is essentially the envelope of t
input represented on a logarithmic or decibel scale. The
AD8310 is the latter kind. Demodulation is performed in a total
of nine detector cells. Six are associated with the amplifier
stages, and three are passive detectors that receive a progressively attenuated fraction of the full input. The maximum sig
frequency can be 440 MHz, but, because all the gain stages ar
dc-coupled, operation at very low frequencies is possible.
mpression log amps either provide a baseband
PPVV −
IN
(2)
O
o the
e in RF systems is dB above 1 mW in
P – P ) is dB.
IN O
ps use a cascade of
e
o
n as
he
nal
e
y
Rev. D | Page 9 of 24
AD8310
SLOPE AND INTERCEPT CALIBRATION
All monolithic log amps from Analog Devices use p
design techniques to control the logarithmic slope and
intercept. The primary source of this calibration is a pair of
accurate voltage references that provide supply- and
temperature-independent scaling. The slope is set to 24 mV/dB
by the bias chosen for the detector cells and the subsequent gain
of the postdetector output interface. With this slope, the full
95 dB dynamic range can be easily accommodated within the
output swing capacity, when operating from a 2.7 V supply.
Intercept positioning at−108 dBV (−95 dBm re 50
likewise been chosen to provide an output centered in the
available voltage range.
Precise control of the slo
with stable scaling parameters, making it a true measurement
device as, for example, a calibrated received signal strength
indicator (RSSI). In this application, the input waveform is
invariably sinusoidal. The input level is correctly specified in
dBV. It can alternatively be stated as an equivalent power, in
dBm, but in this case, it is necessary to specify the impedance
which this power is presumed to be measured. In RF practice, it
is common to assume a reference impedance of 50 Ω, in which
0 dBm (1 mW) corresponds to a sinusoidal amplitude of
316.2 mV (223.6 mV rms). However, the power metric is correct
only when the input impedance is lowered to 50 Ω, either by a
termination resistor added across INHI and INLO, or by the use
of a narrow-band matching network.
Note that log amps do not inherently r
the voltage applied to their input. The AD8310 presents a
nominal input impedance much higher than 50 Ω (typically
1 kΩ at low frequencies). A simple input matching network c
considerably improve the power sensitivity of this type of log
amp. This increases the voltage applied to the input and,
therefore, alters the intercept. For a 50 Ω reactive match, the
voltage gain is about 4.8, and the whole dynamic range move
down by 13.6 dB. The effective intercept is a function of
waveform. For example, a square-wave input reads 6 dB higher
than a sine wave of the same amplitude, and a Gaussian noise
input reads 0.5 dB higher than a sine wave of the same rms
value.
pe and intercept results in a log amp
espond to power, but to
recision
Ω) has
in
an
s
OFFSET CONTROL
In a monolithic log amp, direct coupling is used between the
stages for several reasons. First, it avoids the need for coupling
capacitors, which typically have a chip area at least as large as
that of a basic gain cell, considerably increasing die size. Second,
the capacitor values predetermine the lowest frequency at which
the log amp can operate. For moderate values, this can be as
high as 30 MHz, limiting the application range. Third, the
parasitic back-plate capacitance lowers the bandwidth of the
cell, further limiting the scope of applications.
However, the very high dc gain of a direct-coupled amplifier
raises a practical issue. An offset voltage in the early stages of
the chain is indistinguishable from a real signal. If it were as
high as 400 µV, it would be 18 dB larger than the smallest ac
signal (50 µV), potentially reducing the dynamic range by this
amount. This problem can be averted by using a global feedback
path from the last stage to the first, which corrects this offset in
a similar fashion to the dc negative feedback applied around an
op amp. The high frequency components of the feedback signal
must, of course, be removed to prevent a reduction of the HF
gain in the forward path.
An on-chip filter capacitor of 33 pF provides sufficient suppression of HF feedback to allow operation above 1 MHz. The
−3 dB point in the high-pass response is at 2 MHz, but the
usable range extends well below this frequency. To further lower
the frequency range, an external capacitor can be added at
OFLT (Pin 3). For example, 300 pF lowers it by a factor of 10.
Operation at low audio frequencies requires a capacitor of about
1 µF. Note that this filter has no effect for input levels well above
the offset voltage, where the frequency range would extend
down to dc (for a signal applied directly to the input pins). The
dc offset can optionally be nulled by adjusting the voltage on
the OFLT pin (see the Applications section).
Rev. D | Page 10 of 24
AD8310
PRODUCT OVERVIEW
The AD8310 has six main amplifier/limiter stages. These six
cells and their and associated
g
styled full-wave detectors
m
handle the lower two-thirds of the dynamic range. Three topend detectors, placed at 14.3 dB taps on a passive attenuator,
handle the upper third of the 95 dB range. The first amplifier
stage provides a low noise spectral density (1.28 nV/√Hz).
Biasing for these cells is provided by two references: one
determines their gain, and the other is a band gap circuit that
determines the logarithmic slope and stabilizes it against supply
and temperature variations. The AD8310 can be enabled or
disabled by a CMOS-compatible level at ENBL (Pin 7).
The differential current-mode outputs of the nine detectors are
summed and then converted to single-sided form, nominally
scaled 2 µA/dB. The output voltage is developed by applying
this current to a 3 kΩ load resistor followed by a high speed
gain-of-four buffer amplifier, resulting in a logarithmic slope of
24 mV/dB (480 mV/decade) at VOUT (Pin 4). The unbuffered
voltage can be accessed at BFIN (Pin 6), allowing certain
functional modifications such as the addition of an external
postdemodulation filter capacitor and the alteration or
adjustment of slope and intercept.
MIRROR
2
2
/dB
µ
A
COMM
AD8310
Ω
3k
1k
COMM
+
–
3k
Ω
COMM
ENBL
ENABLE
BFIN
BUFFER
INPUT
VOUT
OUTPUT
Ω
OFLT
OFFSET
FILTER
33pF
SUPPLY
+INPUT
–INPUT
COMMON
VPOS
INHI
1.0k
INLO
COMM
BAND GAP REFERENCE
8mA
Ω
3
AND BIASING
SIX 14.3dB 900MHz
AMPLIFIER STAGES
NINE DETECTOR CELLS
SPACED 14.3dB
INPUT-OFFSET
COMPENSATION LOOP
Figure 22. Main Features of the AD8310
The last gain stage also includes an offset-sensing cell. This
generates a bipolarity output current, if the main signal path
exhibits an imbalance due to accumulated dc offsets. This
current is integrated by an on-chip capacitor that can be
increased in value by an off-chip component at OFLT (Pin 3).
The resulting voltage is used to null the offset at the output of
the first stage. Because it does not involve the signal input
connections, whose ac-coupling capacitors otherwise introduce
a second pole into the feedback path, the stability of the offset
correction loop is assured.
The AD8310 is built on an advanced, dielectrically isolated,
complementary bipolar process. In the following interface
diagrams, resistors labeled as R are thin-film resistors that have
a low temperature coefficient of resistance (TCR) and high
linearity under large-signal conditions. Their absolute tolerance
is typically within ±20%. Similarly, capacitors labeled as C have
01084-022
a typical tolerance of ±15% and essentially zero temperature or
voltage sensitivity. Most interfaces have additional small
junction capacitances associated with them, due to active
devices or ESD protection, which might not be accurate or
stable. Component numbering in these interface diagrams is
local.
ENABLE INTERFACE
The chip-enable interface is shown in Figure 23. The currents in
the diode-connected transistors control the turn-on and turnoff states of the band gap reference and the bias generator. They
are a maximum of 100 µA when ENBL is taken to 5 V under
worst-case conditions. For voltages below 1 V, the AD8310 is
disabled and consumes a sleep current of under 1 µA. When
tied to the supply or a voltage above 2 V, it is fully enabled. The
internal bias circuitry is very fast (typically <100 ns for either
off or on). In practice, however, the latency period before the log
amp exhibits its full dynamic range is more likely to be limited
by factors relating to the use of ac-coupling at the input or the
settling of the offset-control loop (see the following sections).
AD8310
TO BIAS
STAGES
01084-023
ENBL
40kΩ
7
2
COMM
Figure 23. Enable Interface
INPUT INTERFACE
Figure 24 shows the essentials of the input interface. CP and CM
are parasitic capacitances, and C
capacitance, largely due to Q1 and Q2. In most applications,
both input pins are ac-coupled. The S switches close when
enable is asserted. When disabled, bias current I
the inputs float; therefore, the coupling capacitors remain
charged. If the log amp is disabled for long periods, small
leakage currents discharge these capacitors. Then, if they are
poorly matched, charging currents at power-up can generate a
transient input voltage that can block the lower reaches of the
dynamic range until it becomes much less than the signal.
A single-sided signal can be applied via a blocking capacitor to
either Pin 1 or Pin 8, with the other pin ac-coupled to ground.
Under these conditions, the largest input signal that can be
handled is 0 dBV (a sine amplitude of 1.4 V) when using a 3 V
supply; a 5 dBV input (2.5 V amplitude) can be handled with a
5 V supply. When using a fully balanced drive, this maximum
input level is permissible for supply voltages as low as 2.7 V.
Above 10 MHz, this is easily achieved using an LC matching
network. Such a network, having an inductor at the input,
usefully eliminates the input transient noted above.
is the differential input
D
is shut off and
E
Rev. D | Page 11 of 24
AD8310
VPOS
5
S
6kΩ
6kΩ
S
2
COMM
INHI
INLO
COM
COM
4kΩ
2kΩ
TOP-END
DETECTORS
TYP 2.2V FOR
3V SUPPLY,
3.2V AT 5V
C
P
8
C
D
1
C
M
Figure 24. Signal Input Interface
Occasionally, it might be desirable to use the dc-coupled
potential of the AD8310 in baseband applications. The main
challenge here is to present the signal at the elevated commonmode input level, which might require the use of low noise, low
offset buffer amplifiers. In some cases, it might be possible to
use dual supplies of ±3 V, which allow the input pins to operate
at ground potential. The output, which is internally referenced
to the COMM pin (now at −3 V), can be positioned back to
ground level, with essentially no sensitivity to the particular
value of the negative supply.
OFFSET INTERFACE
The input-referred dc offsets in the signal path are nulled via
the interface associated with Pin 3, shown in Figure 25. Q1 and
Q2 are the first-stage input transistors, having slightly unbalanced load resistors, resulting in a deliberate offset voltage of
about 1.5 mV referred to the input pins. Q3 generates a small
current to null this error, dependent on the voltage at the OFLT
pin. When Q1 and Q2 are perfectly matched, this voltage is
about 1.75 V. In practice, it can range from approximately 1 V to
2.5 V for an input-referred offset of ±1.5 mV.
~3kΩ
Q1
125Ω
Q2
I
E
2.4mA
01084-024
VPOS
g
m
33pF
5
TO LAST
DETECTOR
AVERAGE
ERROR
CURRENT
2
COMM
INPUT
STAGE
BIAS,
1.2V
Q1
125Ω
MAIN GAIN
STAGES
16µA AT
Q2
BALANCE
Q3
36kΩ
Q4
48kΩ
OFLT
3
C
S
OFLT
Figure 25. Offset Interface and Offset-Nulling Path
In normal operation using an ac-coupled input signal, the OFLT
pin should be left unconnected. The
g
cell, which is gated off
m
when the chip is disabled, converts a residual offset (sensed at a
point near the end of the cascade of amplifiers) to a current.
This is integrated by the on-chip capacitor, C
external capacitance, C
, to generate the voltage that is
OFLT
, plus any added
HP
applied back to the input stage in the polarity needed to null the
output offset. From a small-signal perspective, this feedback
alters the response of the amplifier, which exhibits a zero in its
ac transfer function, resulting in a closed-loop high-pass −3 dB
corner at about 2 MHz. An external capacitor lowers the highpass corner to arbitrarily low frequencies; using 1 µF, the 3 dB
corner is at 60 Hz.
OUTPUT INTERFACE
The nine detectors generate differential currents, having an
average value that is dependent on the signal input level, plus a
fluctuation at twice the input frequency. These are summed at
nodes LGP and LGN in Figure 26. Further currents are added at
these nodes to position the intercept by slightly raising the
output for zero input and to provide temperature compensation.
01084-025
VPOS
FROM ALL
DETECTORS
COMM
5
LGP
LGN
BIAS
60µA
2
0.4pF
1.25kΩ1.25kΩ
0.4pF1.25kΩ1.25kΩ
2µA/dB
R1
3kΩ
0.2pF
BIAS
4kΩ4kΩ
BFIN
6
3kΩ
1kΩ
4
VOUT
01084-026
Figure 26. Simplified Output Interface
Rev. D | Page 12 of 24
AD8310
For zero-signal conditions, all the detector output currents are
equal. For a finite input of either polarity, their difference is
converted by the output interface to a single-sided unipolar
current, nominally scaled 2 µA/dB (40 µA/decade), at the
output pin BFIN. An on-chip resistor, R1, of ~3 kΩ converts this
current to a voltage of 6 mV/dB. This is then amplified by a
factor of 4 in the output buffer, which can drive a current of up
to 25 mA in a grounded load resistor. The overall rise time of
the AD8310 is under 15 ns. There is also a delay time of about
6 ns when the log amp is driven by an RF burst, starting at zero
amplitude.
When driving capacitive loads, it is desirable to add a low value
of load resistor to speed up the return to the baseline; the buffer
is stable for loads of a least 100 pF. The output bandwidth can
be lowered by adding a grounded capacitor at BFIN. The timeconstant of the resulting single-pole filter is formed with the
3 kΩ internal load resistor (with a tolerance of 20%). Therefore,
to set the –3 dB frequency to 20 kHz, use a capacitor of 2.7 nF.
Using 2.7 µF, the filter corner is at 20 Hz.
Rev. D | Page 13 of 24
AD8310
R
USING THE AD8310
The AD8310 has very high gain and bandwidth. Consequently,
it is susceptible to all signals that appear at the input terminals
within a very broad frequency range. Without the benefit of
filtering, these are indistinguishable from the desired signal and
have the effect of raising the apparent noise floor (that is,
lowering the useful dynamic range). For example, while the
signal of interest has an IF of 50 MHz, any of the following can
easily be larger than the IF signal at the lower extremities of its
dynamic range: a few hundred mV of 60 Hz hum picked up due
to poor grounding techniques, spurious coupling from a digital
clock source on the same PC board, local radio stations, and so
on. Careful shielding and supply decoupling is, therefore,
essential. A ground plane should be used to provide a low
impedance connection to the common pin COMM, for the
decoupling capacitor(s) used at VPOS, and for the output
ground.
BASIC CONNECTIONS
Figure 27 shows the connections needed for most applications.
A supply voltage between 2.7 V and 5.5 V is applied to VPOS
and is decoupled using a 0.01 µF capacitor close to the pin.
Optionally, a small series resistor can be placed in the power
line to give additional filtering of power-supply noise. The
ENBL input, which has a threshold of approximately 1.3 V (see
Figure 21), should be tied to VPOS when this feature is not
needed.
4.7Ω
OPTIONAL
NC
NC
SIGNAL
INPUT
52.3Ω
C2
0.01µF
8765
INHI ENBL BFIN VPOS
INLO COMM OFLT VOUT
C1
0.01µF
1234
NC = NO CONNECT
AD8310
Figure 27. Basic Connections
While the AD8310’s input can be driven differentially, the input
signal is, in general, single-ended. C1 is tied to ground, and the
input signal is coupled in through C2. Capacitors C1 and C2
should have the same value to minimize start-up transients
when the enable feature is used; otherwise, their values need not
be equal.
C4
0.01µF
V
S
(2.7V–5.5V)
V
(RSSI)
OUT
01084-027
The coupling time constant, 50 × C
with a 3 dB attenuation at f
= C
. In high frequency applications, fHP should be as large as
C
HP
/2, forms a high-pass corner
C
= 1/( π × 50 × CC ), where C1 = C2
possible to minimize the coupling of unwanted low frequency
signals. In low frequency applications, a simple RC network
forming a low-pass filter should be added at the input for
similar reasons. This should generally be placed at the generator
side of the coupling capacitors, thereby lowering the required
capacitance value for a given high-pass corner frequency.
For applications in which the ground plane might not be an
equipotential (possibly due to noise in the ground plane), the
low input of an unbalanced source should generally be
ac-coupled through a separate connection of the low associated
with the source. Furthermore, it is good practice in such
situations to break the ground loop by inserting a small
resistance to ground in the low side of the input connector (see
Figure 28).
4.7Ω
SIGNAL
INPUT
GENERATO
COMMON
4.7Ω
C2
0.01µF
8765
INHI ENBL BFIN VPOS
52.3Ω
INLO COMM OFLT VOUT
C1
1234
0.01µF
BOARD-LEVEL
GROUND
OPTIONAL
NC
AD8310
NC
NC = NO CONNECT
C4
0.01µF
V
S
(2.7V–5.5V)
V
(RSSI)
OUT
01084-028
Figure 28. Connections for Isolation of Source Ground from Device Ground
Figure 29 shows the output vs. the input level for sine inputs at
10 MHz, 50 MHz, and 100 MHz. Figure 30 shows the logarithmic conformance under the same conditions.
3.0
2.5
2.0
1.5
OUTPUT (V)
1.0
0.5
10MHz
50MHz
100MHz
The 52.3 Ω resistor combines with the 1.1 kΩ input impedance
of the AD8310 to yield a simple broadband 50 Ω input match.
An input matching network can also be used (see the Input
Matching section).
Rev. D | Page 14 of 24
0
–120 –100
(–87dBm)
INTERCEPT
–80 –60 –40 –20 0
INPUT LEVEL (dBV)
(+13dBm)
20
Figure 29. Output vs. Input Level at 10 MHz, 50 MHz, and 100 MHz
01084-029
AD8310
5
4
3
2
1
0
–1
ERROR (dB)
–2
–3
–4
–5
–120 20–100
(–87dBm)
Figure 30. Log Conformance Errors vs. Input Level at 10 MHz,
±3dB DYNAMIC RANGE
±1dB DYNAMIC RANGE
50MHz
100MHz
–80 –60 –40 –20 0
INPUT LEVEL (dBV)
50 MHz, and 100 MHz
(+13dBm)
10MHz
01084-030
TRANSFER FUNCTION IN TERMS OF SLOPE AND INTERCEPT
The transfer function of the AD8310 is characterized in terms
of its slope and intercept. The logarithmic slope is defined as the
change in the RSSI output voltage for a 1 dB change at the input.
For the AD8310, slope is nominally 24 mV/dB. Therefore, a
10 dB change at the input results in a change at the output of
approximately 240 mV. The plot of log conformance shows the
range over which the device maintains its constant slope. The
dynamic range of the log amp is defined as the range over
which the slope remains within a certain error band, usually
±1 dB or ±3 dB. In Figure 30, for example, the ±1 dB dynamic
range is approximately 95 dB (from +4 dBV to −91 dBV).
The intercept is the point at which the extrapolated linear
response would intersect the horizontal axis (see Figure 29).
For the AD8310, the intercept is calibrated to be −108 dBV
(−95 dBm). Using the slope and intercept, the output voltage
can be calculated for any input level within the specified input
range using the following equation:
V
= V
OUT
where:
V
is the demodulated and filtered RSSI output.
OUT
V
is the logarithmic slope expressed in V/dB.
SLOPE
P
is the input signal expressed in dB relative to some reference
IN
level (either dBm or dBV in this case).
P
is the logarithmic intercept expressed in dB relative to the
O
same reference level.
× (PIN − PO)
SLOPE
dBV vs. dBm
The most widely used convention in RF systems is to specify
power in dBm, decibels above 1 mW in 50 Ω. Specification of
the log amp input level in terms of power is strictly a concession
to popular convention. As mentioned previously, log amps do
not respond to power (power absorbed at the input), but to the
input voltage. The use of dBV, defined as decibels with respect
to a 1 V rms sine wave, is more precise. However, this is still
ambiguous, because waveform is also involved in the response
of a log amp, which, for a complex input such as a CDMA
signal, does not follow the rms value exactly. Because most users
specify RF signals in terms of power (more specifically, in
dBm/50 Ω) both dBV and dBm are used to specify the performance of the AD8310, showing equivalent dBm levels for the
special case of a 50 Ω environment. Values in dBV are converted
to dBm re 50 Ω by adding 13 dB.
Table 4. Correction for Signals with Differing Crest Factors
Signal Type Correction Factor1 (dB)
Sine wave 0
Square wave or dc
−3.01
Triangular wave 0.9
GSM channel (all time slots on) 0.55
CDMA channel (forward link, nine
channels on) 3.55
CDMA channel (reverse link) 0.5
PDC channel (all time slots on) 0.58
__________________________________________
1
Add to the measured input level.
INPUT MATCHING
Where higher sensitivity is required, an input matching network
is useful. Using a transformer to achieve the impedance transformation also eliminates the need for coupling capacitors,
lowers the offset voltage generated directly at the input, and
balances the drive amplitude to INLO and INHI.
The choice of turns ratio depends somewhat on the frequency.
At frequencies below 50 MHz, the reactance of the input
capacitance is much higher than the real part of the input
impedance. In this frequency range, a turns ratio of about 1:4.8
lowers the input impedance to 50 Ω, while raising the input
voltage lowers the effect of the short-circuit noise voltage by the
same factor. The intercept is also lowered by the turns ratio; for
a 50 Ω match, it is reduced by 20 log
noise is reduced by a somewhat smaller factor, because there is a
small contribution from the input noise current.
(4.8) or 13.6 dB. The total
10
For example, for an input level of −33 dBV (−20 dBm), the
output voltage is
V
= 0.024 V/dB × (−33 dBV − (−108 dBV)) = 1.8 V
OUT
Rev. D | Page 15 of 24
AD8310
NARROW-BAND MATCHING
Transformer coupling is useful in broadband applications.
However, a magnetically coupled transformer might not be
convenient in some situations. Table 5 lists narrow-band
matching values.
Table 5. Narrow-Band Matching Values
FC
(MHz)
Z
IN
(Ω)
C1
(pF)
10 45 160 150 3300 13.3
20 44 82 75 1600 13.4
50 46 30 27 680 13.4
100 50 15 13 270 13.4
150 57 10 8.2 220 13.2
200 57 7.5 6.8 150 12.8
250 50 6.2 5.6 100 12.3
500 54 3.9 3.3 39 10.9
10 103 100 91 5600 10.4
20 102 51 43 2700 10.4
50 99 22 18 1000 10.6
100 98 11 9.1 430 10.5
150 101 7.5 6.2 260 10.3
200 95 5.6 4.7 180 10.3
250 92 4.3 3.9 130 9.9
500 114 2.2 2.0 47 6.8
At high frequencies, it is often preferable to use a narrow-band
matching network, as shown in Figure 31. This has several
advantages. The same voltage gain is achieved, providing
increased sensitivity, but a measure of selectivity is also
introduced. The component count is low: two capacitors and an
inexpensive chip inductor. Additionally, by making these
capacitors unequal, the amplitudes at INP and INM can be
equalized when driving from a single-sided source, that is, the
network also serves as a balun. Figure 32 shows the response for
a center frequency of 100 MHz; note the very high attenuation
at low frequencies. The high frequency attenuation is due to the
input capacitance of the log amp.
SIGNAL
INPUT
Figure 31. Reactive Matching Network
C1
C2
C2
(pF)
L
M
L
M
(nH)
8
INHI
AD8310
INLO
1
Voltage Gain
(dB)
01084-031
14
13
12
11
10
9
8
7
6
DECIBELS
5
4
3
2
1
0
–1
60 15080
70 90 120 140
Figure 32. Response of 100 MHz Matching Network
100 110 130
FREQUENCY (MHz)
GAIN
INPUT
01084-032
GENERAL MATCHING PROCEDURE
For other center frequencies and source impedances, the
following steps can be used to calculate the basic matching
parameters.
Step 1: Tune Out C
At a center frequency, fC, the shunt impedance of the input
capacitance,
CIN, can be made to disappear by resonating with a
temporary inductor,
L
=
IN
ω
where C
= 1.4 pF. For example, at fC = 100 MHz, LIN = 1.8 µH.
IN
Step 2: Calculate CO and L
Now, having a purely resistive input impedance, calculate the
nominal coupling elements, C
C
=
O
π
2
For the AD8310, R
needed, at f
= 100 MHz, CO must be 7.12 pF and LO must be
C
356 nH.
Step 3: Split CO into Two Parts
To provide the desired fully balanced form of the network
shown in Figure 31, two capacitors C1 and C2, each of
nominally twice C
14.24 pF in this example. Under these conditions, the voltage
amplitudes at INHI and INLO are similar. A somewhat better
balance in the two drives can be achieved when C1 is made
slightly larger than C2, which also allows a wider range of
choices in selecting from standard values.
IN
L
, whose value is given by
IN
1
2
C
IN
O
and LO, using
O
()
1
RRf
C
is 1 kΩ. Therefore, if a match to 50 Ω is
IN
, can be used. This requires a value of
O
L
=
;
O
MIN
RR
MIN
f
π
2
C
For example, capacitors of C1 = 15 pF and C2 = 13 pF can be
used, making C
Rev. D | Page 16 of 24
= 6.96 pF.
O
AD8310
M
Ω
Step 4: Calculate L
M
The matching inductor required to provide both LIN and LO is
the parallel combination of these.
LL
IN
O
LL
+
O
With L
=
L
M
()
IN
= 1.8 µH and LO = 356 nH, the value of LM to complete
IN
this example of a match of 50 Ω at 100 MHz is 297.2 nH.
The nearest standard value of 270 nH can be used with only a
slight loss of matching accuracy. The voltage gain at resonance
depends only on the ratio of impedances, as given by
=
GAIN log10log20
⎛
⎜
⎜
⎝
⎞
R
IN
⎟
=
⎟
R
S
⎠
⎞
⎛
R
IN
⎟
⎜
⎟
⎜
R
S
⎠
⎝
SLOPE AND INTERCEPT ADJUSTMENTS
Where system (that is, software) calibration is not available, the
adjustments shown in Figure 33 can be used, either singly or in
combination, to trim the absolute accuracy of the AD8310.
The log slope can be raised or lowered by VR1; the values
shown provide a calibration range of ±10% (22.6 mV/dB to
27.4 mV/dB), which includes full allowance for the variability in
the value of the internal resistances. The adjustment can be
made by alternately applying two fixed input levels, provided by
an accurate signal generator, spaced over the central portion of
the dynamic range, for example, −60 dBV and –20 dBV.
Alternatively, an AM-modulated signal at about the center of
the dynamic range can be used. For a modulation depth M,
expressed as a fraction, the decibel range between the peaks and
troughs over one cycle of the modulation period is given by
M++
1
=∆
log20dB
10
For example., using a generator output of −40 dBm with a 70%
modulation depth (M = 0.7), the decibel range is 15 dB, because
the signal varies from −47.5 dBm to −32.5 dBm.
The log intercept is adjustable by VR2 over a −3 dB range with
the component values shown. VR2 is adjusted while applying an
accurately known CW signal, preferably near the lower end of
the dynamic range, to minimize the effect of any residual
uncertainty in the slope. For example, to position the intercept
to −80 dBm, a test level of −65 dBm can be applied, and VR2
adjusted to produce a dc output of 15 dB above 0 at 24 mV/dB,
which is 360 mV.
(3)
1
+V
(2.7V–5.5V)
VR2
100kΩ
(RSSI)
OUT
S
SIGNAL
INPUT
0.01µF
C2
0.01µF
52.3Ω
0.01µF
NC = NO CONNECT
8765
INHI ENBL BFIN VPOS
AD8310
INLO COMM OFLT VOUT
C1
1234
Figure 33. Slope and Intercept Adjustments
NC
10kΩ
VR1
10kΩ
4.7Ω
R
S
= 3V, RS = 500kΩ
FOR V
POS
FOR V
= 5V, RS = 850kΩ
POS
25kΩ
V
24mV/dB ±10%
INCREASING THE SLOPE TO A FIXED VALUE
It is also possible to increase the slope to a new fixed value and,
therefore, to increase the change in output for each decibel of
input change. A common example of this is the need to map the
output swing of the AD8310 into the input range of an analogto-digital converter (ADC) with a rail-to-rail input swing.
Alternatively, a situation might arise when only a part of the
total dynamic range is required—say, just 20 dB—in an
application where the nominal input level is more tightly
constrained and a higher sensitivity to a change in this level is
required. Of course, the maximum output is limited either by
the load resistance and the maximum output current rating of
25 mA or by the supply voltage (see the Specifications section).
The slope can easily be raised by adding a resistor from VOUT
to BFIN, as shown in Figure 34. This alters the gain of the
output buffer, by means of stable positive feedback, from its
normal value of 4 to an effective value that can be as high as 16,
corresponding to a slope of 100 mV/dB.
SIGNAL
INPUT
52.3Ω
The resistor, R
R
SLOPE
0.01µF
C2
0.01µF
8765
INHI ENBL BFIN VPOS
AD8310
INLO COMM OFLT VOUT
C1
0.01µF
=
1234
Figure 34. Raising the Slope to 100 mV/dB
, is set according to the equation
SLOPE
NC
NC = NO CONNECT
k22.9
mV/dB24
1
+
Slope
4.7Ω
R
SLOPE
12.1kΩ
V
S
(2.7V–5.5V)
V
100mV/dB
OUT
01084-033
01084-034
Rev. D | Page 17 of 24
AD8310
OUTPUT FILTERING
For applications in which maximum video bandwidth and,
consequently, fast rise time are desired, it is essential that the
BFIN pin be left unconnected and free of any stray capacitance.
The nominal output video bandwidth of 25 MHz can be
reduced by connecting a ground-referenced capacitor (C
the BFIN pin, as shown in Figure 35. This is generally done to
reduce output ripple (at twice the input frequency for a
symmetric input waveform such as sinusoidal signals).
AD8310
2µA/dB
3kΩ
V
+4
OUT
BFIN
C
FILT
FILT
) to
LOWERING THE HIGH-PASS CORNER FREQUENCY
OF THE OFFSET COMPENSATION LOOP
In normal operation using an ac-coupled input signal, the OFLT
pin should be left unconnected. Input-referred dc offsets of
about 1.5 mV in the signal path are nulled via an internal offset
control loop. This loop has a high-pass −3 dB corner at about
2 MHz. In low frequency ac-coupled applications, it is necessary
to lower this corner frequency to prevent input signals from
being misinterpreted as offsets. An external capacitor on OFLT
lowers the high-pass corner to arbitrarily low frequencies
(Figure 36). For example, by using a 1 µF capacitor, the 3 dB
corner is reduced to 60 Hz.
AD8310
C
= 1/(2π× 3kΩ× VIDEO BANDWIDTH) – 2.1pF
FILT
Figure 35. Lowering the Postdemodulation Video Bandwidth
C
is selected using the following equation:
FILT
=
C
FILT
()
1
k32
×Ω×π
idthVideoBandw
01084-035
pF1.2
−
The video bandwidth should typically be set at a frequency
equal to about one-tenth the minimum input frequency. This
ensures that the output ripple of the demodulated log output,
which is at twice the input frequency, is well filtered.
In many log amp applications, it might be necessary to lower the
corner frequency of the postdemodulation filtering to achieve
low output ripple while maintaining a rapid response time to
changes in signal level. An example of a 4-pole active filter is
shown in the AD8307 data sheet.
OFLT
C
OFLT
(SEE TEXT)
01084-036
Figure 36. Lowering the High-Pass Corner Frequency
of the Offset Control Loop
The corner frequency is set by the following equation:
f
where C
CORNER
=
()
is the capacitor connected to OFLT.
OFLT
1
C
××π
26252
OFLT
Rev. D | Page 18 of 24
AD8310
APPLICATIONS
The AD8310 is highly versatile and easy to use. It needs only a
few external components, most of which can be immediately
accommodated using the simple connections shown in the
Using the AD8310 section.
A few examples of more specialized applications are provided in
the following sections. See the AD8307 data sheet for more
applications (note the slightly different pin configuration).
CABLE-DRIVING
For a supply voltage of 3 V or greater, the AD8310 can drive a
grounded 100 Ω load to 2.5 V. If reverse-termination is required
when driving a 50 Ω cable, it should be included in series with
the output, as shown in Figure 37. The slope at the load is then
12 mV/dB. In some cases, it might be permissible to operate the
cable without a termination at the far end, in which case, the
slope is not lowered. Where a further increase in slope is
desirable, the scheme shown in Figure 34 can be used.
AD8310
50Ω
VOUT
Figure 37. Output Response of Cable-Driver Application
50Ω
01084-037
DC-COUPLED INPUT
It might occasionally be necessary to provide response to dc
inputs. Because the AD8310 is internally dc-coupled, there is no
reason why this cannot be done. However, its differential inputs
must be positioned at least 2 V above the COM potential for
proper biasing of the first stage. Usually, the source is a singlesided ground-referenced signal, so level-shifting and a singleended-to-differential conversion must be provided to correctly
drive the AD8310’s inputs.
Figure 38 shows how a level-shift to midsupply (2.5 V in this
example) and a single-ended-to-differential conversion can be
accomplished using the AD8138 differential amplifier. The four
499 Ω resistors set up a gain of unity. An output common-mode
(or bias) voltage of 2.5 is achieved by applying 2.5 V from a
supply-referenced resistive divider to the AD8138’s V
The differential outputs of the AD8138 directly drive the 1.1 kΩ
input impedance of the AD8310.
OCM
pin.
499Ω
5V
0.1µF
5V
10kΩ
10kΩ
499Ω
0.1µF
AD8138
499Ω
499Ω
50Ω
NC = NO CONNECT
SIGNAL
INPUT
2.5V
Figure 38. DC-Coupled Log Amp
In this application the offset voltage of the AD8138 must be
trimmed. The internal offset compensation circuitry of the
AD8310 is disabled by applying a nominal voltage of ~1.9 V to
the OFLT pin, so the trim on the AD8138 is effectively
trimming both devices’ offsets. The trim is done by grounding
the circuit’s input and slightly varying the gain resistors on the
AD8138’s inverting input (a 50 Ω potentiometer is used in this
example) until the voltage on the AD8310’s output reaches a
minimum.
After trimming, the lower end of the dynamic range is limited
by the broadband noise at the output of the AD8138, which is
approximately 425 µV p-p. A differential low-pass filter can be
added between the AD8138 and the AD8310 when the very fast
pulse response of the circuit is not required.
2.7
2.5
2.3
2.1
1.9
1.7
1.5
RSSI OUTPUT (V)
1.3
1.1
0.9
0.7
0.1
1
INPUT LEVEL (mV)
Figure 39. Transfer Function of DC-Coupled Log Amp Application
0.01µF
8765
INHI ENBL BFIN VPOS
NC
AD8310
INLO COMM OFLT VOUT
1234
3.01kΩ1.87kΩ
10 100 1000
5V
V
OUT
5V
01084-039
01084-038
Rev. D | Page 19 of 24
AD8310
0
0
EVALUATION BOARD
An evaluation board is available, which has been carefully laid
out and tested to demonstrate the specified high speed
performance of the AD8310. Figure 40 shows the schematic of
the evaluation board, which follows the basic connections
schematic shown in Figure 27.
Connectors INHI, INLO, and VOUT are of the SMA type.
Supply and ground are connected to the TP1 and TP2 vector
pins. The layout and silkscreen for the component side of the
board are shown in Figure 41 and Figure 42. Switches and
component settings for different setups are described in Table 6.
For ordering information, see the
found.
.
SW1
A
R4
R1
INHI
Ω
INLO
Ω
C2
0.01µF
52.3Ω
C1
0.01µF
R2
0Ω
B
INHI ENBL BFIN VPOS
R3
INLO COMM OFLT VOUT
Figure 40. Evaluation Board Schematic
C3
OPEN
8765
AD8310
123
TP2
Error! Reference source not
TP1
R5
C7
OPEN
0Ω
C4
0.01µF
R6
4
0Ω
VPOS
C5
OPEN
W1 W2
C6
OPEN
R7
OPEN
V
OUT
01084-040
Figure 41. Layout of the Component Side of the Evaluation Board
01084-041
01084-042
Figure 42. Component Side Silkscreen of the Evaluation Board
Rev. D | Page 20 of 24
Preliminary Technical Data AD8310
Table 6. Evaluation Boards Setup Options
Component Function Default Condition
TP1, TP2 Supply and Ground Vector Pins. Not Applicable
SW1
R1/R4
C1, C2, R3
C3
C4, C5, R5
R6
W1,W2, C6, R7
C7
Device Enable. When in Position A, the ENBL pin is connected to +V
and the AD8310 is in normal
S
operating mode. In Position B, the ENBL pin is connected to ground, putting the device into sleep
mode.
SMA Connector Grounds.Connects common of INHI and INLO SMA connectors to ground. Can be
used to isolate the generator ground from the evaluation board ground (see Figure 28).
Input Interface. R3 (52.3 Ω) combines with the AD8310’s 1 kΩ input impedance to give an overall
broadband input impedance of 50 Ω. C1, C2, and the AD8310’s input impedance combine to set a
high-pass input corner of 32 kHz. Alternatively, R3, C1, and C2 can be replaced by an indicator and
matching capacitors to form an input matching network. See the Input Matching section for
details.
RSSI (Video) Bandwidth Adjust. The addition of C3 (farads) lowers the RSSI bandwidth of the VLOG
output according to the following equation:
= 1/(2π × 3 kΩ Video Bandwidth) – 2.1 pF
C
FILT
Supply Decoupling. The normal supply decoupling of 0.01 µF (C4) can be augmented by a larger
capacitor in C5. An inductor or small resistor can be placed in R5 for additional decoupling.
Output Source Impedance. In cable-driving applications, a resistor (typically 50 Ω or 75 Ω) can be
placed in R6 to give the circuit a back-terminated output impedance.
Output Loading. Resistors and capacitors can be placed in C6 and R7 to load-test VOUT. Jumpers
W1 and W2 are used to connect or disconnect the loads.
Offset Compensation Loop. A capacitor in C7 reduces the corner frequency of the offset control
loop in low frequency applications.
SW1 = A
R1 = R4 = 0 Ω
R3 = 52.3 Ω,
C1 = C2 = 0.01 µF
C3 = Open
C4 = 0.01 µF,
C5 = Open, R5 = 0
R6 = 0 Ω
C6 = R7 = Open,
W1 = W2 = Installed
C7 = Open
Rev. D | Page 21 of 24
AD8310 Preliminary Technical Data
OUTLINE DIMENSIONS
3.00
BSC
8
5
3.00
BSC
PIN 1
1
0.65 BSC
4.90
BSC
4
0.15
0.00
0.38
0.22
COPLANARITY
0.10
COMPLIANT TO JEDEC STANDARDS MO-187AA
1.10 MAX
SEATING
PLANE
0.23
0.08
8°
0°
0.80
0.60
0.40
Figure 43. 8-Lead Mini Small Outline Package [MSOP]
(RM-8)
Dimensions shown in millimeters
ORDERING GUIDE
Model Temperature Range Package Description Package Option Branding
AD8310ARM −40°C to +85°C 8-Lead MSOP, Tube RM-8 J6A
AD8310ARM-REEL7 −40°C to +85°C 8-Lead MSOP, 7 Tape and Reel RM-8 J6A
AD8310ARMZ-REEL
AD8310ARMZ-REEL7
AD8310-EVAL Evaluation Board
1
Z = Pb-free part.
1
−40°C to +85°C 8-Lead MSOP, 13 Tape and Reel RM-8 J6A
1
−40°C to +85°C 8-Lead MSOP, 7 Tape and Reel RM-8 J6A
Rev. D | Page 22 of 24
Preliminary Technical Data AD8310
NOTES
Rev. D | Page 23 of 24
AD8310 Preliminary Technical Data
NOTES
© 2004 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C01084–0–10/04(D)
Rev. D | Page 24 of 24
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