Analog Devices AD734SQ-883B, AD734SCHIPS, AD734BQ, AD734BN, AD734AQ Datasheet

REV. C

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
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which may result from its use. No license is granted by implication or
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a

AD734

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] World Wide Web Site: http://www.analog.com
Fax: [phone] © Analog Devices, Inc., 1999

10 MHz, 4-Quadrant

Multiplier/Divider

FEATURES
High Accuracy

0.1% Typical Error

High Speed

10 MHz Full-Power Bandwidth
450 V/␮s Slew Rate
200 ns Settling to 0.1% at Full Power

Low Distortion

–80 dBc from Any Input
Third-Order IMD Typically –75 dBc at 10 MHz

Low Noise

94 dB SNR, 10 Hz to 20 kHz
70 dB SNR, 10 Hz to 10 MHz

Direct Division Mode

2 MHz BW at Gain of 100

APPLICATIONS
High Performance Replacement for AD534
Multiply, Divide, Square, Square Root
Modulator, Demodulator
Wideband Gain Control, RMS-DC Conversion
Voltage-Controlled Amplifiers, Oscillators, and Filters
Demodulator with 40 MHz Input Bandwidth

demodulator with input frequencies as high as 40 MHz as long
as the desired output frequency is less than 10 MHz.

The AD734AQ and AD734BQ are specified for the industrial

temperature range of –40°C to +85°C and come in a 14-lead

ceramic DIP. The AD734SQ/883B, available processed to

MIL-STD-883B for the military range of –55°C to +125°C, is

available in a 14-lead ceramic DIP.

PRODUCT HIGHLIGHTS

The AD734 embodies more than two decades of experience in
the design and manufacture of analog multipliers, to provide:

1. A new output amplifier design with more than twenty times

the slew-rate of the AD534 (450 V/µs versus 20 V/µs) for a

full power (20 V pk-pk) bandwidth of 10 MHz.

2. Very low distortion, even at full power, through the use of
circuit and trimming techniques that virtually eliminate all of
the spurious nonlinearities found in earlier designs.

3. Direct control of the denominator, resulting in higher
multiplier accuracy and a gain-bandwidth product at small
denominator values that is typically 200 times greater than
that of the AD534 in divider modes.

4. Very clean transient response, achieved through the use of a
novel input stage design and wide-band output amplifier,
which also ensure that distortion remains low even at high
frequencies.

5. Superior noise performance by careful choice of device
geometries and operating conditions, which provide a
guaranteed 88 dB of dynamic range in a 20 kHz bandwidth.

CONNECTION DIAGRAM

14-Lead DIP

(Q Package and N Package)

14
13
12
11
10

9
8

1
2
3
4

7

6

5

TOP VIEW

(Not to Scale)

X1

Z1

W OUTPUT

DD DENOMINATOR DISABLE

VP POSITIVE SUPPLY
X2
U0
U1

AD734

VN NEGATIVE SUPPLY

ER REFERENCE VOLTAGE

Z2

U2
Y1

Y2

Z INPUT

X INPUT

DENOMINATOR

INTERFACE

Y INPUT

PRODUCT DESCRIPTION

The AD734 is an accurate high speed, four-quadrant analog
multiplier that is pin-compatible with the industry-standard
AD534 and provides the transfer function W = XY/U. The
AD734 provides a low-impedance voltage output with a full­power (20 V pk-pk) bandwidth of 10 MHz. Total static error
(scaling, offsets, and nonlinearities combined) is 0.1% of full
scale. Distortion is typically less than –80 dBc and guaranteed.
The low capacitance X, Y and Z inputs are fully differential. In
most applications, no external components are required to
define the function.

The internal scaling (denominator) voltage U is 10 V, derived
from a buried-Zener voltage reference. A new feature provides
the option of substituting an external denominator voltage,
allowing the use of the AD734 as a two-quadrant divider with a
1000:1 denominator range and a signal bandwidth that remains
10 MHz to a gain of 20 dB, 2 MHz at a gain of 40 dB and
200 kHz at a gain of 60 dB, for a gain-bandwidth product of
200 MHz.

The advanced performance of the AD734 is achieved by a
combination of new circuit techniques, the use of a high speed
complementary bipolar process and a novel approach to laser­trimming based on ac signals rather than the customary dc
methods. The wide bandwidth (>40 MHz) of the AD734’s
input stages and the 200 MHz gain-bandwidth product of the
multiplier core allow the AD734 to be used as a low distortion

–2–

REV. C

AD734–SPECIFICATIONS

TRANSFER FUNCTION

W = A

O

X

1−X2

()

Y

1−Y2

()

U

1−U2

()

− Z1−Z

2

()

















ABS

Parameter Conditions Min Typ Max Min Typ Max Min Typ Max Units

MULTIPLIER PERFORMANCE

Transfer Function W = XY/10 W = XY/10 W = XY/10
Total Static Error

1

–10 V ≤ X, Y ≤ 10 V 0.1 0.4 0.1 0.25 0.1 0.4 %

Over T

MIN

to T

MAX

1 0.6 1.25 %

vs. Temperature T

MIN

to T

MAX

0.004 0.003 0.004 %/°C

vs. Either Supply ±V

S

= 14 V to 16 V 0.01 0.05 0.01 0.05 0.01 0.05 %/V

Peak Nonlinearity –10 V ≤ X ≤ +10 V, Y = +10 V 0.05 0.05 0.05 %

–10 V ≤ Y ≤ +10 V, X = +10 V 0.025 0.025 0.025 %

THD

2

X = 7 V rms, Y = +10 V, f ≤ 5 kHz –58 –66 –58 dBc

T

MIN

to T

MAX

–55 –63 –55 dBc

Y = 7 V rms, X = +10 V, f ≤ 5 kHz –60 –80 –60 dBc

T

MIN

to T

MAX

–57 –74 –57 dBc

Feedthrough X = 7 V rms, Y = nulled, f ≤ 5 kHz –85 –60 –85 –70 –85 –60 dBc

Y = 7 V rms, X = nulled, f ≤ 5 kHz –85 –66 –85 –76 –85 –66 dBc

Noise (RTO) X = Y = 0

Spectral Density 100 Hz to 1 MHz 1.0 1.0 1.0 µV/√Hz

Total Output Noise 10 Hz to 20 kHz –94 –88 –94 –88 –94 –88 dBc

T

MIN

to T

MAX

–85 –85 –85 dBc

DIVIDER PERFORMANCE (Y = 10 V)

Transfer Function W = XY/U W = XY/U W = XY/U
Gain Error Y = 10 V, U = 100 mV to 10 V 1 1 1 %

X Input Clipping Level Y ≤ 10 V 1.25 × U 1.25 × U 1.25 × UV

U Input Scaling Error

3

0.3 0.15 0.3 %

T

MIN

to T

MAX

0.8 0.65 1 %

(Output to 1%) U = 1 V to 10 V Step, X = 1 V 100 100 100 ns

INPUT INTERFACES (X, Y, & Z)

3 dB Bandwidth 40 40 40 MHz

Operating Range Differential or Common Mode ±12.5 ±12.5 ±12.5 V

X Input Offset Voltage 15 5 15 mV

T

MIN

to T

MAX

25 15 25 mV

Y Input Offset Voltage 10 5 10 mV

T

MIN

to T

MAX

12 6 12 mV

Z Input Offset Voltage 20 10 20 mV

T

MIN

to T

MAX

50 50 90 mV

Z Input PSRR (Either Supply) f ≤ 1 kHz 54 70 66 70 54 70 dB

T

MIN

to T

MAX

50 56 50 dB

CMRR f = 5 kHz 70 85 70 85 70 85 dB

Input Bias Current (X, Y, Z Inputs) 50 300 50 150 50 300 nA

T

MIN

to T

MAX

400 300 500 nA

Input Resistance Differential 50 50 50 kΩ

Input Capacitance Differential 2 2 2 pF

DENOMINATOR INTERFACES (U0, U1, & U2)

Operating Range VN to VP-3 VN to VP-3 VN to VP-3 V
Denominator Range 1000:1 1000:1 1000:1

Interface Resistor U1 to U2 28 28 28 kΩ

OUTPUT AMPLIFIER (W)

Output Voltage Swing T

MIN

to T

MAX

±12 ±12 ±12 V

Open-Loop Voltage Gain X = Y = 0, Input to Z 72 72 72 dB

Dynamic Response From X or Y Input, CL ≤ 20 pF

3 dB Bandwidth W ≤ 7 V rms 8 10 8 10 8 10 MHz
Slew Rate 450 450 450 V/µs

Settling Time +20 V or –20 V Output Step

To 1% 125 125 125 ns
To 0.1% 200 200 200 ns

Short-Circuit Current T

MIN

to T

MAX

20 50 80 20 50 80 20 50 80 mA

POWER SUPPLIES, ±V

S

Operating Supply Range ±8 ±16.5 ±8 ±16.5 ±8 ±16.5 V

Quiescent Current T

MIN

to T

MAX

6912 6912 6912 mA

NOTES

1

Figures given are percent of full scale (e.g., 0.01% = 1 mV).

2

dBc refers to deciBels relative to the full-scale input (carrier) level of 7 V rms.

3

See Figure 10 for test circuit.
All min and max specifications are guaranteed.
Specifications subject to change without notice.

(TA = +25ⴗC, +VS = VP = +15 V, –VS = VN = –15 V, R

L

≥ 2 k⍀)

AD734

–3–

REV. C

ABSOLUTE MAXIMUM RATINGS

1

Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±18 V

Internal Power Dissipation

2

for T

J

max = 175°C . . . . . . . . . . . . . . . . . . . . . . . . 500 mW

X, Y and Z Input Voltages . . . . . . . . . . . . . . . . . . . . VN to VP

Output Short Circuit Duration . . . . . . . . . . . . . . . . Indefinite

Storage Temperature Range

Q . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to +150°C

Operating Temperature Range

AD734A, B (Industrial) . . . . . . . . . . . . . . . –40°C to +85°C

AD734S (Military) . . . . . . . . . . . . . . . . . . –55°C to +125°C

Lead Temperature Range (soldering 60 sec) . . . . . . . . +300°C

Transistor Count . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 81

ESD Rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 500 V

NOTES

1

Stresses above those listed under Absolute Maximum Ratings may cause perma­nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied.

2

14-Lead Ceramic DIP: θ

JA

= 110°C/W.

CHIP DIMENSIONS & BONDING DIAGRAM

Dimensions shown in inches and (mm).
(Contact factory for latest dimensions.)

ORDERING GUIDE

Temperature Package Package

Model Range Description Option

AD734AN –40°C to +85°C Plastic DIP N-14
AD734BN –40°C to +85°C Plastic DIP N-14
AD734AQ –40°C to +85°C Cerdip Q-14
AD734BQ –40°C to +85°C Cerdip Q-14
AD734SQ/883B –55°C to +125°C Cerdip Q-14
AD734SCHIPS –55°C to +125°CDie

AD734

–4–

REV. C

is typically less than 5 mV, which corresponds to a bias current
of only 100 nA. This low bias current ensures that mismatches
in the sources resistances at a pair of inputs does not cause an
offset error. These currents remain low over the full temperature
range and supply voltages.

The common-mode range of the X, Y and Z inputs does not
fully extend to the supply rails. Nevertheless, it is often possible
to operate the AD734 with one terminal of an input pair con­nected to either the positive or negative supply, unlike previous
multipliers. The common-mode resistance is several megohms.

The full-scale output of ±10 V can be delivered to a load resis­tance of 1 kΩ (although the specifications apply to the standard
multiplier load condition of 2 kΩ). The output amplifier is

stable driving capacitive loads of at least 100 pF, when a slight
increase in bandwidth results from the peaking caused by this

capacitance. The 450 V/µs slew rate of the AD734’s output am-

plifier ensures that the bandwidth of 10 MHz can be maintained
up to the full output of 20 V pk-pk. Operation at reduced supply

voltages is possible, down to ±8 V, with reduced signal levels.

Available Transfer Functions

The uncommitted (open-loop) transfer function of the AD734 is

W = A

O

X

1

− X

2

()

Y

1−Y2

()

U

− Z1− Z

2

()

















, (1)

where A

O

is the open-loop gain of the output op-amp, typically
72 dB. When a negative feedback path is provided, the circuit
will force the quantity inside the brackets essentially to zero,
resulting in the equation

(X

1

– X2)(Y1 – Y2) = U (Z1 – Z2) (2)

This is the most useful generalized transfer function for the
AD734; it expresses a balance between the product XY and the
product UZ. The absence of the output, W, in this equation
only reflects the fact that we have not yet specified which of the
inputs is to be connected to the op amp output.

Most of the functions of the AD734 (including division, unlike
the AD534 in this respect) are realized with Z

1

connected to W.

So, substituting W in place of Z

1

in the above equation results in

an output.

W =

X

1

− X

2

()

Y

1−Y2

()

U

+ Z

2

.

(3)

The free input Z2 can be used to sum another signal to the
output; in the absence of a product signal, W simply follows the
voltage at Z2 with the full 10 MHz bandwidth. When not
needed for summation, Z2 should be connected to the ground
associated with the load circuit. We can show the allowable
polarities in the following shorthand form:

±W

()

=

±X

()

±Y

()

+U

()

+±Z.

(4)

In the recommended direct divider mode, the Y input is set to a
fixed voltage (typically 10 V) and U is varied directly; it may
have any value from 10 mV to 10 V. The magnitude of the ratio
X/U cannot exceed 1.25; for example, the peak X-input for U

= 1 V is ±1.25 V. Above this level, clipping occurs at the

positive and negative extremities of the X-input. Alternatively,

Ru

DENOMINATOR

CONTROL

DD

ER

∑

XY/U – Z

∞

HIGH-ACCURACY
TRANSLINER
MULTIPLIER CORE

AD734

X1
X2

U0
U1

U2

Y1

Y2

W

Z1
Z2

U

X = X1 – X

2

Y = Y1 – Y

2

Z = Z1 – Z

2

WIF

ZIF

A

O

XIF

YIF

XZ

U

Figure 1. AD734 Block Diagram

FUNCTIONAL DESCRIPTION

Figure 1 is a simplified block diagram of the AD734. Operation
is similar to that of the industry-standard AD534 and in many
applications these parts are pin-compatible. The main functional
difference is the provision for direct control of the denominator
voltage, U, explained fully on the following page. Internal sig­nals are actually in the form of currents, but the function of the
AD734 can be understood using voltages throughout, as shown
in this figure. Pins are named using upper-case characters (such
as X1, Z2) while the voltages on these pins are denoted by sub­scripted variables (for example, X

1

, Z2).

The AD734’s differential X, Y and Z inputs are handled by
wideband interfaces that have low offset, low bias current and
low distortion. The AD734 responds to the difference signals
X = X

1

– X2, Y = Y1 – Y2 and Z = Z1 – Z2, and rejects

common-mode voltages on these inputs. The X, Y and Z

interfaces provide a nominal full-scale (FS) voltage of ±10 V,

but, due to the special design of the input stages, the linear

range of the differential input can be as large as ±17 V. Also

unlike previous designs, the response on these inputs is not

clipped abruptly above ±15 V, but drops to a slope of one half.

The bipolar input signals X and Y are multiplied in a translinear
core of novel design to generate the product XY/U. The
denominator voltage, U, is internally set to an accurate,
temperature-stable value of 10 V, derived from a buried-Zener
reference. An uncalibrated fraction of the denominator voltage
U appears between the voltage reference pin (ER) and the
negative supply pin (VN), for use in certain applications where
a temperature-compensated voltage reference is desirable. The
internal denom-inator, U, can be disabled, by connecting the
denominator disable Pin 13 (DD) to the positive supply pin
(VP); the denom-inator can then be replaced by a fixed or
variable external volt-age ranging from 10 mV to more than 10 V.

The high-gain output op-amp nulls the difference between
XY/U and an additional signal Z, to generate the final output
W. The actual transfer function can take on several forms, de­pending on the connections used. The AD734 can perform all
of the functions supported by the AD534, and new functions
using the direct-division mode provided by the U-interface.

Each input pair (X1 and X2, Y1 and Y2, Z1 and Z2) has a

differential input resistance of 50 kΩ; this is formed by “real”

resistors (not a small-signal approximation) and is subject to a

tolerance of ±20%. The common-mode input resistance is

several megohms and the parasitic capacitance is about 2 pF.

The bias currents associated with these inputs are nulled by
laser-trimming, such that when one input of a pair is optionally
ac-coupled and the other is grounded, the residual offset voltage

AD734

–5–

REV. C

After temperature-correction (block TC), the reference voltage
is applied to transistor Qd and trimmed resistor Rd, which
generate the required reference current. Transistor Qu and
resistor Ru are not involved in setting up the internal denominator,
and their associated control pins U0, U1 and U2 will normally
be grounded. The reference voltage is also made available, via

the 100 kΩ resistor Rr, at Pin 9 (ER); the purpose of Qr is

explained below.

When the control pin DD (denominator disable) is connected to
VP, the internal source of Iu is shut off, and the collector cur­rent of Qu must provide the denominator current. The resistor
Ru is laser-trimmed such that the multiplier denominator is
exactly equal to the voltage across it (that is, across pins U1 and
U2). Note that this trimming only sets up the correct internal

ratio; the absolute value of Ru (nominally 28 kΩ) has a
tolerance of ±20%. Also, the alpha of Qu, (typically 0.995)

which might be seen as a source of scaling error, is canceled by
the alpha of other transistors in the complete circuit.

In the simplest scheme (Figure 3), an externally-provided
control voltage, V

G

, is applied directly to U0 and U2 and the

resulting voltage across Ru is therefore reduced by one V

BE

. For

example, when V

G

= 2 V, the actual value of U will be about

1.3 V. This error will not be important in some closed-loop
applications, such as automatic gain control (AGC), but clearly
is not acceptable where the denominator value must be well­defined. When it is required to set up an accurate, fixed value of
U, the on-chip reference may be used. The transistor Qr is
provided to cancel the V

BE

of Qu, and is biased by an external
resistor, R2, as shown in Figure 4. R1 is chosen to set the de­sired value of U and consists of a fixed and adjustable resistor.

Ru
28kV

Qu

Iu

U0

U1

U2

VP

DD

AD734

~

60mA

+V

S

–V

S

NC

NC

ER

VN

Qr

Rr

100kV

V

G

3

4

5

9

8

13

14

Figure 3. Low-Accuracy Denominator Control

R1

Ru
28kV

Qu

Iu

U0

U1

U2

VP

DD

AD734

+V

S

–V

S

NOM
8V

NC

ER

VN

Qr

Rr

100kV

3

4

5

9

8

13

14

R2

Figure 4. Connections for a Fixed Denominator

Table I shows useful values of the external components for set­ting up nonstandard denominator values.

the AD734 can be operated using the standard (AD534) divider
connections (Figure 8), when the negative feedback path is
established via the Y

2

input. Substituting W for Y2 in Equation

(2), we get

W =U

Z

2

− Z

1

()

X

1

− X

2

()

+Y

1

.

(5)

In this case, note that the variable X is now the denominator,

and the above restriction (X/U ≤ 1.25) on the magnitude of the

X input does not apply. However, X must be positive in order
for the feedback polarity to be correct. Y

1

can be used for
summing purposes or connected to the load ground if not
needed. The shorthand form in this case is

±W

()

=+U

()

±Z

()

+X

()

+±Y

()

.

(6)

In some cases, feedback may be connected to two of the avail­able inputs. This is true for the square-rooting connections
(Fig-ure 9), where W is connected to both X

1

and Y2. Setting

X

1

= W and Y2 = W in Equation (2), and anticipating the

possibility of again providing a summing input, so setting X

2

= S

and Y

1

= S, we find, in shorthand form

±W

()

=+U

()

+Z

()

+±S

()

.

(7)

This is seen more generally to be the geometric-mean function,
since both U and Z can be variable; operation is restricted to
one quadrant. Feedback may also be taken to the U-interface.
Full details of the operation in these modes is provided in the
appropriate section of this data sheet.

Direct Denominator Control

A valuable new feature of the AD734 is the provision to replace
the internal denominator voltage, U, with any value from +10 mV
to +10 V. This can be used (1) to simply alter the multiplier
scaling, thus improve accuracy and achieve reduced noise levels
when operating with small input signals; (2) to implement an
accurate two-quadrant divider, with a 1000:1 gain range and an
asymptotic gain-bandwidth product of 200 MHz; (3) to achieve
certain other special functions, such as AGC or rms.

Figure 2 shows the internal circuitry associated with denomina­tor control. Note first that the denominator is actually proportional

to a current, Iu, having a nominal value of 356 µA for U = 10 V,

whereas the primary reference is a voltage, generated by a buried­Zener circuit and laser-trimmed to have a very low temperature
coefficient. This voltage is nominally 8 V with a tolerance of

±10%.

Ru

28kV

Rd
NOM

22.5kV

Qu

Qd

NOM
8V

Rr

100kV

TC

Qr

NEGATIVE SUPPLY

NOMINALLY
356mA for
U = 10V

Iu

U0

U1

U2

VP

DD

ER

VN

AD734

LINK TO
DISABLE

3

4

5

8

9

14

13

Figure 2. Denominator Control Circuitry

AD734

–6–

REV. C

Table I. Component Values for Setting Up Nonstandard
Denominator Values

Denominator R1 (Fixed) R1 (Variable) R2

5 V 34.8 kΩ 20 kΩ 120 kΩ
3 V 64.9 kΩ 20 kΩ 220 kΩ
2 V 86.6 kΩ 50 kΩ 300 kΩ
1 V 174 kΩ 100 kΩ 620 kΩ

The denominator can also be current controlled, by grounding
Pin 3 (U0) and withdrawing a current of Iu from Pin 4 (U1).

The nominal scaling relationship is U = 28 × Iu, where u is

expressed in volts and Iu is expressed in milliamps. Note,
however, that while the linearity of this relationship is very good,

it is subject to a scale tolerance of ±20%. Note that the common

mode range on Pins 3 through 5 actually extends from 4 V to
36 V below VP, so it is not necessary to restrict the connection
of U0 to ground if it should be desirable to use some other
voltage.

The output ER may also be buffered, re-scaled and used as a
general-purpose reference voltage. It is generated with respect to
the negative supply line Pin 8 (VN), but this is acceptable when
driving one of the signal interfaces. An example is shown in
Figure 12, where a fixed numerator of 10 V is generated for a
divider application. There, Y

2

is tied to VN but Y1 is 10 V above
this; therefore the common-mode voltage at this interface is still
5 V above VN, which satisfies the internal biasing requirements
(see Specifications table).

OPERATION AS A MULTIPLIER

All of the connection schemes used in this section are essentially
identical to those used for the AD534, with which the AD734 is
pin-compatible. The only precaution to be noted in this regard
is that in the AD534, Pins 3, 5, 9, and 13 are not internally
connected and Pin 4 has a slightly different purpose. In many
cases, an AD734 can be directly substituted for an AD534 with
immediate benefits in static accuracy, distortion, feedthrough,
and speed. Where Pin 4 was used in an AD534 application to
achieve a reduced denominator voltage, this function can now be
much more precisely implemented with the AD734 using alter­native connections (see Direct Denominator Control, page 5).

Operation from supplies down to ±8 V is possible. The supply

current is essentially independent of voltage. As is true of all
high speed circuits, careful power-supply decoupling is impor­tant in maintaining stability under all conditions of use. The
decoupling capacitors should always be connected to the load
ground, since the load current circulates in these capacitors at
high frequencies. Note the use of the special symbol (a triangle
with the letter ‘L’ inside it) to denote the load ground.

Standard Multiplier Connections

Figure 5 shows the basic connections for multiplication. The X
and Y inputs are shown as optionally having their negative
nodes grounded, but they are fully differential, and in many
applications the grounded inputs may be reversed (to facilitate
interfacing with signals of a particular polarity, while achieving
some desired output polarity) or both may be driven.

The AD734 has an input resistance of 50 kΩ ± 20% at the X,

Y, and Z interfaces, which allows ac-coupling to be achieved
with moderately good control of the high-pass (HP) corner

frequency; a capacitor of 0.1 µF provides a HP corner frequency

1
2
3
4
5
6
7

10

8

9

11

13
12

14

W

ER
VN

VP
DD

Z1

Z2

X1
X2

U1
U2

U0

Y1
Y2

AD734

NC

NC

LOAD
GROUND

0.1

mF

0.1

mF

X – INPUT

610V FS

Y – INPUT

610V FS

+15V

–15V

OPTIONAL
SUMMING INPUT
±10V FS

W =

(X

1

– X2)

10V

(Y

1

– Y2)

+ Z

2

L

L

Z

2

Figure 5. Basic Multiplier Circuit

of 32 Hz. When a tighter control of this frequency is needed, or
when the HP corner is above about 100 kHz, an external resis­tor should be added across the pair of input nodes.

At least one of the two inputs of any pair must be provided with
a dc path (usually to ground). The careful selection of ground
returns is important in realizing the full accuracy of the AD734.
The Z2 pin will normally be connected to the load ground,
which may be remote, in some cases. It may also be used as an
optional summing input (see Equations (3) and (4), above)

having a nominal FS input of ±10 V and the full 10 MHz

bandwidth.

In applications where high absolute accuracy is essential, the
scaling error caused by the finite resistance of the signal source(s)

may be troublesome; for example, a 50 Ω source resistance at

just one input will introduce a gain error of –0.1%; if both the

X- and Y-inputs are driven from 50 Ω sources, the scaling error

in the product will be –0.2%. Provided the source resistance(s)
are known, this gain error can be completely compensated by

including the appropriate resistance (50 Ω or 100 Ω, respectively,

in the above cases) between the output W (Pin 12) and the Z1
feedback input (Pin 11). If Rx is the total source resistance
associated with the X1 and X2 inputs, and Ry is the total source
resistance associated with the Y1 and Y2 inputs, and neither Rx

nor Ry exceeds 1 kΩ, a resistance of Rx+Ry in series with pin

Z1 will provide the required gain restoration.

Pins 9 (ER) and 13 (DD) should be left unconnected in this
application. The U-inputs (Pins 3, 4 and 5) are shown
connected to ground; they may alternatively be connected to
VN, if desired. In applications where Pin 2 (X2) happens to be
driven with a high-amplitude, high-frequency signal, the
capacitive coupling to the denominator control circuitry via an
ungrounded Pin 3 can cause high-frequency distortion. However,
the AD734 can be operated without modification in an AD534
socket, and these three pins left unconnected, with the above
caution noted.

1
2
3
4
5
6
7

10

8

9

11

13
1214W

ER
VN

VP
DD

Z1

Z2

X1
X2

U1
U2

U0

Y1
Y2

AD734

NC

NC

L

0.1mF

0.1mF

X – INPUT

610V FS

Y – INPUT

610V FS

+15V

–15V

I

W

=

(X1 – X2)

10V

(Y1 – Y2)

1

R

S

+

1

50kV

R

S

L

610mA MAX FS
610V MAXIMUM

LOAD VOLTAGE

L

LOAD

I

W

Figure 6. Conversion of Output to a Current

AD734

–7–

REV. C

Current Output

It may occasionally be desirable to convert the output voltage to
a current. In correlation applications, for example, multiplica­tion is followed by integration; if the output is in the form of a
current, a simple grounded capacitor can perform this function.
Figure 6 shows how this can be achieved. The op amp forces
the voltage across Z1 and Z2, and thus across the resistor R

S

, to
be the product XY/U. Note that the input resistance of the
Z interface is in shunt with R

S

, which must be calculated

accordingly.

The smallest FS current is simply ±10 V/50 kΩ, or ±200 µA,

with a tolerance of about 20%. To guarantee a 1% conversion
tolerance without adjustment, R

S

must be less than 2.5 kΩ. The

maximum full scale output current should be limited to about

±10 mA (thus, R

S

= 1 kΩ). This concept can be applied to all

connection modes, with the appropriate choice of terminals.

Squaring and Frequency-Doubling

Squaring of an input signal, E, is achieved simply by connecting
the X and Y inputs in parallel; the phasing can be chosen to
produce an output of E

2

/U or –E2/U as desired. The input may
have either polarity, but the basic output will either always be
positive or negative; as for multiplication, the Z2 input may be
used to add a further signal to the output.

When the input is a sine wave, a squarer behaves as a frequency­doubler, since

(Esinwt)

2

= E2 (1 – cos2wt)/2 (8)

Equation (8) shows a dc term at the output which will vary
strongly with the amplitude of the input, E. This dc term can be
avoided using the connection shown in Figure 7, where an
RC-network is used to generate two signals whose product has
no dc term. The output is

W = 4

E

2

sin wt +

π

4



















E

2

sin wt −

π

4



















1

10 V











(9)

for w = 1/CR1, which is just

W = E

2

(cos2wt)/( 10 V) (10)

which has no dc component. To restore the output to ±10 V

when E = 10 V, a feedback attenuator with an approximate ratio
of 4 is used between W and Z1; this technique can be used
wherever it is desired to achieve a higher overall gain in the
transfer function.

In fact, the values of R3 and R4 include additional compensa-

tion for the effects of the 50 kΩ input resistance of all three

interfaces; R2 is included for a similar reason. These resistor
values should not be altered without careful calculation of the
consequences; with the values shown, the center frequency f

0

is
100 kHz for C = 1 nF. The amplitude of the output is only a
weak function of frequency: the output amplitude will be 0.5%
too low at f = 0.9f

0

and f = 1.1f0. The cross-connection is simply
to produce the cosine output with the sign shown in Equation
(10); however, the sign in this case will rarely be important.

R2

1.6kV

R1

1.6kV

C

Esinωt

R3
13kV

R4

4.32kV

E cos2ωt2/10V

1
2
3
4
5
6
7

10

8

9

11

13
12

14

W

ER
VN

VP
DD

Z1

Z2

X1
X2

U1
U2

U0

Y1
Y2

AD734

NC

NC

0.1mF

0.1mF

+15V

–15V

L

L

L

Figure 7. Frequency Doubler

OPERATION AS A DIVIDER

The AD734 supports two methods for performing analog
division. The first is based on the use of a multiplier in a feed­back loop. This is the standard mode recommended for
multipliers having a fixed scaling voltage, such as the AD534,
and will be described in this Section. The second uses the
AD734’s unique capability for externally varying the scaling
(denominator) voltage directly, and will be described in the
next section.

Feedback Divider Connections

Figure 8 shows the connections for the standard (AD534)
divider mode. Feedback from the output, W, is now taken to the
Y2 (inverting) input, which, provided that the X-input is posi­tive, establishes a negative feedback path. Y1 should normally
be connected to the ground associated with the load circuit, but
may optionally be used to sum a further signal to the output. If
desired, the polarity of the Y-input connections can be reversed,
with W connected to Y1 and Y2 used as the optional summation
input. In this case, either the polarity of the X-input connections
must be reversed, or the X-input voltage must be negative.

Z INPUT

610V FS

X INPUT

+0.1V TO

+10V

OPTIONAL

SUMMING

INPUT

610V FS

W = 10

+Y1

(Z

2

– Z1)

(X1 – X2)

1
2
3
4
5
6
7

10

8

9

11

13
12

14

W

ER
VN

VP
DD

Z1

Z2

X1
X2

U1
U2

U0

Y1
Y2

AD734

NC

NC

0.1mF

0.1

mF

+15V

–15V

L

L

Y

1

L

Figure 8. Standard (AD534) Divider Connection

The numerator input, which is differential and can have either
polarity, is applied to pins Z1 and Z2. As with all dividers based
on feedback, the bandwidth is directly proportional to the
denominator, being 10 MHz for X = 10 V and reducing to
100 kHz for X = 100 mV. This reduction in bandwidth, and the
increase in output noise (which is inversely proportional to the
denominator voltage) preclude operation much below a denomi­nator of 100 mV. Division using direct control of the denominator
(Figure 10) does not have these shortcomings.

AD734

–8–

REV. C

(10V)W = (Z2 – Z1)+ S

S

OPTIONAL

SUMMING

INPUT

610V FS

Z INPUT
+10mV TO
+10V

D

1
2
3
4
5
6
7

10

8

9

11

13
12

14

W

ER
VN

VP
DD

Z1

Z2

X1
X2

U1
U2

U0

Y1
Y2

AD734

NC

NC

0.1mF

0.1mF

+15V

–15V

L

L

L

Figure 9. Connection for Square Rooting

Connections for Square-Rooting

The AD734 may be used to generate an output proportional to
the square-root of an input using the connections shown in
Figure 9. Feedback is now via both the X and Y inputs, and is
always negative because of the reversed-polarity between these
two inputs. The Z input must have the polarity shown, but
because it is applied to a differential port, either polarity of
input can be accepted with reversal of Z1 and Z2, if necessary.
The diode D, which can be any small-signal type (1N4148
being suitable) is included to prevent a latching condition which
could occur if the input momentarily was of the incorrect
polarity of the input, the output will be always negative.

Note that the loading on the output side of the diode will be

provided by the 25 kΩ of input resistance at X1 and Y2, and by

the user’s load. In high speed applications it may be beneficial

to include further loading at the output (to 1 kΩ minimum) to

speed up response time. As in previous applications, a further
signal, shown here as S, may be summed to the output; if this
option is not used, this node should be connected to the load
ground.

DIVISION BY DIRECT DENOMINATOR CONTROL

The AD734 may be used as an analog divider by directly vary­ing the denominator voltage. In addition to providing much
higher accuracy and bandwidth, this mode also provides greater
flexibility, because all inputs remain available. Figure 10 shows
the connections for the general case of a three-input multiplier
divider, providing the function

W =

X

1

− X

2

()

Y

1−Y2

()

U

1−U2

()

+ Z

2,

(11)

where the X, Y, and Z signals may all be positive or negative,
but the difference U = U

1

– U2 must be positive and in the
range +10 mV to +10 V. If a negative denominator voltage must
be used, simply ground the noninverting input of the op amp.
As previously noted, the X input must have a magnitude of less
than 1.25U.

2MV

X – INPUT

Y – INPUT

U – INPUT

1
2
3
4
5
6
7

10

8

9

11

13
12

14

W

ER
VN

VP
DD

Z1

Z2

X1
X2

U1
U2

U0

Y1
Y2

AD734

NC

LOAD
GROUND

0.1mF

0.1mF

+15V

–15V

OPTIONAL
SUMMING INPUT
610V FS

W =

(X

1

– X2)

(Y1 – Y2)

+ Z

2

U1 – U

2

L

L

Z

2

U

1

U

2

Figure 10. Three-Variable Multiplier/Divider Using Direct
Denominator Control

This connection scheme may also be viewed as a variable-gain
element, whose output, in response to a signal at the X input, is
controllable by both the Y input (for attenuation, using Y less
than U) and the U input (for amplification, using U less than
Y). The ac performance is shown in Figure 11; for these results,
Y was maintained at a constant 10 V. At U = 10 V, the gain is
unity and the circuit bandwidth is a full 10 MHz. At U = 1 V,
the gain is 20 dB and the bandwidth is essentially unaltered. At
U = 100 mV, the gain is 40 dB and the bandwidth is 2 MHz.
Finally, at U = 10 mV, the gain is 60 dB and the bandwidth is
250 kHz, corresponding to a 250 MHz gain-bandwidth product.

70

60

50

40

30

20

10

0

GAIN – dB

10k 100k 1M 10M

FREQUENCY – Hz

U = 10mV

U = 100mV

U = 1V

U = 10V

Figure 11. Three-Variable Multiplier/Divider Performance

The 2 MΩ resistor is included to improve the accuracy of the

gain for small denominator voltages. At high gains, the X input
offset voltage can cause a significant output offset voltage. To
eliminate this problem, a low-pass feedback path can be used
from W to X2; see Figure 13 for details.

Where a numerator of 10 V is needed, to implement a two­quadrant divider with fixed scaling, the connections shown in
Figure 12 may be used. The reference voltage output appearing
between Pin 9 (ER) and Pin 8 (VN) is amplified and buffered
by the second op amp, to impose 10 V across the Y1/Y2 input.
Note that Y2 is connected to the negative supply in this applica­tion. This is permissible because the common-mode voltage is
still high enough to meet the internal requirements. The transfer
function is

W =10 V

X

1

− X

2

U

1−U2











+ Z

2

.

(12)

The ac performance of this circuit remains as shown in Figure 11.

100kV

SCALE

ADJUST

200kV

10V

W =

(X1– X2)

+ Z

2

U1–U

2

OP AMP = AD712 DUAL

2MV

X – INPUT

U – INPUT

1
2
3
4
5
6
7

10

8

9

11

13
12

14

W

ER
VN

VP
DD

Z1

Z2

X1
X2

U1
U2

U0

Y1
Y2

AD734

L

LOAD
GROUND

0.1mF

0.1mF

+15V

–15V

OPTIONAL
SUMMING
INPUT
610V FS

L

Z

2

U

1

U

2

Figure 12. Two-Quadrant Divider with Fixed 10 V Scaling

AD734

–9–

REV. C

A PRECISION AGC LOOP

The variable denominator of the AD734 and its high gain­bandwidth product make it an excellent choice for precise
automatic gain control (AGC) applications. Figure 13 shows a
suggested method. The input signal, E

IN

, which may have a
peak amplitude of from 10 mV to 10 V at any frequency from
100 Hz to 10 MHz, is applied to the X input, and a fixed posi­tive voltage E

C

to the Y input. Op amp A2 and capacitor C2
form an integrator having a current summing node at its invert­ing input. (The AD712 dual op amp is a suitable choice for this
application.) In the absence of an input, the current in D2 and
R2 causes the integrator output to ramp negative, clamped by
diode D3, which is included to reduce the time required for the
loop to establish a stable, calibrated, output level once the
circuit has received an input signal. With no input to the
denominator (U0 and U2), the gain of the AD734 is very high
(about 70 dB), and thus even a small input causes a substantial
output.

E

OUT

D3

1N914

R2

1MVR11MV

C1

1mF

E

IN

R3

1MV

OP AMP = AD712 DUAL

1
2

3
4
5
6
7

10

8

9

11

13
12

14

W

ER
VN

VP
DD

Z1

Z2

X1
X2

U1
U2

U0

Y1
Y2

AD734

L

0.1mF

0.1mF

+15V

–15V

A1

NC

C2

1mF

A2

D2
1N914

+1V TO

+10V

E

C

D1

1N914

C1

1mF

L

Figure 13. Precision AGC Loop

Diode D1 and C1 form a peak detector, which rectifies the out­put and causes the integrator to ramp positive. When the
current in R1 balances the current in R2, the integrator output
holds the denominator output at a constant value. This occurs
when there is sufficient gain to raise the amplitude of E

IN

to that

required to establish an output amplitude of E

C

over the range
of +1 V to +10 V. The X input of the AD734, which has finite
offset voltage, could be troublesome at the output at high gains.
The output offset is reduced to that of the X input (one or two
millivolts) by the offset loop comprising R3, C3, and buffer A1.
The low pass corner frequency of 0.16 Hz is transformed to a
high-pass corner that is multiplied by the gain (for example,
160 Hz at a gain of 1000).

In applications not requiring operation down to low frequencies,
amplifier A1 can he eliminated, but the AD734’s input

resistance of 50 kΩ between X1 and X2 will reduce the time
constant and increase the input offset. Using a non-polar 20 µF

tantalum capacitor for C1 will result in the same unity-gain
high-pass corner; in this case, the offset gain increases to 20, still
very acceptable.

Figure 14 shows the error in the output for sinusoidal inputs at
100 Hz, 100 kHz, and 1 MHz, with E

C

set to +10 V. The out­put error for any frequency between 300 Hz and 300 kHz is
similar to that for 100 kHz. At low signal frequencies and low
input amplitudes, the dynamics of the control loop determine
the gain error and distortion; at high frequencies, the 200 MHz
gain-bandwidth product of the AD734 limit the available gain.

The output amplitude tracks E

C

over the range +1 V to slightly

more than +10 V.

+2

+1

0

–1

–2

ERROR – dB

10m 100m 1 10

INPUT AMPLITUDE – Volts

100kHZ

100HZ

1MHZ

Figure 14. AGC Amplifier Output Error vs. Input Voltage

WIDEBAND RMS-DC CONVERTER
USING U INTERFACE

The AD734 is well suited to such applications as implicit RMS­DC conversion, where the AD734 implements the function

V

RMS

=

avg V

IN

2

[]

V

RMS

(13)

using its direct divide mode. Figure 15 shows the circuit.

L

1/2

AD708

V

IN

2

VO =

V

O

R1

3.32kV

V

IN

1
2
3
4
5
6
7

10

8

9

11

13
12

14

W

ER
VN

VP

DD

Z1

Z2

X1
X2

U1
U2

U0

Y1
Y2

AD734

0.1mF

0.1mF

+15V

–15V

U2a

C1

47mF

U1

L

L

L

C2
1mF

L

L

L

1/2

AD708

U2b

Figure 15. A 2-Chip, Wideband RMS-DC Converter

In this application, the AD734 and an AD708 dual op amp
serve as a 2-chip RMS-DC converter with a 10 MHz bandwidth.
Figure 16 shows the circuit’s performance for square-, sine-,
and triangle-wave inputs. The circuit accepts signals as high as
10 V p-p with a crest factor of 1 or 1 V p-p with a crest factor of

10. The circuit’s response is flat to 10 MHz with an input of
10 V, flat to almost 5 MHz for an input of 1 V, and to almost
1 MHz for inputs of 100 mV. For accurate measurements of
input levels below 100 mV, the AD734’s output offset (Z inter­face) voltage, which contributes a dc error, must be trimmed out.

In Figure 15’s circuit, the AD734 squares the input signal, and
its output (V

IN

2

) is averaged by a low-pass filter that consists of
R1 and C1 and has a corner frequency of 1 Hz. Because of the
implicit feedback loop, this value is both the output value, V

RMS

,
and the denominator in Equation (13). U2a and U2b, an
AD708 dual dc precision op amp, serve as unity-gain buffers,
supplying both the output voltage and driving the U interface.

AD734

–10–

REV. C

SQUARE WAVE
SINE WAVE
TRI-WAVE

10k 100k 1M 10M

INPUT FREQUENCY – Hz

100

10

1

100m

10m

1m

100m

OUTPUT VOLTAGE – Volts

Figure 16. RMS-DC Converter Performance

LOW DISTORTION MIXER

The AD734’s low noise and distortion make it especially
suitable for use as a mixer, modulator, or demodulator.
Although the AD734’s –3 dB bandwidth is typically 10 MHz
and is established by the output amplifier, the bandwidth of its
X and Y interfaces and the multiplier core are typically in excess
of 40 MHz. Thus, provided that the desired output signal is less
than 10 MHz, as would typically be the case in demodulation,
the AD734 can be used with both its X and Y input signals as
high as 40 MHz. One test of mixer performance is to linearly
combine two closely spaced, equal-amplitude sinusoidal signals
and then mix them with a third signal to determine the mixer’s
2-tone Third-Order Intermodulation Products.

1
2
3
4
5
6
7

10

8

9

11

13
12

14

W

ER
VN

VP

DD

Z1

Z2

X1
X2

U1
U2

U0

Y1
Y2

AD734

0.1mF

0.1mF

+15V

–

2kV

HP3326A

COMBINE

A + B

DATEL

DVC-8500

HP3326A

HIGH VOLTAGE

OPTION

HP3585A
WITH 10X PROBE
dBm REF TO 50V

AD707

Figure 17. AD734 Mixer Test Circuit

Figure 17 shows a test circuit for measuring the AD734’s perfor­mance in this regard. In this test, two signals, at 10.05 MHz and

9.95 MHz are summed and applied to the AD734’s X interface.
A second 9 MHz signal is applied to the AD734’s Y interface.
The voltage at the U interface is set to 2 V to use the full
dynamic range of the AD734. That is, by connecting the W and
Z1 pins together, grounding the Y2 and X2 pins, and setting
U = 2 V, the overall transfer function is

W =

X

1Y1

2 V

(14)

and W can be as high as 20 V p-p when X1 = 2 V p-p and Y1 =
10 V p-p. The 2 V p-p signal level corresponds to +10 dBm into

a 50 Ω input termination resistor connected from X1 or Y1 to

ground.

If the two X1 inputs are at frequencies f

1

and f2 and the

frequency at the Y1 input is f

0

, then the two-tone third-order

intermodulation products should appear at frequencies 2f

1

– f

2

±

f

0

and 2f2 – f

1

± f

0

. Figures 18 and 19 show the output spectra of

the AD734 with f

1

= 9.95 MHz, f2 = 10.05 MHz, and f0 =

9.00 MHz for a signal level of f

1

& f2 of 6 dBm and f0 of

+24 dBm in Figure 18 and f

1

& f2 of 0 dBm and f0 of +24 dBm
in Figure 19. This performance is without external trimming of
the AD734’s X and Y input-offset voltages.

The possible Two Tone Intermodulation Products are at 2 ×

9.95 MHz – 10.05 MHz ± 9.00 MHz and 2 × 10.05 –

9.95 MHz ± 9.00 MHz; of these only the third-order products

at 0.850 MHz and 1.150 MHz are within the 10 MHz band­width of the AD734; the desired output signals are at

0.950 MHz and 1.050 MHz. Note that the difference (Figure

18) between the desired outputs and third-order products is
approximately 78 dB, which corresponds to a computed
third-order intercept point of +46 dBm.

Figure 18. AD734 Third-Order Intermodulation Performance
for f

1

= 9.95 MHz, f2 = 10.05 MHz, and f0 = 9.00 MHz and for

Signal Levels of f

1

& f2 of 6 dBm and f0 of +24 dBm. All Dis­played Signal Levels Are Attenuated 20 dB by the 10X Probe
Used to Measure the Mixer’s Output

Figure 19. AD734 Third-Order Intermodulation Performance
for f

1

= 9.95 MHz, f2 = 10.05 MHz, and f0 = 9.00 MHz and for

Signal Levels of f

1

& f2 of 0 dBm and f0 of +24 dBm. All Dis­played Signal Levels Are Attenuated 20 dB by the 10X Probe
Used to Measure the Mixer’s Output

AD734

–11–

REV. C

SIGNAL AMPLITUDE

DIFFERENTIAL PHASE – Degrees

–0.1

–2V 2V

0

–0.05

0.05
0

0.1

VS = 615V
R

LOAD

= 2kV

C

LOAD

= 20pF

Figure 21. Differential Phase at

3.58 MHz and R

L

= 2 k

Ω

FREQUENCY – Hz

100

0

PSRR – dB

80

60

40

20

1k 10k 10M100k 1M

VP

VN

Figure 24. PSRR vs. Frequency

FREQUENCY – Hz

0

dBc

–20

–40

–60

–80

1k 10k 10M100k 1M

Y INPUT

X INPUT

TEST INPUT = 7V RMS
OTHER INPUT = 10V DC
R

LOAD

= $2kV

Figure 27. THD vs. Frequency,
U = 10 V

FREQUENCY – Hz

GAIN FLATNESS

100k 1M 10M

0.3

0.2

0.1
0

–0.1
–0.2

–0.3
–0.4

VS = 615V
X = 1.4V RMS

Y = 10V
R

LOAD

= 500V

C

LOAD

= 20pF

Figure 22. Gain Flatness, 300 kHz to
10 MHz, R

L

= 500

Ω

FEEDTHROUGH – dBc

FREQUENCY – Hz

0

–40

–60

–80

–100

1k 10k 10M100k 1M

Y INPUT, X NULLED

INPUT SIGNAL = 7V RMS

X INPUT, Y NULLED

Figure 25. Feedthrough vs. Frequency

SIGNAL LEVEL

dBc

0

–100

–10dBm

70.7mV RMS

–20

–40

–60

–80

10dBm

707mV RMS

30dBm

7V RMS

FREQUENCY = 1MHz
VP = 15V
VN = –15V
R

LOAD

= 2kV

Y INPUT. X = 10V DC

X INPUT. Y = 10V DC

Figure 28. THD vs. Signal Level,
f = 1 MHz

SIGNAL AMPLITUDE

DIFFERENTIAL GAIN – dB

–0.04

–2V 2V

0

–0.02

–0.06
–0.08

0.02
0

0.06

0.04

VS = 615V
R

LOAD

= 2kV

C

LOAD

= 20pF

Figure 20. Differential Gain at

3.58 MHz and R

L

= 2 k

Ω

FREQUENCY – Hz

100

0

dB

80

60

40

20

1k 10k 10M100k 1M

X INPUT, Y = 10V

Y INPUT, X = 10V

COMMON-MODE
SIGNAL = 7V RMS

Figure 23. CMRR vs. Frequency

FREQUENCY – Hz

0

dBc

–20

–40

–60

–80

1k 10k 10M100k 1M

Y INPUT

X INPUT

TEST INPUT = 1V RMS
U = 2V
OTHER INPUT = 2V DC

Figure 26. THD vs. Frequency,
U = 2 V

Typical Characteristics–

–12–

C1442a–0–10/99

PRINTED IN U.S.A.

AD734–Typical Characteristics

REV. C

Figure 31. Pulse Response vs.
C

LOAD

, C

LOAD

= 0 pF, 47 pF,

100 pF, 200 pF

TEMPERATURE – 8C

DEVIATION OF INPUT

OFFSET VOLTAGE – mV

8

–55 –35

–15 5 25 45 65 85 105 125

6

4

2

0

–2

–4

INPUT OFFSET VOLTAGE
DRIFT WILL TYPICALLY BE
WITHIN SHADED AREA

–6

Figure 36. VOS Drift, Y Input

TEMPERATURE – 8C

DEVIATION OF INPUT

OFFSET VOLTAGE – mV

20

–15

–55 –35

–15 5 25 45 65 85 105 125

15

10

5

0

–5

–10

INPUT OFFSET VOLTAGE
DRIFT WILL TYPICALLY BE
WITHIN SHADED AREA

Figure 34. VOS Drift, X Input

TEMPERATURE – 8C

DEVIATION OF INPUT

OFFSET VOLTAGE – mV

60

–55 –35

–15 5 25 45 65 85 105 125

40

20

0

–20

–40

–60

INPUT OFFSET VOLTAGE
DRIFT WILL TYPICALLY BE
WITHIN SHADED AREA

Figure 35. VOS Drift, Z Input

Y1 FREQUENCY – MHz

OUTPUT AMPLITUDE – dB

10 1006020 30 40 50 70 80 90

–30

–20

0

–10

X1 FREQ =
Y1 FREQ –1MHz
(

E.G., Y

1

– X1 = 1MHz

FOR ALL CURVES)

U = 1V

U = 2V

U = 5V

U = 10V

Figure 33. Output Amplitude vs. Input
Frequency, When Used as Demodulator

SUPPLY VOLTAGE – 6V

S

OUTPUT SWING – Volts

–5

818

13

0

–10

–20

10

5

20

15

–15

9101112 14151617

Figure 32. Output Swing vs. Supply
Voltage

FREQUENCY – Hz

AMPLITUDE – dB

5
4

–5

100k 1M

10M

3
2

1

0
–1
–2
–3
–4

VS = 615V
X = 1.4V rms

Y = 10V
R

LOAD

= 500V

C

LOAD

= 20pF, 47pF,

100pF

INCREASING

C

LOAD

Figure 29. Gain vs. Frequency vs.
C

LOAD

FREQUENCY – Hz

PHASE SHIFT – Degrees

100k 1M 10M

0

–30
–60

–90
–120
–150
–180

VS = 615V
X = 1.4V rms
Y = 10V
R

LOAD

= 500V

C

LOAD

= 20pF, 47pF, 100pF

INCREASING

C

LOAD

Figure 30. Phase vs. Frequency
vs. C

LOAD

14-Lead Ceramic DIP (Q) Package14-Lead Plastic DIP (N) Package

0.210

(5.33)

MAX

0.160 (4.06)

0.115 (2.93)

0.795 (20.19)

0.725 (18.42)

0.022 (0.558)

0.014 (0.356)

0.100

(2.54)

BSC

PIN 1

0.280 (7.11)

0.240 (6.10)

0.325 (8.25)

0.300 (7.62)

0.015 (0.381)

0.008 (0.204)

0.195 (4.95)

0.115 (2.93)

0.070 (1.77)

0.045 (1.15)

SEATING
PLANE

0.060 (1.52)

0.015 (0.38)

0.130
(3.30)
MIN

7

8

14

1

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm).