RESISTOR MATCHING (%)
PERCENTAGE OF UNITS (%)
0.04
1996 TA01b
–0.02
0
0.02
40
35
30
25
20
15
10
5
0
–0.04
LT1996A
G = 81
FEATURES
■
Pin Configurable as a Difference Amplifier,
Inverting and Noninverting Amplifier
■
Difference Amplifier
Gain Range 9 to 117
CMRR >80dB
■
Noninverting Amplifier
Gain Range 0.008 to 118
■
Inverting Amplifier
Gain Range –0.08 to –117
■
Gain Error: <0.05%
■
Gain Drift: < 3ppm/°C
■
Wide Supply Range: Single 2.7V to Split ±18V
■
Micropower Operation: 100µA Supply
■
Input Offset Voltage: 50µV (Max)
■
Gain Bandwidth Product: 560kHz
■
Rail-to-Rail Output
■
Space Saving 10-Lead MSOP and DFN Packages
U
APPLICATIO S
LT1996
Precision, 100µA
Gain Selectable Amplifier
U
DESCRIPTIO
The LT®1996 combines a precision operational amplifier
with eight precision resistors to form a one-chip solution
for accurately amplifying voltages. Gains from –117 to
118 with a gain accuracy of 0.05% can be achieved without
any external components. The device is particularly well
suited for use as a difference amplifier, where the excellent
resistor matching results in a common mode rejection
ratio of greater than 80dB.
The amplifier features a 50µV maximum input offset
voltage and a gain bandwidth product of 560kHz. The
device operates from any supply voltage from 2.7V to 36V
and draws only 100µA supply current on a 5V supply. The
output swings to within 40mV of either supply rail.
The internal resistors have excellent matching characteristics; variation is 0.05% over temperature with a guaranteed matching temperature coefficent of less than 3ppm/°C.
The resistors are also extremely stable over voltage,
exhibiting a nonlinearity of less than 10ppm.
■
Handheld Instrumentation
■
Medical Instrumentation
■
Strain Gauge Amplifiers
■
Differential to Single-Ended Conversion
, LTC and LT are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners. Patents Pending.
TYPICAL APPLICATIO
V
M(IN)
∆V
IN
V
P(IN)
INPUT RANGE
±60V
= 100kΩ
R
IN
The LT1996 is fully specified at 5V and ±15V supplies and
from –40°C to 85°C. The device is available in space
saving 10-lead MSOP and DFN packages. For an amplifier
with selectable gains from –13 to 14, see the LT1991 data
sheet.
U
Rail-to-Rail Gain = 9 Difference Amplifier Distribution of Resistor Matching
V
= V
REF
1996 TA01
+ 9 • ∆V
IN
1996f
1
V
REF
OUT
SWING 40mV TO
EITHER RAIL
15V
450k/81
450k/27
–
+
450k/9
–
450k/9
+
450k/27
450k/81
450k
4pF
LT1996
450k
4pF
–15V
LT1996
WW
W
ABSOLUTE AXI U RATI GS
U
(Note 1)
Total Supply Voltage (V+ to V–) ............................... 40V
Input Voltage (Pins P9/M9, Note 2) ....................... ±60V
Input Current
(Pins P27/M27/P81/M81, Note 2) .................. ±10mA
Output Short-Circuit Duration (Note 3) ............ Indefinite
Operating Temperature Range (Note 4) ...–40°C to 85°C
Specified Temperature Range (Note 5) ....–40°C to 85°C
UUW
PACKAGE/ORDER I FOR ATIO
TOP VIEW
10
P9
1
P27
2
3
P81
4
V
EE
5
REF
10-LEAD (3mm × 3mm) PLASTIC DFN
UNDERSIDE METAL CONNECTED TO V
DD PACKAGE
T
= 125°C, θJA = 160°C/W
JMAX
(PCB CONNECTION OPTIONAL)
M9
M27
9
M81
8
7
V
CC
6
OUT
EE
ORDER PART
NUMBER
LT1996CDD
LT1996IDD
LT1996ACDD
LT1996AIDD
DD PART MARKING*
LBPC
Maximum Junction Temperature
DD Package ......................................................125°C
MS Package ..................................................... 150°C
Storage Temperature Range
DD Package .......................................–65°C to 125°C
MS Package ......................................–65°C to 150°C
MSOP–Lead Temperature (Soldering, 10 sec)...... 300°C
ORDER PART
NUMBER
TOP VIEW
1
P9
2
P27
3
P81
4
V
EE
REF
5
MS PACKAGE
10-LEAD PLASTIC MSOP
= 150°C, θJA = 230°C/W
JMAX
10
M9
9
M27
8
M81
7
V
CC
OUT
6
LT1996CMS
LT1996IMS
LT1996ACMS
LT1996AIMS
MS PART MARKING*T
LTBPB
*Temperature and electrical grades are identified by a label on the shipping container. Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The
● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Difference amplifier configuration, VS = 5V, 0V or ±15V;
VCM = V
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
∆G Gain Error VS = ±15V, V
GNL Gain Nonlinearity VS = ±15V; V
∆G/∆T Gain Drift vs Temperature (Note 6) VS = ±15V; V
CMRR Common Mode Rejection Ratio, VS = ±15V; G = 9; VCM = ±15.3V
= half supply, unless otherwise noted.
REF
Referred to Inputs (RTI) LT1996AMS
= ±10V; RL = 10k
G = 81; LT1996AMS
G = 27; LT1996AMS
G = 9; LT1996AMS
G = 81; LT1996ADD ● ±0.02 ±0.05 %
G = 27; LT1996ADD
G = 9; LT1996ADD
G = 81; LT1996 ● ±0.04 ±0.12 %
G = 27; LT1996
G = 9; LT1996
LT1996ADD
LT1996
VS = ±15V; G = 27; VCM = –14.5V to 14.3V
LT1996AMS
LT1996ADD
LT1996
OUT
= ±10V; RL = 10k; G = 9 ● 1 10 ppm
OUT
= ±10V; RL = 10k ● 0.3 3 ppm/°C
OUT
● ±0.02 ±0.05 %
● ±0.03 ±0.06 %
● ±0.03 ±0.07 %
● ±0.02 ±0.07 %
● ±0.03 ±0.08 %
● ±0.04 ±0.12 %
● ±0.04 ±0.12 %
● 80 100 dB
● 80 100 dB
● 70 100 dB
● 95 105 dB
● 90 105 dB
● 75 105 dB
1996f
2
LT1996
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Difference amplifier configuration, VS = 5V, 0V or ±15V;
= V
V
CM
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
CMRR Common Mode Rejection Ratio (RTI) VS = ±15V; G = 81; VCM = –14.1V to 13.9V
V
CM
V
OS
∆VOS/∆T Op Amp Offset Voltage Drift (Note 6) ● 0.3 1 µV/°C
I
B
I
OS
e
n
R
IN
= half supply, unless otherwise noted.
REF
Input Voltage Range (Note 7) P9/M9 Inputs
Op Amp Offset Voltage (Note 8) LT1996AMS, VS = 5V, 0V 15 50 µV
Op Amp Input Bias Current 2.5 5 nA
Op Amp Input Offset Current LT1996A 50 500 pA
Op Amp Input Noise Voltage 0.01Hz to 1Hz 0.35 µV
Input Noise Voltage Density G = 9; f = 1kHz 46 nV/√Hz
(Includes Resistor Noise) G = 117; f = 1kHz 18 nV/√Hz
Input Impedance (Note 10) P9 (M9 = Ground) ● 350 500 650 kΩ
LT1996AMS
LT1996ADD
LT1996
= ±15V; V
V
S
V
= 5V, 0V; V
S
= 3V, 0V; V
V
S
= 0V ● –15.5 15.3 V
REF
= 2.5V ● 0.84 3.94 V
REF
= 1.25V ● 0.98 1.86 V
REF
● 105 120 dB
● 100 120 dB
● 85 120 dB
P9/M9 Inputs, P81/M81 Connected to REF
V
= ±15V; V
S
= 5V, 0V; V
V
S
= 3V, 0V; V
V
S
= 0V ● –60 60 V
REF
= 2.5V ● –12.6 15.6 V
REF
= 1.25V ● –1.25 6.8 V
REF
P27/M27 Inputs
= ±15V; V
V
S
= 5V, 0V; V
V
S
V
= 3V, 0V; V
S
= 0V ● –14.5 14.3 V
REF
= 2.5V ● 0.95 3.84 V
REF
= 1.25V ● 1 1.82 V
REF
P81/M81 Inputs
= ±15V; V
V
S
= 5V, 0V; V
V
S
V
= 3V, 0V; V
S
= 0V ● –14.1 13.9 V
REF
= 2.5V ● 0.99 3.81 V
REF
= 1.25V ● 1 1.8 V
REF
● 135 µV
LT1996AMS, VS = ±15V 15 80 µV
● 160 µV
LT1996MS 25 100 µV
● 200 µV
LT1996DD 25 150 µV
● 250 µV
● 7.5 nA
● 750 pA
LT1996 50 1000 pA
● 1500 pA
0.01Hz to 1Hz 0.07 µV
0.1Hz to 10Hz 0.25 µV
0.1Hz to 10Hz 0.05 µV
P27 (M27 = Ground)
P81 (M81 = Ground)
● 326.9 467 607.1 kΩ
● 319.2 456 592.8 kΩ
P-P
RMS
P-P
RMS
M9 (P9 = Ground) ● 35 50 65 kΩ
M27 (P27 = Ground)
M81 (P81 = Ground)
● 11.69 16.7 21.71 kΩ
● 3.85 5.5 7.15 kΩ
1996f
3
LT1996
ELECTRICAL CHARACTERISTICS
temperature range, otherwise specifications are at T
VCM = V
= half supply, unless otherwise noted.
REF
The ● denotes the specifications which apply over the full operating
= 25°C. Difference amplifier configuration, VS = 5V, 0V or ±15V;
A
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
∆R Resistor Matching (Note 9) G = 81; LT1996AMS ● ±0.02 ±0.05 %
G = 27; LT1996AMS
G = 9; LT1996AMS
● ±0.03 ±0.06 %
● ±0.03 ±0.07 %
G = 81; LT1996ADD ● ±0.02 ±0.05 %
G = 27; LT1996ADD
G = 9; LT1996ADD
● ±0.02 ±0.07 %
● ±0.03 ±0.08 %
G = 81; LT1996 ● ±0.04 ±0.12 %
G = 27; LT1996
G = 9; LT1996
● ±0.04 ±0.12 %
● ±0.04 ±0.12 %
∆R/∆T Resistor Temperature Coefficient (Note 6) Resistor Matching ● 0.3 3 ppm/°C
Absolute Value
● –30 ppm/°C
PSRR Power Supply Rejection Ratio VS = ±1.35V to ±18V (Note 8) ● 105 135 dB
Minimum Supply Voltage ● 2.4 2.7 V
V
OUT
Output Voltage Swing (to Either Rail) No Load
= 5V, 0V 40 55 mV
V
S
V
= 5V, 0V ● 65 mV
S
= ±15V ● 110 mV
V
S
1mA Load
V
= 5V, 0V 150 225 mV
S
= 5V, 0V ● 275 mV
V
S
= ±15V ● 300 mV
V
S
I
SC
Output Short-Circuit Current (Sourcing) Drive Output Positive; 8 12 mA
Short Output to Ground
● 4mA
Output Short-Circuit Current (Sinking) Drive Output Negative; 8 21 mA
Short Output to V
or Midsupply ● 4mA
S
BW –3dB Bandwidth G = 9 38 kHz
G = 27 17 kHz
G = 81 7 kHz
GBWP Op Amp Gain Bandwidth Product f = 10kHz 560 kHz
tr, t
f
Rise Time, Fall Time G = 9; 0.1V Step; 10% to 90% 8 µs
G = 81; 0.1V Step; 10% to 90% 40 µs
t
S
SR Slew Rate VS = 5V, 0V; V
I
S
Settling Time to 0.01% G = 9; VS = 5V, 0V; 2V Step 85 µs
G = 9; V
G = 9; V
G = 9; V
V
= 5V, 0V; –2V Step 85 µs
S
= ±15V; 10V Step 110 µs
S
= ±15V; –10V Step 110 µs
S
= 1V to 4V ● 0.06 0.12 V/µs
= ±15V; V
S
OUT
= ±10V ● 0.08 0.12 V/µs
OUT
Supply Current VS = 5V, 0V 100 110 µA
● 150 µA
VS = ±15V 130 160 µA
● 210 µA
Note 1: Absolute Maximum Ratings are those beyond which the life of the
device may be impaired.
Note 2: The P27/M27 and P81/M81 inputs are protected by ESD diodes to
the supply rails. If one of these four inputs goes outside the rails, the input
current should be limited to less than 10mA. The P9/M9 inputs can
4
withstand ±60V if P81/M81 are grounded and VS = ±15V (see Applications
Information section about “High Voltage CM Difference Amplifiers”).
Note 3: A heat sink may be required to keep the junction temperature
below absolute maximum ratings.
1996f
GAIN (V/V)
9
INPUT OFFSET VOLTAGE (µV)
150
100
50
0
–50
–100
–150
10881 90 99 11745
1996 G06
18 27 72635436
VS = 5V, 0V
REPRESENTATIVE PARTS
ELECTRICAL CHARACTERISTICS
LT1996
Note 4: Both the LT1996C and LT1996I are guaranteed functional over the
–40°C to 85°C temperature range.
Note 5: The LT1996C is guaranteed to meet the specified performance
from 0°C to 70°C and is designed, characterized and expected to meet
specified performance from –40°C to 85°C but is not tested or QA
sampled at these temperatures. The LT1996I is guaranteed to meet
specified performance from –40°C to 85°C.
Note 6: This parameter is not 100% tested.
Note 7: Input voltage range is guaranteed by the CMRR test at V
= ±15V.
S
For the other voltages, this parameter is guaranteed by design and through
correlation with the ±15V test. See the Applications Information section to
determine the valid input voltage range under various operating
conditions.
Note 8: Offset voltage, offset voltage drift and PSRR are defined as
referred to the internal op amp. You can calculate output offset as follows.
In the case of balanced source resistance, V
I
• 450k + IB • 450k • (1 – RP/RN) where RP and RN are the total
OS
resistance at the op amp positive and negative terminal respectively.
Note 9: Resistors connected to the minus inputs. Resistor matching is not
tested directly, but is guaranteed by the gain error test.
Note 10: Input impedance is tested by a combination of direct
measurements and correlation to the CMRR and gain error tests.
UW
TYPICAL PERFOR A CE CHARACTERISTICS
Output Voltage Swing vs
Supply Current vs Supply Voltage
200
175
150
125
100
SUPPLY CURRENT (µA)
TA = 25°C
75
50
25
0
42 6 10 14 18
0
SUPPLY VOLTAGE (±V)
8
TA = 85°C
12
TA = –40°C
16
1996 G01
20
Temperature
VS = 5V, 0V
NO LOAD
60
40
OUTPUT LOW
OUTPUT VOLTAGE SWING (mV)
(LEFT AXIS)
20
V
EE
–50
050
–25 25 75 125
TEMPERATURE (°C)
OUTPUT HIGH
(RIGHT AXIS)
OS, OUT
(Difference Amplifier Configuration)
Output Voltage Swing vs Load
Current (Output Low)
V
CC
1400
–20
–40
–60
100
1996 G02
VS = 5V, 0V
1200
1000
800
600
400
OUTPUT VOLTAGE (mV)
200
V
EE
0
1
TA = 25°C
2
34
LOAD CURRENT (mA)
= VOS • Noise Gain +
TA = 85°C
TA = –40°C
1098765
1996 G03
Output Voltage Swing vs Load
Current (Output High)
V
CC
–100
–200
–300
–400
–500
–600
–700
–800
OUTPUT VOLTAGE SWING (mV)
–900
–1000
0123
TA = 85°C
TA = 25°C
4
5
LOAD CURRENT (mA)
VS = 5V, 0V
TA = –40°C
678910
1996 G04
Output Short-Circuit Current vs
Temperature
25
VS = 5V, 0V
20
15
10
SOURCING
5
OUTPUT SHORT-CIRCUIT CURRENT (mA)
0
–50
0
–25
TEMPERATURE (°C)
SINKING
50
25
75
100
Input Offset Voltage vs
Difference Gain
125
1996 G05
1996f
5
LT1996
UW
TYPICAL PERFOR A CE CHARACTERISTICS
Output Offset Voltage vs
Difference Gain Gain Error vs Load Current Slew Rate vs Temperature
10.0
7.5
5.0
2.5
0
–2.5
–5.0
OUTPUT OFFSET VOLTAGE (mV)
–7.5
–10.0
18 27 72635436
9
VS = 5V, 0V
REPRESENTATIVE PARTS
10881 90 99 11745
GAIN (V/V)
1996 G07
Bandwidth vs Gain CMRR vs Frequency PSRR vs Frequency
40
35
30
25
20
15
–3dB BANDWIDTH (kHz)
10
5
0
27 45 63 81
9
GAIN (V/V)
VS = 5V, 0V
= 25°C
T
A
99 11718 36 54 72 90 108
1996 G10
0.04
GAIN = 81
= ±15V
V
S
0.03
V
= ±10V
OUT
= 25°C
T
A
0.02
0.01
0
–0.01
GAIN ERROR (%)
–0.02
–0.03
–0.04
130
120
110
100
90
80
70
60
CMRR (dB)
50
40
30
20
10
0
12 4
0
GAIN = 81
GAIN = 27
GAIN = 9
10 1k 10k 1M
100 100k
REPRESENTATIVE UNITS
3
LOAD CURRENT (mA)
FREQUENCY (Hz)
(Difference Amplifier Configuration)
0.30
GAIN = 9
= ±15V
V
S
= ±10V
V
0.25
OUT
0.20
0.15
0.10
SLEW RATE (V/µs)
0.05
0
–50
–25 0
120
110
100
90
80
70
60
50
40
30
20
10
0
10 1k 10k
1996 G08
VS = 5V, 0V
= 25°C
T
A
1996 G11
5
PSRR (dB)
SR– (FALLING EDGE)
TEMPERATURE (°C)
100 100k
FREQUENCY (Hz)
SR+ (RISING EDGE)
50 100 125
25 75
GAIN = 9
GAIN = 27
1996 G09
VS = 5V, 0V
= 25°C
T
A
GAIN = 81
1996 G12
Output Impedance vs Frequency CMRR vs Temperature Gain Error vs Temperature
1000
VS = 5V, 0V
= 25°C
T
A
100
10
GAIN = 81
1
GAIN = 27
OUTPUT IMPEDANCE (Ω)
0.01
GAIN = 9
0.1
1 100 1k 100k10k
10
FREQUENCY (Hz)
1996 G13
120
GAIN = 9
V
100
80
60
CMRR (dB)
40
20
0
–50
= ±15V
S
–25 0
REPRESENTATIVE UNITS
50 100 125
25 75
TEMPERATURE (°C)
1996 G14
0.030
GAIN = 9
= ±15V
V
S
0.025
0.020
0.015
GAIN ERROR (%)
0.010
0.005
0
–50
–25 0
TEMPERATURE (°C)
6
REPRESENTATIVE UNITS
50 100 125
25 75
1996 G15
1996f
UW
TYPICAL PERFOR A CE CHARACTERISTICS
LT1996
(Difference Amplifier Configuration)
50
40
30
GAIN (dB)
20
10
0
0.5 10 100 5001
50mV/DIV
VS = 5V, 0V
TA = 25°C
GAIN = 81
GAIN = 27
GAIN = 9
FREQUENCY (kHz)
1996 G16
Small Signal Transient Response,
Gain = 9
Gain and Phase vs FrequencyGain vs Frequency
40
PHASE
(RIGHT AXIS)
30
20
GAIN
(LEFT AXIS)
GAIN (dB)
10
0
–10
0.1
50mV/DIV
1996 G17
400
0
–20
–40
–60
–80
–100
–120
–140
–160
–180
–200
1
FREQUENCY (kHz)
10 100
VS = 5V, 0V
TA = 25°C
GAIN = 9
Small Signal Transient Response,
Gain = 27
0.01Hz to 1Hz Voltage Noise
VS = ±15V
= 25°C
T
A
MEASURED IN G =117
REFERRED TO OP AMP INPUTS
PHASE (deg)
OP AMP VOLTAGE NOISE (100nV/DIV)
0 102030405060 70 8090100
Small Signal Transient Response,
Gain = 81
50mV/DIV
TIME (s)
1996 G21
10µs/DIV
U
UU
PI FU CTIO S
(Difference Amplifier Configuration)
1996 G18
P9 (Pin 1): Noninverting Gain-of-9 input. Connects a 50k
internal resistor to the op amp’s noninverting input.
P27 (Pin 2): Noninverting Gain-of-27 input. Connects a
(50k/3) internal resistor to the op amp’s noninverting input.
P81 (Pin 3): Noninverting Gain-of-81 input. Connects a
(50k/9) internal resistor to the op amp’s noninverting input.
VEE (Pin 4): Negative Power Supply. Can be either ground
(in single supply applications), or a negative voltage (in
split supply applications).
REF (Pin 5): Reference Input. Sets the output level when
difference between inputs is zero. Connects a 450k internal
20µs/DIV
1996 G19
50µs/DIV
1996 G20
resistor to the op amp’s noninverting input.
OUT (Pin 6): Output. V
(V
– VM3) + 81 • (VP9 – VM9).
P3
OUT
= V
+ 9 • (VP1 – VM1) + 27 •
REF
VCC (Pin 7): Positive Power Supply. Can be anything from
2.7V to 36V above the VEE voltage.
M81 (Pin 8): Inverting Gain-of-81 input. Connects a
(50k/9) internal resistor to the op amp’s inverting input.
M27 (Pin 9): Inverting Gain-of-27 input. Connects a
(50k/3) internal resistor to the op amp’s inverting input.
M9 (Pin 10): Inverting Gain-of-9 input. Connects a 50k
internal resistor to the op amp’s inverting input.
1996f
7
LT1996
BLOCK DIAGRA
W
9 8
M9 M27 M81
450k/81
450k/27
450k/9
450k/9
450k/27
450k/81
P9 P27 P81
2 3 4 5
1
U
WUU
APPLICATIO S I FOR ATIO
Introduction
The LT1996 may be the last op amp you ever have to stock.
Because it provides you with several precision matched
resistors, you can easily configure it into several different
classical gain circuits without adding external components. The several pages of simple circuits in this data
sheet demonstrate just how easy the LT1996 is to use. It
can be configured into difference amplifiers, as well as into
inverting and noninverting single ended amplifiers. The
fact that the resistors and op amp are provided together in
such a small package will often save you board space and
reduce complexity for easy probing.
The Op Amp
The op amp internal to the LT1996 is a precision device
with 15µV typical offset voltage and 3nA input bias cur-
rent. The input offset current is extremely low, so matching the source resistance seen by the op amp inputs will
provide for the best output accuracy. The op amp inputs
are not rail-to-rail, but extend to within 1.2V of VCC and 1V
7 610
4pF
–
+
4pF
OUT
V
450k
V
EE
CC
450k
OUT
LT1996
REF
1996 BD
of VEE. For many configurations though, the chip inputs
will function rail-to-rail because of effective attenuation to
the +input. The output is truly rail-to-rail, getting to within
40mV of the supply rails. The gain bandwidth product of
the op amp is about 560kHz. In noise gains of 2 or more,
it is stable into capacitive loads up to 500pF. In noise gains
below 2, it is stable into capacitive loads up to 100pF.
The Resistors
The resistors internal to the LT1996 are very well matched
SiChrome based elements protected with barrier metal.
Although their absolute tolerance is fairly poor (±30%),
their matching is to within 0.05%. This allows the chip to
achieve a CMRR of 80dB, and gain errors within 0.05%.
The resistor values are (450k/9), (450k/27), (450k/81)
and 450k, connected to each of the inputs. The resistors
have power limitations of 1watt for the 450k and (450k/81)
resistors, 0.3watt for the (450k/27) resistors and 0.5watt
for the (450k/9) resistors; however, in practice, power
dissipation will be limited well below these values by the
8
1996f
LT1996
U
WUU
APPLICATIO S I FOR ATIO
maximum voltage allowed on the input and REF pins. The
50k resistors connected to the M9 and P9 inputs are
isolated from the substrate, and can therefore be taken
beyond the supply voltages. The naming of the pins “P9,”
“P27,” “P81,” etc., is based on their admittances relative
to the feedback and REF admittances. Because it has 9
times the admittance, the voltage applied to the P9 input
has 9 times the effect of the voltage applied to the REF
input.
Bandwidth
The bandwidth of the LT1996 will depend on the gain you
select (or more accurately the noise gain resulting from
the gain you select). In the lowest configurable gain of 1,
the –3dB bandwidth is limited to 450kHz, with peaking of
about 2dB at 280kHz. In the highest configurable gains,
bandwidth is limited to 5kHz.
Input Noise
The LT1996 input noise is comprised of the Johnson noise
of the internal resistors (√4kTR), and the input voltage
noise of the op amp. Paralleling all four resistors to the
+input gives a 3.8kΩ resistance, for 8nV/√Hz of voltage
noise. The equivalent network on the –input gives another
8nV/√Hz, and the op amp 14nV/√Hz. Taking their RMS
sum gives a total 18nV/√Hz input referred noise floor.
Output noise depends on configuration and noise gain.
Input Resistance
The LT1996 input resistances vary with configuration, but
once configured are apparent on inspection. Note that
resistors connected to the op amp’s –input are looking
into a virtual ground, so they simply parallel. Any feedback
resistance around the op amp does not contribute to input
resistance. Resistors connected to the op amp’s +input
are looking into a high impedance, so they add as parallel
or series depending on how they are connected, and
whether or not some of them are grounded. The op amp
+input itself presents a very high GΩ impedance. In the
classical noninverting op amp configuration, the LT1996
presents the high input impedance of the op amp, as is
usual for the noninverting case.
Common Mode Input Voltage Range
The LT1996 valid common mode input range is limited by
three factors:
1. Maximum allowed voltage on the pins
2. The input voltage range of the internal op amp
3. Valid output voltage
The maximum voltage allowed on the P27, M27, P81 and
M81 inputs includes the positive and negative supply plus
a diode drop. These pins should not be driven more than
a diode drop outside of the supply rails. This is because
they are connected through diodes to internal manufacturing post-package trim circuitry, and through a substrate
diode to VEE. If more than 10mA is allowed to flow through
these pins, there is a risk that the LT1996 will be detrimmed
or damaged. The P9 and M9 inputs do not have clamp
diodes or substrate diodes or trim circuitry and can be
taken well outside the supply rails. The maximum allowed
voltage on the P9 and M9 pins is ±60V.
The input voltage range of the internal op amp extends to
within 1.2V of VCC and 1V of VEE. The voltage at which the
op amp inputs common mode is determined by the
voltage at the op amp’s +input, and this is determined by
the voltages on pins P9, P27, P81 and REF. (See “Calculating Input Voltage Range” section.) This is true provided
that the op amp is functioning and feedback is maintaining
the inputs at the same voltage, which brings us to the third
requirement.
For valid circuit function, the op amp output must not be
clipped. The output will clip if the input signals are attempting to force it to within 40mV of its supply voltages. This
usually happens due to too large a signal level, but it can
also occur with zero input differential and must therefore
be included as an example of a common mode problem.
1996f
9
LT1996
U
WUU
APPLICATIO S I FOR ATIO
Consider Figure 1. This shows the LT1996 configured as
a gain of 117 difference amplifier on a single supply with
5V
7
450k/81
8
450k/27
9
450k/9
10
–
V
DM
0V
450k/9
+
1
V
CM
2.5V
2
3
450k/27
450k/81
Figure 1. Difference Amplifier Cannot Produce 0V on a Single
Supply. Provide a Negative Supply, or Raise Pin 5, or Provide
400µV of V
DM
the output REF connected to ground. This is a great circuit,
but it does not support VDM = 0V at any common mode
because the output clips into ground while trying to
produce 0V
. It can be fixed simply by declaring the
OUT
valid input differential range not to extend below +0.4mV,
or by elevating the REF pin above 40mV, or by providing
a negative supply.
Calculating Input Voltage Range
450k
4pF
4pF
–
+
4pF
4pF
450k
4
REF
LT1996
1996 F01
6
V
= 117 • V
OUT
5
DM
R
F
V
R
G
V
EXT
V
INT
R
G
CC
–
+
V
EE
V
REF
R
F
1996 F02
Figure 2. Calculating CM Input Voltage Range
These two voltages represent the high and low extremes
of the common mode input range, if the other limits have
not already been exceeded (1 and 3, above). In most cases,
the inverting inputs M9 through M81 can be taken further
than these two extremes because doing this does not
move the op amp input common mode. To calculate the
limit on this additional range, see Figure 3. Note that, with
R
F
V
R
V
MORE
MAX OR MIN
G
V
V
EXT
INT
R
G
Figure 3. Calculating Additional Voltage Range of
Inverting Inputs
CC
–
+
V
EE
V
REF
R
F
1996 F03
Figure 2 shows the LT1996 in the generalized case of a
difference amplifier, with the inputs shorted for the common mode calculation. The values of RF and RG are
dictated by how the P inputs and REF pin are connected.
By superposition we can write:
V
= V
INT
Or, solving for V
V
EXT
But valid V
• (RF/(RF + RG)) + V
EXT
:
EXT
= V
• (1 + RG/RF) – V
INT
voltages are limited to VCC – 1.2V and V
INT
• (RG/(RF + RG))
REF
• RG/R
REF
F
+
EE
1V, so:
MAX V
EXT
= (V
– 1.2) • (1 + RG/RF) – V
CC
REF
• RG/R
F
and:
MIN V
EXT
= (V
+ 1) • (1 + RG/RF) – V
EE
REF
• RG/R
F
10
V
= 0, the op amp output is at V
MORE
V
(the high cm limit), as V
EXT
MORE
amp output will go more negative from V
V
MORE
V
OUT
• RF/RG, so:
= V
REF
– V
MORE
• RF/R
G
. From the max
REF
goes positive, the op
by the amount
REF
Or:
V
= (V
MORE
The most negative that V
Max V
MORE
REF
= (V
– V
) • RG/R
OUT
OUT
– VEE – 0.04V) • RG/R
REF
F
can go is VEE + 0.04V, so:
F
(should be positive)
The situation where this function is negative, and therefore
problematic, when V
= 0 and VEE = 0, has already been
REF
dealt with in Figure 1. The strength of the equation is
demonstrated in that it provides the three solutions
1996f
LT1996
U
WUU
APPLICATIO S I FOR ATIO
suggested in Figure 1: raise V
some negative V
MORE
.
Likewise, from the lower common mode extreme, making
the negative input more negative will raise the output
voltage, limited by V
MORE
= (V
MIN V
– 0.04V.
CC
– VCC + 0.04V) • RG/R
REF
(should be negative)
Again, the additional input range calculated here is only
available provided the other remaining constraint is not
violated, the maximum voltage allowed on the pin.
The Classical Noninverting Amplifier: High Input Z
Perhaps the most common op amp configuration is the
noninverting amplifier. Figure 4 shows the textbook
R
G
CLASSICAL NONINVERTING OP AMP CONFIGURATION.
YOU PROVIDE THE RESISTORS.
450k/81
8
450k/27
9
450k/9
10
450k/9
1
450k/27
2
450k/81
3
V
IN
CLASSICAL NONINVERTING OP AMP CONFIGURATION
IMPLEMENTED WITH LT1991. R
GAIN IS ACHIEVED BY GROUNDING, FLOATING OR FEEDING BACK
THE AVAILABLE RESISTORS TO ARRIVE AT DESIRED R
WE PROVIDE YOU WITH <0.1% RESISTORS.
–
V
+
IN
, lower VEE, or provide
REF
R
F
V
OUT
V
= GAIN • V
OUT
GAIN = 1 + RF/R
450k
4pF
4pF
IN
G
–
+
4pF
4pF
450k
LT1996
5
= 45k, RG = 5.6k, GAIN = 9.1.
F
F
6
AND RG.
F
V
OUT
1996 F04
representation of the circuit on the top. The LT1996 is
shown on the bottom configured in a precision gain of 9.1.
One of the benefits of the noninverting op amp configuration is that the input impedance is extremely high. The
LT1996 maintains this benefit. Given the finite number of
available feedback resistors in the LT1996, the number of
gain configurations is also finite. The complete list of such
Hi-Z input noninverting gain configurations is shown in
Table 1. Many of these are also represented in Figure 5 in
schematic form. Note that the P-side resistor inputs have
been connected so as to match the source impedance
seen by the internal op amp inputs. Note also that gain and
noise gain are identical, for optimal precision.
Table 1. Configuring the M Pins for Simple Noninverting Gains.
The P Inputs are driven as shown in the examples on the next
page
M81, M27, M9 Connection
Gain M81 M27 M9
1 Output Output Output
1.08 Output Output Grounded
1.11 Output Float Grounded
1.30 Output Grounded Output
1.32 Float Output Grounded
1.33 Output Grounded Float
1.44 Output Grounded Grounded
3.19 Grounded Output Output
3.7 Float Grounded Output
3.89 Grounded Output Float
4.21 Grounded Output Grounded
9.1 Grounded Float Output
10 Float Float Grounded
11.8 Grounded Grounded Output
28 Float Grounded Float
37 Float Grounded Grounded
82 Grounded Float Float
91 Grounded Float Grounded
109 Grounded Grounded Float
118 Grounded Grounded Grounded
Figure 4. The LT1996 as a Classical Noninverting Op Amp
1996f
11
LT1996
U
WUU
APPLICATIO S I FOR ATIO
+
V
8
M81
9
M27
10
M9
1
P9
2
P27
V
V
IN
3
IN
P81
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
S
7
V
LT1996
V
EE
CC
REF
OUT
5
6
V
OUT
V
IN
4
–
V
S
GAIN = 1 GAIN = 10 GAIN = 3.893
+
V
S
7
V
LT1996
V
EE
CC
REF
OUT
5
6
V
OUT
4
–
V
S
V
IN
GAIN = 28 GAIN = 37 GAIN = 9.1
V
LT1996
V
EE
4
–
V
S
V
LT1996
V
EE
4
–
V
S
+
V
S
7
CC
REF
OUT
6
V
OUT
5
+
V
S
7
CC
REF
OUT
6
V
OUT
5
+
V
V
LT1996
V
EE
4
–
V
S
V
LT1996
V
EE
4
–
V
S
S
7
CC
REF
OUT
6
V
OUT
5
V
+
V
S
7
CC
REF
OUT
6
V
OUT
5
V
IN
8
M81
9
M27
10
M9
1
P9
2
P27
3
IN
P81
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
LT1996
V
EE
V
S
GAIN = 91
V
OUT
+
V
S
7
V
CC
REF
OUT
6
V
OUT
5
4
–
1996 F05
+
V
V
LT1996
V
EE
4
–
V
S
S
7
CC
REF
OUT
6
V
OUT
5
V
IN
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
8
M81
9
M27
10
M9
1
P9
2
P27
V
3
IN
P81
V
LT1996
V
EE
4
–
V
S
+
V
S
7
CC
REF
OUT
6
V
OUT
5
V
IN
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
GAIN = 11.8 GAIN = 82
V
LT1996
V
EE
4
–
V
S
+
V
S
7
CC
REF
OUT
6
5
+
V
8
M81
9
M27
10
M9
1
P9
2
P27
V
3
IN
P81
S
7
V
LT1996
V
EE
CC
REF
OUT
5
6
V
OUT
4
–
V
S
GAIN = 109 GAIN = 118
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
V
IN
Figure 5. Some Implementations of Classical Noninverting
Gains Using the LT1996. High Input Z Is Maintained
12
1996f
LT1996
U
WUU
APPLICATIO S I FOR ATIO
Attenuation Using the P Input Resistors
Attenuation happens as a matter of fact in difference
amplifier configurations, but it is also used for reducing
peak signal level or improving input common mode range
even in single ended systems. When signal conditioning
indicates a need for attenuation, the LT1996 resistors are
ready at hand. The four precision resistors can provide
several attenuation levels, and these are tabulated in
Table 2 as a design reference.
V
IN
R
A
V
INT
R
G
V
A = RG/(RA + RG)
CLASSICAL ATTENUATOR
Figure 6. LT1996 Provides for Easy Attenuation to the Op Amp’s
+Input. The P9 Input Can Be Taken Well Outside of the Supplies
Because the attenuations and the noninverting gains are
set independently, they can be combined. This provides
high gain resolution, about 700 unique gains between
0.0085 and 118, as plotted in Figure 7. This is too large a
number to tabulate, but the designer can calculate achievable gain by taking the vector product of the gains and
attenuations in Tables 1 and 2, and seeking the best match.
Average gain resolution is 1.5%, with worst case steps of
about 50% as seen in Figure 7.
1000
GAIN
0.001
Figure 7. Over 600 Unique Gain Settings Achievable with the
LT1996 by Combining Attenuation with Noninverting Gain
INT
100
0.01
10
0.1
= A • V
1
0
OKAY UP
TO ±60V
V
IN
IN
100 200 400
450k/9
1
450k/27
2
450k/81
3
LT1991 ATTENUATING TO THE +INPUT BY
DRIVING AND GROUNDING AND FLOATING
INPUTS R
300
COUNT
V
INT
+
450k
LT1996
5
= 50k, RG = 50k/9, SO A = 0.1.
A
600
500 700
1996 F07
1996 F06
Table 2. Configuring the P Pins for Various Attenuations. Those
Shown in Bold Are Functional Even When the Input Drive
Exceeds the Supplies
P81, P27, P9, REF Connection
A P81 P27 P9 REF
0.0085 Grounded Grounded Grounded Driven
0.0092 Grounded Grounded Float Driven
0.0110 Grounded Float Grounded Driven
0.0122 Grounded Float Float Driven
0.0270 Float Grounded Grounded Driven
0.0357 Float Grounded Float Driven
0.0763 Grounded Grounded Driven Grounded
0.0769 Grounded Grounded Driven Float
0.0847 Grounded Grounded Driven Driven
0.0989 Grounded Float Driven Grounded
0.1 Grounded Float Driven Float
0.110 Grounded Float Driven Driven
0.229 Grounded Driven Grounded Grounded
0.231 Grounded Driven Grounded Float
0.237 Grounded Driven Grounded Driven
0.243 Float Grounded Driven Grounded
0.248 Grounded Driven Float Grounded
0.25 Float Grounded Driven Float
0.25 Grounded Driven Float Float
0.257 Grounded Driven Float Driven
0.270 Float Grounded Driven Driven
0.305 Grounded Driven Driven Grounded
0.308 Grounded Driven Driven Float
0.314 Grounded Driven Driven Driven
0.686 Driven Grounded Grounded Grounded
0.692 Driven Grounded Grounded Float
0.695 Driven Grounded Grounded Driven
0.730 Float Driven Grounded Grounded
0.743 Driven Grounded Float Grounded
0.75 Float Driven Grounded Float
0.752 Driven Grounded Float Driven
0.757 Float Driven Grounded Driven
0.763 Driven Grounded Driven Grounded
0.769 Driven Grounded Driven Float
0.771 Driven Grounded Driven Driven
0.890 Driven Float Grounded Grounded
0.9 Float Float Driven Grounded
0.901 Driven Float Grounded Driven
0.915 Driven Driven Grounded Grounded
0.923 Driven Driven Grounded Float
0.924 Driven Driven Grounded Driven
0.964 Float Driven Float Grounded
0.973 Float Driven Driven Grounded
0.988 Driven Float Float Grounded
0.989 Driven Float Driven Grounded
0.991 Driven Driven Float Grounded
0.992 Driven Driven Driven Grounded
1996f
13
LT1996
U
WUU
APPLICATIO S I FOR ATIO
Inverting Configuration
The inverting amplifier, shown in Figure 8, is another
classical op amp configuration. The circuit is actually
identical to the noninverting amplifier of Figure 4, except
that VIN and GND have been swapped. The list of available
gains is shown in Table 3, and some of the circuits are
shown in Figure 9. Noise gain is 1+|Gain|, as is the usual
case for inverting amplifiers. Again, for the best DC performance, match the source impedance seen by the op amp
inputs.
R
F
R
G
IN
CLASSICAL INVERTING OP AMP CONFIGURATION.
YOU PROVIDE THE RESISTORS.
450k/81
8
450k/27
9
–
+
V
= GAIN • V
OUT
GAIN = – RF/R
450k
4pF
4pF
V
OUT
IN
G
V
(DRIVE)
V
IN
Table 3. Configuring the M Pins for Simple Inverting Gains
M81, M27, M9 Connection
Gain M81 M27 M9
–0.083 Output Output Drive
–0.110 Output Float Drive
–0.297 Output Drive Output
–0.321 Float Output Drive
–0.329 Output Drive Float
–0.439 Output Drive Drive
–2.19 Drive Output Output
–2.7 Float Drive Output
–2.89 Drive Output Float
–3.21 Drive Output Drive
–8.1 Drive Float Output
–9 Float Float Drive
–10.8 Drive Drive Output
–27 Float Drive Float
–36 Float Drive Drive
–81 Drive Float Float
–90 Drive Float Drive
–108 Drive Drive Float
–117 Drive Drive Drive
450k/9
10
450k/9
1
450k/27
2
450k/81
3
CLASSICAL INVERTING OP AMP CONFIGURATION IMPLEMENTED
WITH LT1991. R
GAIN IS ACHIEVED BY GROUNDING, FLOATING OR FEEDING BACK
THE AVAILABLE RESISTORS TO ARRIVE AT DESIRED R
WE PROVIDE YOU WITH <0.1% RESISTORS.
= 45k, RG = 5.55k, GAIN = –8.1.
F
–
+
450k
4pF
4pF
5
LT1996
6
AND RG.
F
V
OUT
1996 F08
Figure 8. The LT1996 as a Classical Inverting Op Amp. Note the
Circuit Is Identical to the Noninverting Amplifier, Except that V
IN
and Ground Have Been Swapped
1996f
14
LT1996
U
WUU
APPLICATIO S I FOR ATIO
+
V
8
M81
9
M27
V
IN
10
M9
1
P9
2
P27
3
P81
8
M81
9
V
IN
M27
10
M9
1
P9
2
P27
3
P81
S
7
V
LT1996
V
EE
CC
REF
OUT
6
V
OUT
5
V
IN
4
–
V
S
GAIN = –0.321 GAIN = –9 GAIN = –2.89
+
V
S
7
V
LT1996
V
EE
CC
REF
OUT
6
V
OUT
5
V
IN
4
–
V
S
GAIN = –27 GAIN = –36 GAIN = –8.1
V
LT1996
V
EE
4
–
V
S
V
LT1996
V
EE
4
–
V
S
+
V
S
7
CC
REF
OUT
6
V
OUT
5
+
V
S
7
CC
REF
OUT
6
V
OUT
5
+
V
LT1996
V
EE
4
–
V
S
LT1996
V
EE
4
–
V
S
S
7
V
CC
REF
OUT
6
V
OUT
5
+
V
S
V
7
V
CC
REF
OUT
6
IN
V
OUT
5
8
V
IN
M81
9
M27
10
M9
1
P9
2
P27
3
P81
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
V
LT1996
V
EE
4
–
V
S
GAIN = –90
V
OUT
+
V
S
7
CC
REF
OUT
6
V
OUT
5
1996 F09
+
V
V
LT1996
V
EE
4
–
V
S
S
7
CC
REF
OUT
6
V
OUT
5
8
V
IN
M81
9
M27
10
M9
1
P9
2
P27
3
P81
V
IN
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
LT1996
V
EE
4
–
V
S
+
V
S
V
7
V
CC
REF
OUT
6
IN
V
OUT
5
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
GAIN = –10.8 GAIN = –81
V
LT1996
V
EE
4
–
V
S
+
V
S
7
CC
REF
OUT
6
5
+
V
LT1996
V
EE
4
–
V
S
S
7
V
CC
REF
OUT
6
V
OUT
5
8
M81
9
M27
10
V
IN
M9
1
P9
2
P27
3
P81
8
V
IN
M81
9
M27
10
M9
1
P9
2
P27
3
P81
GAIN = –108 GAIN = –117
Figure 9. It Is Simple to Get Precision Inverting Gains with the LT1996.
Input Impedance Varies from 3.8kΩ (Gain = –117) to 50kΩ (Gain = –9)
1996f
15
LT1996
U
WUU
APPLICATIO S I FOR ATIO
Difference Amplifiers
The resistors in the LT1996 allow it to easily make difference amplifiers also. Figure 10 shows the basic 4-resistor
difference amplifier and the LT1996. A difference gain of
27 is shown, but notice the effect of the additional dashed
connections. By connecting the 50k resistors in parallel,
the gain is reduced by a factor of 10. Of course, with so
many resistors, there are many possible gains. Table 4
shows the difference gains and how they are achieved.
Note that, as for inverting amplifiers, the noise gain is 1
more than the signal gain.
Table 4. Connections Giving Difference Gains for the LT1996
Gain V
+
IN
0.083 P9 M9 M27, M81 P27, P81
0.110 P9 M9 M81 P81
0.297 P27 M27 M9, M81 P9, P81
0.321 P9 M9 M27 P27
0.329 P27 M27 M81 P81
0.439 P9, P27 M9, M27 M81 P81
2.189 P81 M81 M9, M27 P9, P27
2.700 P27 M27 M9 P9
2.893 P81 M81 M27 P27
3.214 P9, P81 M9, M81 M27 P27
8.1 P81 M81 M9 P9
9 P9 M9
10.8 P27, P81 M27, M81 M9 P9
27 P27 M27
36 P9, P27 M9, M27
81 P81 M81
90 P9, P81 M9, M81
108 P27, P81 M27, M81
117 P9, P27, P81 M9, M27, M81
–
V
IN
Output GND (REF)
R
F
R
G
–
V
PARALLEL
TO CHANGE
, R
R
F
V
IN
R
G
+
V
IN
CLASSICAL DIFFERENCE AMPLIFIER USING THE LT1991
450k/81
8
450k/27
–
IN
G
V
IN
9
450k/9
10
450k/9
1
450k/27
2
+
450k/81
3
CLASSICAL DIFFERENCE AMPLIFIER IMPLEMENTED
WITH LT1991. R
ADDING THE DASHED CONNECTIONS CONNECTS THE
TWO 450k RESISTOR IN PARALLEL, SO RF IS REDUCED
TO 45k. GAIN BECOMES 45k/16.7k = 2.7.
–
+
R
F
V
= GAIN • (V
OUT
GAIN = R
450k
4pF
4pF
–
+
4pF
4pF
450k
= 450k, RG = 16.7k, GAIN = 3.
F
F/RG
V
OUT
LT1996
+
–
– V
)
IN
IN
6
V
OUT
5
1996 F10
Figure 10. Difference Amplifier Using the LT1996. Gain Is Set
Simply by Connecting the Correct Resistors or Combinations of
Resistors. Gain of 27 Is Shown, with Dashed Lines Modifying It
to Gain of 2.7. Noise Gain Is Optimal
16
1996f
LT1996
U
WUU
APPLICATIO S I FOR ATIO
+
V
8
M81
9
M27
–
V
IN
+
V
IN
10
M9
1
P9
2
P27
3
P81
8
–
V
IN
+
V
IN
M81
9
M27
10
M9
1
P9
2
P27
3
P81
S
7
V
LT1996
V
EE
CC
REF
OUT
6
V
5
OUT
V
IN
V
IN
4
–
V
S
GAIN = 0.321
+
V
S
V
LT1996
V
EE
7
CC
REF
OUT
6
V
5
OUT
V
IN
V
IN
4
–
V
S
GAIN = 27 GAIN = 36 GAIN = 8.1
V
LT1996
V
EE
4
–
V
S
+
V
S
7
CC
REF
OUT
6
V
OUT
5
+
V
LT1996
V
EE
4
–
V
S
S
7
V
CC
REF
OUT
6
V
OUT
5
8
–
V
IN
V
IN
M81
9
M27
10
M9
1
P9
2
P27
3
+
P81
8
M81
9
M27
10
–
M9
1
+
P9
2
P27
3
P81
GAIN = 9 GAIN = 2.89
V
LT1996
V
EE
4
–
V
S
+
V
S
7
CC
REF
OUT
6
V
OUT
5
+
V
LT1996
V
EE
4
–
V
S
S
7
V
CC
REF
OUT
6
V
5
OUT
–
V
IN
+
V
IN
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
8
M81
–
9
M27
10
M9
1
P9
+
2
P27
3
P81
LT1996
V
EE
V
S
GAIN = 90
V
OUT
+
V
S
7
V
CC
REF
OUT
6
V
OUT
5
4
–
1996 F11
+
V
V
LT1996
V
EE
4
–
V
S
S
7
CC
REF
OUT
6
V
OUT
5
–
V
V
8
IN
+
IN
M81
9
M27
10
M9
1
P9
2
P27
3
P81
–
V
IN
+
V
IN
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
LT1996
V
EE
4
–
V
S
+
V
S
7
V
CC
REF
OUT
6
V
5
OUT
–
V
IN
+
V
IN
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
GAIN = 10.8 GAIN = 81
V
LT1996
V
EE
4
–
V
S
+
V
S
7
CC
REF
OUT
6
5
+
V
V
LT1996
V
EE
4
–
V
S
S
7
CC
REF
OUT
6
V
OUT
5
–
V
V
8
IN
+
IN
M81
9
M27
10
M9
1
P9
2
P27
3
P81
–
V
V
8
IN
+
IN
M81
9
M27
10
M9
1
P9
2
P27
3
P81
GAIN = 108 GAIN = 117
Figure 11. Many Difference Gains Are Achievable Just by Strapping the Pins
1996f
17
LT1996
U
WUU
APPLICATIO S I FOR ATIO
R
F
R
G
–
V
IN
+
V
IN
–
R
G
+
V
= GAIN • (V
OUT
GAIN = R
R
F
CLASSICAL DIFFERENCE AMPLIFIER
F/RG
V
OUT
+
–
– V
)
IN
IN
Figure 12. Another Method of Selecting Difference Gain Is “Cross-Coupling.”
The Additional Method Means the LT1996 Provides Extra Integer Gains
Difference Amplifier: Additional Integer Gains Using
Cross-Coupling
Figure 12 shows the basic difference amplifier as well as
the LT1996 in a difference gain of 27. But notice the effect
of the additional dashed connections. This is referred to as
“cross-coupling” and has the effect of reducing the differential gain from 27 to 18. Using this method, additional
integer gains are achievable, as shown in Table 5 below.
Note that the equations can be written by inspection from
the V
+
connections, and that the V
IN
–
connections are
IN
simply the opposite (swap P for M and M for P). The
method is the same as for the LT1991, except that the
LT1996 applies a multiplier of 9. Noise gain, bandwidth,
and input impedance specifications for the various cases
are also tabulated, as these are not obvious. Schematics
are provided in Figure 13.
Table 5. Connections Using Cross-Coupling. Note That Equations
Can Be Written by Inspection of the V
Gain V
+
IN
V
IN
18 P27, M9 M27, P9 27 – 9 39 14 46 16
45 P81, M27, M9 M81, P27, P9 81 – 27 – 9 117 5 12 6
54 P81, M27 M81, P27 81 – 27 108 5 16 6
63 P81, P9, M27 M81, M9, P27 81 + 9 – 27 117 5 16 5
72 P81, M9 M81, P9 81 – 9 90 6 45 6
99 P81, P27, M9 M81, M27, P9 81 + 27 – 9 117 5 45 4
Gain Noise –3dB BW R
–
Equation Gain kHz Typ kΩ Typ kΩ
+
Column
IN
+
IN
V
COUPLING
–
R
IN
450k/81
8
450k/27
–
IN
CROSS-
V
IN
9
450k/9
10
450k/9
1
450k/27
2
+
450k/81
3
CLASSICAL DIFFERENCE AMPLIFIER IMPLEMENTED
WITH LT1991. R
GAIN CAN BE ADJUSTED BY "CROSS COUPLING." MAKING THE
DASHED CONNECTIONS REDUCE THE GAIN FROM 3 T0 2.
WHEN CROSS COUPLING, SEE WHAT IS CONNECTED TO THE
+
VOLTAGE. CONNECTING P27 AND M9 GIVES 27 – 9 = 18.
V
IN
CONNECTIONS TO V
–
V
IN
+
V
IN
–
V
IN
+
V
IN
–
V
IN
+
V
IN
= 450k, RG = 16.7k, GAIN = 27.
F
–
ARE SYMMETRIC: M27 AND P9.
IN
8
9
10
1
2
3
8
9
10
1
2
3
8
9
10
1
2
3
V
M81
M27
V
M9
LT1996
P9
P27
V
EE
P81
4
–
V
S
GAIN = 18 GAIN = 54
M81
M27
V
M9
LT1996
P9
P27
V
EE
P81
4
–
V
S
GAIN = 45
V
M81
M27
V
M9
LT1996
P9
P27
V
EE
P81
4
–
V
S
GAIN = 72
450k
4pF
4pF
–
+
4pF
4pF
450k
+
S
7
CC
V
S
7
CC
+
S
7
CC
6
OUT
REF
5
+
6
OUT
REF
5
6
OUT
REF
5
Figure 13. Integer Gain Difference
Amplifiers Using Cross-Coupling
6
V
OUT
5
LT1996
1996 F10
+
V
M81
M27
M9
P9
P27
P81
M81
M27
M9
P9
P27
P81
M81
M27
M9
P9
P27
P81
GAIN = 99
V
LT1996
V
EE
4
–
V
S
V
V
LT1996
V
EE
4
–
V
S
GAIN = 63
V
V
LT1996
V
EE
4
–
V
S
S
7
CC
+
S
7
CC
+
S
7
CC
REF
REF
REF
5
5
5
OUT
OUT
OUT
6
V
OUT
6
6
V
OUT
1996 F13
IN
V
OUT
V
IN
V
IN
V
OUT
V
IN
V
IN
V
OUT
V
IN
9
10
1
2
+
3
–
8
9
10
1
2
3
+
–
8
9
10
1
2
+
3
8
–
V
18
1996f
LT1996
U
WUU
APPLICATIO S I FOR ATIO
High Voltage CM Difference Amplifiers
This class of difference amplifier remains to be discussed.
Figure 14 shows the basic circuit on the top. The effective
input voltage range of the circuit is extended by the fact
that resistors RT attenuate the common mode voltage
seen by the op amp inputs. For the LT1996, the most
useful resistors for RG are the M9 and P9 50kΩ resistors,
because they do not have diode clamps to the supplies and
therefore can be taken outside the supplies. As before, the
input CM of the op amp is the limiting factor and is set by
the voltage at the op amp +input, V
we can write:
V
= V
INT
(RF + RG||RT) + V
Solving for V
V
= (1 + RG/(RF||RT)) • (V
EXT
(RF + RG||RT) – V
• (RF||RT)/(RG + RF||RT) + V
EXT
• (RF||RG)/(RT + RF||RG)
TERM
:
EXT
INT
• (RF||RG)/(RT + RF||RG))
TERM
Given the values of the resistors in the LT1996, this
equation has been simplified and evaluated, and the resulting equations provided in Table 6. As before, substituting VCC – 1.2 and VEE + 1 for V
upper and lower common mode extremes respectively.
Following are sample calculations for the case shown in
Figure 14, right-hand side. Note that P81 and M81 are
terminated so row 3 of Table 6 provides the equation:
MAX V
= 91/9 • (VCC – 1.2V) – V
EXT
= (10.11) • (10.8) – 0.11(2.5) – 9(10) =
18.9V
and:
MIN V
= 91/9 • (VEE + 1V) – V
EXT
= (10.11)(1) – 0.11(2.5) – 9(10) = –80.2V
. By superposition
INT
• (RG||RT)/
REF
– V
LIM
• (RG||RT)/
REF
will give the valid
/9 – 9 • V
REF
/9 – 9 • V
REF
TERM
TERM
Table 6. HighV CM Connections Giving Difference Gains
for the LT1996
Noise (Substitute V
R
Gain V
T
Gain V
+
–
V
IN
IN
9 P9 M9 10 10/9 • V
9 P9 M9 P27, M27 37 37/9 • V
9 P9 M9 P81, M81 91 91/9 • V
9 P9 M9 P27||P81 118 118/9 • V
LIM
LIM
LIM
M27||M81
R
F
V
CC
–
+
CONNECTED TO V
V
R
12V
EE
F
7
V
OUT
GAIN = R
450k
–
+
450k
4
4pF
4pF
4pF
4pF
–
V
IN
+
V
IN
(= V
)
EXT
+
–
V
V
IN
IN
INPUT CM RANGE
= –60V TO 18.9V
R
G
R
G
R
10V
V
R
TERM
8
9
10
1
2
3
T
T
HIGH CM VOLTAGE DIFFERENCE AMPLIFIER
INPUT CM TO OP AMP IS ATTENUATED BY
RESISTORS R
T
450k/81
450k/27
450k/9
450k/9
450k/27
450k/81
Max, Min V
EE
= GAIN • (V
TERM.
+ 1 for V
LIM
– V
REF
– V
REF
– V
REF
F/RG
LT1996
EXT
– 1.2,
CC
)
LIM
- V
REF
/9 – 3 • V
/9 – 9 • V
/9 – 12 • V
V
OUT
+
– V
IN
V
REF
REF
/9
TERM
TERM
TERM
–
)
IN
6
V
OUT
5
2.5V
but this exceeds the 60V absolute maximum rating of the
P9, M9 pins, so –60V becomes the de facto negative
common mode limit. Several more examples of high CM
circuits are shown in Figures 15, 16, 17 for various
supplies.
HIGH NEGATIVE CM VOLTAGE DIFFERENCE AMPLIFIER
IMPLEMENTED WITH LT1996.
= 450k, RG = 50k, RT 5.55k, GAIN = 9
R
F
V
= 10V = VCC = 12V, V
TERM
= 2.5V, VEE = 0V.
REF
Figure 14. Extending CM Input Range
1996 F14
1996f
19
LT1996
U
WUU
APPLICATIO S I FOR ATIO
8
M81
9
M27
10
–
V
IN
+
V
IN
–
V
IN
+
V
IN
M9
1
P9
2
P27
3
P81
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
1.25V
3V 3V
7
V
LT1996
V
EE
CC
REF
5
OUT
1.25V
6
V
OUT
V
IN
V
IN
4
V
= 0.97V TO 1.86V
CM
3V
7
V
LT1996
V
EE
4
V
= 0.22V TO 3.5V
CM
CC
REF
5
OUT
1.25V
6
V
OUT
–
V
IN
+
V
IN
8
M81
9
M27
10
–
+
M9
1
P9
2
P27
3
P81
V
LT1996
V
EE
4
= 1.11V TO 2V
V
CM
> 45mV
V
DM
7
CC
REF
OUT
6
V
OUT
V
V
5
3V
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
V
LT1996
V
EE
4
V
= 4V TO 7.26V
CM
7
CC
REF
5
OUT
1.25V
6
V
OUT
–
V
IN
+
V
IN
8
M81
9
M27
10
–
IN
+
IN
M9
1
P9
2
P27
3
P81
V
CM
3V
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
V
7
V
CC
LT1996
REF
V
EE
4
3V
= –.78V TO 1.67V
<– 45mV
V
DM
3V
7
V
CC
LT1996
REF
V
EE
4
= –5V TO –1.74V
CM
5
5
OUT
OUT
1.25V
6
V
OUT
6
V
OUT
3V
8
M81
9
M27
10
–
V
IN
+
V
IN
M9
1
P9
2
P27
3
P81
LT1996
V
EE
4
7
V
CC
REF
5
OUT
1.25V
6
V
OUT
–
V
IN
+
V
IN
8
9
10
1
2
3
3V
M81
M27
M9
P9
P27
P81
3V
V
LT1996
V
EE
4
7
CC
REF
OUT
5
1.25V
6
V
OUT
3V
8
M81
9
M27
–
V
IN
+
V
IN
10
M9
1
P9
2
P27
3
P81
LT1996
V
EE
4
3V
V
CC
1.25V
V
= –1.28V TO 6.8V
CM
3V
8
M81
9
M27
10
–
V
IN
+
V
IN
M9
1
P9
2
P27
3
P81
LT1996
V
EE
4
7
V
CC
REF
5
OUT
1.25V
6
V
OUT
–
V
IN
+
V
IN
8
9
10
1
2
3
V
= 9.97V TO 18V
CM
M81
M27
M9
LT1996
P9
P27
V
EE
P81
V
= –17V TO –8.9V
CM
3V
7
V
CC
REF
4
OUT
5
1.25V
6
V
OUT
3V
8
M81
9
M27
10
–
V
IN
+
V
IN
M9
1
P9
2
P27
3
P81
LT1996
V
EE
4
3V
V
CC
1.25V
V
= –2V TO 8.46V
CM
V
= 12.9V TO 23.4V
CM
V
= –23V TO –12.5V
CM
Figure 15. Common Mode Ranges for Various LT1996 Difference Amp Configurations on VS = 3V, 0V, with Gain = 9
7
6
1.25V
V
OUT
REF
OUT
5
7
6
1.25V
V
OUT
1996 F15
REF
OUT
5
20
1996f
LT1996
U
WUU
APPLICATIO S I FOR ATIO
8
M81
9
M27
10
–
V
IN
+
V
IN
–
V
IN
+
V
IN
M9
1
P9
2
P27
3
P81
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
2.5V
5V 5V
7
V
LT1996
V
EE
CC
REF
OUT
5
2.5V
6
V
OUT
V
V
4
V
= –0.83V TO 3.9V
CM
5V
7
V
CC
LT1996
V
EE
4
V
= –3.7V TO 7.8V
CM
REF
OUT
5
2.5V
6
V
OUT
–
V
IN
+
V
IN
8
M81
9
M27
10
–
IN
+
IN
M9
1
P9
2
P27
3
P81
V
7
V
CC
LT1996
V
EE
4
= 1.1V TO 4.2V
CM
> 5mV
V
DM
REF
OUT
5
6
V
OUT
V
V
5V
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
V
7
V
CC
LT1996
V
EE
4
= 3.8V TO 15.3V
CM
REF
OUT
5
2.5V
–
V
6
V
OUT
IN
+
V
IN
8
M81
9
M27
10
–
IN
+
IN
M9
1
P9
2
P27
3
P81
V
CM
5V
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
V
CM
7
V
CC
LT1996
V
EE
OUT
REF
5
4
5V
= –0.56V TO 3.7V
<– 5mV
V
DM
5V
7
V
CC
LT1996
V
EE
REF
OUT
5
4
= –11.7V TO 0.3V
2.5V
6
V
OUT
6
V
OUT
5V
8
M81
9
M27
–
V
IN
+
V
IN
10
M9
1
P9
2
P27
3
P81
V
LT1996
V
EE
4
7
CC
REF
5
OUT
2.5V
6
V
OUT
–
V
IN
+
V
IN
8
9
10
1
2
3
5V
M81
M27
M9
P9
P27
P81
V
LT1996
V
EE
4
5V
7
CC
REF
5
OUT
2.5V
6
V
OUT
5V
8
M81
9
M27
–
V
IN
+
V
IN
10
M9
1
P9
2
P27
3
P81
V
LT1996
V
EE
4
5V
CC
2.5V
V
= –12.6V TO 15.6V
CM
5V
8
M81
9
M27
10
–
V
IN
+
V
IN
M9
1
P9
2
P27
3
P81
V
LT1996
V
EE
4
7
CC
REF
5
OUT
2.5V
6
V
OUT
–
V
IN
+
V
IN
8
9
10
1
2
3
V
= 9.8V TO 38.1V
CM
M81
M27
M9
LT1996
P9
P27
V
EE
P81
V
= –35.1V TO –6.8V
CM
5V
7
V
CC
REF
4
5
OUT
2.5V
6
V
OUT
5V
8
M81
9
M27
10
–
V
IN
+
V
IN
M9
1
P9
2
P27
3
P81
V
LT1996
V
EE
4
5V
CC
2.5V
V
= –17.1V TO 19.5V
CM
V
= 12.8V TO 49.5V
CM
V
= –47.2V TO –10.5V
CM
Figure 16. Common Mode Ranges for Various LT1996 Difference Amp Configurations on VS = 5V, 0V, with Gain = 9
7
6
2.5V
V
OUT
REF
OUT
5
7
6
2.5V
V
OUT
1996 F16
REF
OUT
5
1996f
21
LT1996
U
WUU
APPLICATIO S I FOR ATIO
8
M81
9
M27
10
–
V
IN
+
V
IN
–
V
IN
+
V
IN
M9
1
P9
2
P27
3
P81
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
2.5V
5V 5V
7
V
LT1996
V
EE
CC
REF
OUT
6
V
OUT
5
V
IN
V
IN
4
–5V
V
= –4.4V TO 4.2V
CM
5V
7
V
LT1996
V
EE
CC
REF
OUT
6
V
OUT
5
–
V
IN
+
V
IN
4
–5V
V
= –23.9V TO 8.1V
CM
8
M81
9
M27
10
–
+
M9
1
P9
2
P27
3
P81
V
LT1996
V
EE
4
7
CC
REF
OUT
6
V
OUT
V
V
5
5V
8
M81
9
M27
10
–
IN
+
IN
M9
1
P9
2
P27
3
P81
V
LT1996
V
EE
4
7
CC
REF
OUT
6
V
OUT
5
–5V
–5V –5V
5V
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
V
V
= –3.9V TO 4.8V
CM
<– 5mV
V
DM
5V
7
V
CC
LT1996
V
EE
4
–5V
= –31.4V TO 0.6V
CM
REF
5
OUT
6
V
OUT
V
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
V
CM
= –5V TO 3.7V
CM
> 5mV
V
DM
5V
7
V
CC
LT1996
V
EE
REF
OUT
5
4
–5V
= –16.4V TO 15.6V
5V
–
V
6
V
OUT
IN
+
V
IN
8
M81
9
M27
–
V
IN
+
V
IN
10
M9
1
P9
2
P27
3
P81
V
LT1996
V
EE
4
7
CC
REF
OUT
6
V
OUT
5
–
V
IN
+
V
IN
–5V
V
= –40.4V TO 38.4V
CM
8
9
10
1
2
3
–5V
5V
5V
8
M81
9
M27
10
–
V
IN
+
V
IN
M9
1
P9
2
P27
3
P81
V
LT1996
V
EE
4
7
CC
REF
OUT
6
V
OUT
V
V
5
–5V
V
= –52.4V TO 49.8V
CM
–
IN
+
IN
–5V –5V
8
9
10
1
2
3
M81
M27
M9
P9
P27
P81
M81
M27
M9
P9
P27
P81
V
CM
V
CM
5V
7
V
CC
LT1996
REF
V
EE
4
–5V
= 4.6V TO 60V
5V
7
V
CC
LT1996
REF
V
EE
4
= 7.6V TO 60V
OUT
5
OUT
5
6
V
OUT
6
V
OUT
5V
8
M81
9
M27
–
V
IN
+
V
IN
10
M9
1
P9
2
P27
3
P81
V
CM
= –60V TO –10.2V
5V
8
M81
9
M27
10
–
V
IN
+
V
IN
M9
1
P9
2
P27
3
P81
V
= –60V TO –10.2V
CM
Figure 17. Common Mode Ranges for Various LT1996 Difference Amp Configurations on VS = ±5V, with Gain = 9
V
LT1996
V
EE
4
–5V
V
LT1996
V
EE
4
–5V
5V
7
CC
REF
OUT
6
V
OUT
5
5V
7
CC
REF
OUT
6
V
OUT
5
1996 F17
22
1996f
PACKAGE DESCRIPTIO
LT1996
U
DD Package
10-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1699)
3.50 ±0.05
0.675 ±0.05
1.65 ±0.05
(2 SIDES)2.15 ±0.05
PACKAGE
OUTLINE
0.25 ± 0.05
2.38 ±0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
(2 SIDES)
0.50
BSC
PIN 1
TOP MARK
(SEE NOTE 6)
0.200 REF
R = 0.115
TYP
3.00 ±0.10
(4 SIDES)
0.75 ±0.05
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).
CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
1.65 ± 0.10
(2 SIDES)
0.00 – 0.05
2.38 ±0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
106
15
0.50 BSC
0.38 ± 0.10
0.25 ± 0.05
(DD10) DFN 1103
0.889 ± 0.127
(.035 ± .005)
5.23
(.206)
MIN
0.305 ± 0.038
(.0120 ± .0015)
TYP
RECOMMENDED SOLDER PAD LAYOUT
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
3.20 – 3.45
(.126 – .136)
0.50
(.0197)
BSC
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
GAUGE PLANE
0.18
(.007)
0.254
(.010)
DETAIL “A”
DETAIL “A”
0° – 6° TYP
0.53 ± 0.152
(.021 ± .006)
SEATING
PLANE
3.00 ± 0.102
(.118 ± .004)
(NOTE 3)
4.90 ± 0.152
(.193 ± .006)
(.043)
0.17 – 0.27
(.007 – .011)
TYP
1.10
MAX
12
0.50
(.0197)
BSC
0.497 ± 0.076
6
45
(.0196 ± .003)
REF
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
0.86
(.034)
REF
0.127 ± 0.076
(.005 ± .003)
MSOP (MS) 0603
1996f
8910
7
3
23
LT1996
U
TYPICAL APPLICATIO
Micropower AV = 90 Instrumentation Amplifier
V
98
+
V
M
1/2 LT6011
–
+
V
P
1/2 LT6011
–
1
450k/81
450k/27
450k/9
450k/9
450k/27
450k/81
23 4 5
7 610
450k
4pF
–
+
4pF
LT1996
450k
Bidirectional Controlled Current Source AC Coupled Amplifier Differential Input/Output G = 9 Amplifier
OUT
1996 TA02
+
V
M81
M27
M9
P9
P27
P81
LT1996
4
–
V
S
S
7
6
R1
5
10k
V
IN
–
V
)
IN
I
LOAD
+ –
9(V
IN
=
10kΩ
0.1µF
8
M81
9
M27
10
M9
1
P9
2
P27
3
P81
8
9
10
–
V
IN
1
+
V
IN
2
3
+
V
S
7
LT1996
4
–
V
S
GAIN = 117
BW = 4Hz TO 5kHz
8
M81
9
M27
10
–
V
6
5
IN
V
OUT
V
IN
USE V
OUTPUT COMMON MODE LEVEL
M9
1
+
P9
2
P27
3
P81
TO SET THE DESIRED
OCM
LT1996
4
–
V
S
+
V
S
7
6
5
10k
–
LT6010
10k
V
OUT
V
OCM
+
V
OUT
1996 TA03
RELATED PARTS
PART NUMBER DESCRIPTION COMMENTS
LT1990 High Voltage Difference Amplifier ±250V Input Common Mode, Micropower, Pin Selectable Gain = 1, 10
LT1991 Precision, 100µA Gain Selectable Amplifier Gain Resistors of 450k, 150k, 50k
LT1995 30MHz, 1000V/µs Gain Selectable Amplifier High Speed, Pin Selectable Gain = –7 to 8
LT6010/LT6011/LT6012 Single/Dual/Quad Precision Op Amp Similar Performance as LT1996 Diff Amp, 135µA, 14nV√Hz,
Rail-to-Rail Out
LT6013/LT6014 Single/Dual Precision Op Amp Lower Noise AV ≥ 5 Version of LT1991, 145µA, 8nV/√Hz,
Rail-to-Rail Out
LTC6910-X Programmable Gain Amplifiers 3 Gain Configurations, Rail-to-Rail Input and Output
+
–
24
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
1996f
LT/TP 0205 1K • PRINTED IN USA
© LINEAR TECHNOLOGY CORPORATION 2005
Loading...