Keithley 427 Service manual

INSTRUCTION MANUAL

MODEL 427

CURRENT AMPLIFIER

0 1975, KEITHLEY INSTRUMENTS, INC.

CLEVELAND, OHIO, U.S.A.

DOCUMENT NUMBER 29103

CONTENTS

CONTENTS

1. GF,Ng3,&.L DESCRIPT~ON-------------------------------.-------- 1

2.

OPERATION-------------------------------------------------- 4

MODEL 427

3. APPLICATIONS----------------------------------------------- 9

ACCESSORIES------------------------------------------------ 10

4.

5. ClRC"IT DESCRIPTION---------------------------------------- 12

6. REp~Cp‘&LE PARTS------------------------------------------ 17

7. CALIBRATION------------------------------------------------ 26

SCNEMATICS---------------------------------------------------- 33

1074

MODEL 427

ILLUSTRATIONS

ILLUSTRATIONS

Figure No. Title

1 Front panel. __-___---___-_____-___________________ 1
2 Front pane1 Controls. ________--______-____________ 3
3 Rear Panel Controls. - 3
4 shunt Method Measurement, -----------------__------ 4
5 Effect of Shunt capacitance. ---------------------- 4
6 Compensation for Shunt Capacitance. --------------- 5
6b Extended Frequency Response. ---------------------- 5
7 Fee&a& Method. _--__-_-__________-_______________ 6
8 Input Voltage N&se. -------------___-------------- 6
9 Bandwidrh of Feedback System. --------------------- 7
9b Effect of filter on Noise spectrum. -------------- 7

9, Effect of Input Capacitance on Noise. ------------- 7
10 Frequency Compensafian. ---------_____--_-__------- 7
11 Plot of Noise-Improvement Contours.
12 Block Diagram of a High-Speed Current Amplifier, -- 12

__- _____ -__-__

Page

9

13 Filter circuit. ------_______--_-__-_______________ 13
14 power s”pp~y Regulator. --------------------------- 13
15 current suppression. ------------__---------------- 13
lb COmpOnent Layout - PC-291. ------------------------ 13
17 component Layout - pc-290, -----------------___---- 14
18 component Layout ­19 COmpOnenf Layout ­20 chassis - Top vie”. ------------___--------------- 16
21 Chassis Assembly - Exploded “few. ----------------- 19
22 Bottom CO”er Assembly. -------------------------- 19
23 Measurement of Input Voltage Drop. ---------------- 27
24 Measurement of Rise Time. -------_--_____-___-_____ 28
25 Measurement of Filter Rise Time. ------------------ 29

pc-2?39.
P&292.

---------_------___-____ 15

---------------__------ 15

1074

iii

SPECIFICATIONS

SPECIFICATIONS

RANGE: 10’ to 10” volts/ampere in eight decade ranges.

(10-13 ampere resolution to 10.” ampere full output).
OUTPUT: *10 volts at up to 3 milliamperes.
OUTPUT RESISTANCE: Less than 10 ohms dc to 30 kHz.
OUTPUT ACCURACY: 12% of reading to the lo9 vattsl

ampere range, +4% of reading on the 1O’O and 10”

volts/ampere ranges exclusive of noise. drift and current

offset.

RISE TIME (10% to 90%): Adjustable in lx and 3.3x steps

from “Fast Rise Time” listed below to 330 rnsec.

NOISE VS. RISE TIME’:

I

FAST RISE TlML

MODEL 427

STABILITY: Current offset doubles per 10°C above 25°C.

Voltage drift is less than 0.005% per “C and less than

0.005% ,,er da” of full outDut after l.hour warmur,.

OFFSET CURRENT: Less than lQLz am&e at 25”C’and up
CURRENT SUPPRESSION: lo-lo ampere to 10-a ampere in

OVERLOAD INDICATION: Lamp indicates pre-filter or post.

CONNECTORS: Input: (Front) ENC. Output: (Front and

DIMENSIONS; WEIGHT: Style M 3%” half.rack, overall bench

,-

to 7001, relative humidity.

eight decade ranges with 0.1% resolution (lO.turn poten.

tiometer). Stability is +O.Zo/. of suppressed value per “C
bO.Z% per day.

INPUT VOLTAGE DROP: Less than 400 /.IV for fullaxle

output on the 10” to 10” volts/ampere ranges when

properly zeroed.

EFFECTIVE INPUT RESISTANCE: Less than 15 ohms on the

10” and 105 volts/ampere ranges. increasing to less than

4 megohms on the 10” volts/ampere range.

MAXIMUM INPUT OVERLOAD: Transient: 1000 volts on any

range for up to 3 seconds using a Keithley (or other 10
mA-limited) highaoltags supply. Continuous: 500 volts
on the 10” to 1O’voltsjampere ranges, decreasing to ‘ZOO
an the 10”. 70 on the lo5 and 20 volts on the 10’ volts,
ampere ranges.

filter overload.

DYNAMIC RESERVE: 10 (20 dB).

Rear) BNC.

POWER: 90.125 or 180.250volts (switch selected), 50.60 Hr.

5 watts.
size4’~highx8%‘widex12L/a”deep(100x217 x310 mm).

Net weight, 7 Ibs. (3.0 kg).

i”

1074

MODEL 427

GENERAL DESCRIPTION

SECTION 1.

l-1. GENERAL; The Model 427 Current Amplifier is a
high-speed, feedback-type amplifier with particular
features useful for automated semiconductor testing,

mass spectrometry, and gas chromatography applications.

l-2. FEATURES.

a. Wide Dynamic Range. Selectable rise times permit
low-noise operation important when resolving small
current levels.

b. High Speed.
out of a 10sSZpere signal with a 100 microsecond
rise time.

Typical resolution is 20 picoamperes

GENERAL DESCRIPTION

C. Built-in Current Suppression.
the signal level can be measured since large ambient
current levels can be easily suppressed.

d. Overload Indication.
assured since overloads are automatically indicated.

e. Variable Cain. The GAIN Switch is designated
in eight gain positions from lo4 to 1011 volts per
ampere - therefore gain adjustment is straight forward.

f. Variable Kise Time. Optimum response can be
selected for each gain setting since a separate RISE

TIME switch is provided on the front panel.

Small changes in

Accurate measurements are

0471

1

GENERAL DESCRIPTION

MODEL 427

TABLE 1-l.

Front Panel Controls and Terminals

PUSH Power Switch (S302)
GAIN Switch (5201)
RISE TIME (5101)
SUPPRESSION

MAX AMPERES Switch (S303)

FINE Control (R333)

POLARITY Switch (5304)

ZERO ADJ Control (R235)

INPUT

Receptacle

OUTPUT Receptacle (5102)

~ OVERLOAD Indicator (DS302)

(5202)

Functional Description

Controls power to instrument.
sets gain in Volts per ampere.

Sets

rise time Fn milliseconds.

Sets maximum suppression.

Adjusts suppression.

Sets polarity of suppression.

Adjusts output zero.

Input source connection.

Output connectFon.

Indicates overload condition.

Rear Panel Controls and Terminals

TABLE 1-2.

Paragraph

2-4, al
2-4, a2

2-4, a3

2-4, a4
2-4, a5
2-4, a6
2-4, a7
2-3, a
2-3, b

2-5, d

Control or Terminal Functional Description Paragraph

Line Switch (5301)

Power Receptacle

Fuse (F301)

0lJTPuT Receptacle

(P305)

(5103) Output connection.

Sets instrument for 117V or 23411.
Connection to line power.
Type 3AG Slow-Blow, 117V @ l/4 A (w-17)

234V @ l/S A (w-20)

WARNING

Using a Line Power Cord other than the one supplied

with your instrument may result in an electrical

shock hazard. If the Line Power cord is lost or
damaged,

replace only with Keithley Part No. CO-7.

2-4, b
2-3, c

2-3, b

0878

MODEL 427

GAIN

SWITC

S20

,----SUPPRESSION-,

FINE

ADJUST

GENERAL DESCRIPTION

POLARITY

SWITCH

INPUT

5202

ZERO ADJ

R235

FIG"RF 2.

OVERIDAD Power

.

DS302 S302

want Panel Controls.

P

OUTFW

5102

I

0471

FIGURE 3.

Rear Panel Controls.

OPERATION

MODEL 427

SECTION 2.

MEASUREMENT CONSIDERATIONS.

2-1.

a. Current-Detection Devices.

‘small electrical c~rrent8 has been the basis for a

number of instrumental methods used by the analyst.
Ion chambers, high-impedance electrodes, many forms
of ch=“metog=aphic detectors, phototubes and multipli­ers are coaronly-used t=ansduce=a which eequire the
measurement of small currents. Devices used for this
measurement a=e often called electraaeters.

b.. In any measure-

ment, if e”“=ce noise greatly exceeds that added by

the inst=“mentatian, optimization of instrumenteti””
is unimportant.
ical minimum, optimization of instrumentation charac-

teristics becomes imperative. TO determine the cate­gory into which this meas”=ement falls. the researcher
needs t” be familiar with the characteristics which

impoee theoretical and practical limitations on his
me*surement .

theoretical limitations present in voltage measueements

The noise inceeases with 8”“=ce realstance, and the

familiar equation for the mean-square noise voltage is

q = 4kTRAf Eq. 1

When source noise approaches theoret-

Most researchers a=e familiar with the

The DleasUrement of

OPERATION

Prom this equation it is irmnediately apparent that the
m%QB”rement of emall current =equi=es large values of
R, i.e., high impedance levels. Howwee, thfe gives
difficulties for meas”=ements requiring wide bandwidths
because of the RC time constant associated with a
high-megohm resistor and even a few picofarads of cir-

cuit capacitance.
generating a voltage across a parallel RC. The fre­quency response of this current measurement is limited

by the RC time constant.

end the -3 dB p”int “CC”=B at a frequency

Lor* noise and high

requirements.
techniques must be used which obtain high speed “sing

high-impedance devices.

C. Hiah Speed Methods.

1. High epeed can, af c”“=se, be obtained in .

shunt-type meae”=eme”t by “sing a low value for the
shunt resist”=.
resistor value int=ad”ces excessive noise into the

meQ8”rement.

Figure 4

speed,

TO optimize a current-measuring system,

As pointed ““t above, such a srmll

shows a c”==ent s”“=ce

Figure 5

therefore, a=e contradictory

shows this response

whe=e k is the Boltzma”” Constant, T ia the absolute

temperature of the s”“=ce resistance R, and
noise bendwidth( 3

single RC rolloff.) In the case of cureent measure-

ments it Is more appropriate to consider the noise

current generated by the ~l”“=ce and load resistances.
The mean squaee noise c”==ent generated by a resistor

is given by Eq. 2.

FIGURE 4.

In the shunt method c”=re”t is measured by

the voltage drop ac=“ee a resistor.

times the 3 dB bandwidth for a

Af

is the

2. A second method to achieve bandwidth is to
keep R large, to accept the frequency roll-off
starting at F”, and t” change the frequency eesponae
of the voltage amplifier a8 ehown in Figure 6a. The
combined effects of the RC time c”“sta”c folloved
by this amplifier is shown in Pigure 6b and it is
seen that the frequency response of the c”==ent

measurement has been extended to Pl. The frequency

at which the amplifier gain sta=‘ts to increase must
be exactly equal t” the frequency F” determined by
the RC time c”nstant in order for this approach t”
result in a flat frequency respanse. Therefore,

FO

FIGURE 5. The frequency respanse of the shunt method

1s limited by omnipresent ahunt capacitance.

LOG FREQUENCY

I

0471

MODEL 427

OPERATION

this oethad is ueeful only far application* where
the shunt capacitance C is constant. Aaide from
thin drawback this is * 1eSitimste approach which
is being wed in low-noise, high-speed current-

meesuring applicatians. In addition to current noise

in the *hunt and in the amplifier input stage, B

maJor source of noise in this system *ri*** from

the voltage-noise generator ssaociated with thb in­put atage (reflected a* current noise in the shunt
resistor) caused by the high-frequency peeking in
the following stages of amplification. More will be
said about this in the discussion on noise behavior.

3. A third method used for speeding up * current

measurement asas guarding techniques to eliminate

the effects of capacit*nces. Unfortunately only

certain type* of capacitance*, such ** cable cap=-

itances, can be conveniently eliminated in this

manner.

itences associated with the *ource itself became*

very cumbersome and m*y not be feasible in many in-

stsnces.

*re identical to those mentioned in the second

system.

4. A fourth circuit configuration combines the
capability of low-noise and high-speed performance
with tolerance for varying input C and eliminate*
need for separate guard by making the ground plane
*n effective guard. This is the current-feedback
technique.
ment of 3 over shunt technique*. Again, the major
sources of noise are identical to those mentioned

in the second system.

d. Noise in Current Measurements. Noise forms *

b*aic limitation in *nv hinh-speed current-measurinn

system. The shunt *y&m give; the simplest curren;

measurement but does not give low-noise performance.

A properly designed feedback *y*tem gives superior

noise - bandwidth performance. Noise in these two

systems will be discussed next.

1. Noise Behavior of the Shunt System. High
speed end low noise *r* contradictory requirements
in any current meesurement because *orw capacitance
is always present. The theoretical performance
limftetion of the shunt *yetem c*n be calculated **

To eli,r,inate the effect of parasitic c*p*c-

The major *ourc** of noise in this *“*tern

This technique gives * typical improve-

The rms thermal noise current (in) generated by *
resistance R is given by

Eq. 4

The equivalent noise bandwidth (.f) of * parallel SC
combination is Af = 1/(4RC) snd the eignal hand­width (3 dB bandwidth) F, = 1/(2nRC). For practical

purposes peak-to-peak noise is taken 88 5 times the
ml* value. The peak-to-peak noise current can now

be written a*

i

UPP =

In practice, e typical value for shunt cape.cit*nce

is 100 picofarads.
rule-of-thumb is obtained. The lowest ratio of

detectable current divided by signal bandwidth using

*hunt-techniques is 2-10-14 ampere/Hertz for B peak­to-peak signal-to-noise ratio equal to 1. A coroll-

ary far this rule-of-thumb expresses the noise cur-

rent in term* of obtainable risetime (lo-SO% rise­time tr = 2.2 RC). The lowest product of detectable
current and risetime using shunt technique* is 7 x

lo-l5 ampere seconds.
assumed that the voltage amplifier does not contri­bute noise to the measurement.

2. Noise Behavior of the Feedback System. There

are three *ource* of noise in the feedback system

that have to be looked at closely. The firat two,

input-stage shot noise and current noise from the
mea*urinS resistor, are rather straight-forward. The

third, voltage noise from the input device of the
amplifier, cau*e* *ome peculiar difficulties in the
measurement. Any resistor connected to the input

injects white current noise (Eq. 4). In the circuit
of Figure 7 the only resistor that is connected to

the input is the feedback resistor R. As in the

shunt system, R must be made large for lowest noi*e.
Beceuse this noise is white, the total contribution
can be calculated by equ.,ting Af to the equivalent
noise bandwidth of the system. The second *ource of
noise is the current noise from the amplifier input.
This component is essentially the shot noise asaoci­ared with the gate leakage current (io) of the input
device. Its rms value equals . . .

2 x 10-9 F,

With this value the following

F

In this derivation it has been

Eq. 5

FIGURE 6.

0471

LOG FREWENCY

FO

ny tailoring the frequency response of the
amplifier (Pig. 6a) the frequency response
of the shunt method c*n be extended.

F>

FIGURE 6b.

FO

Extended frequency response.

LOG FREQUENCY

F,

OPERATION

MODEL 427

T;; = J-zTp-

where e is the electronic charge. The contribution

of this noise generator is also white. N*t only do

these two noise sources generate white current noise,
the noise in a given bandwidth is also independent

of the input capecitence C. The mejor source of

noise in e feedback current meesurement is the noise

contribution aseocisted with the voltage noise of

the input amplifier. The voltage noise ten be rep-

resented by a VOltage noise generator (0,) et the
emplifier input es shown in Figure 8. This wise
generator itself is assumed to be white. However,
its total noise contribution to the current-measuring

system is not white.

reveal that et low frequencies P large em*u*t of feed­beck ie applied around the voltage noise source {en).

However, the SC combination ettenuetes the high-

frequency components of V,,t so that no feedback is
present et high frequencies. Thus, the noise con-

tribution to the output voltage V,,t from the valt­age noise source a* is no longer independent of

frequency. The noise is “colored” and increases in
intensity for ell frequencies higher than F,. The
resulting noise spectrum is shown in Figure 9b. The
tote1 system noise is related to the are* under
this curve.
plotted on the horizontal axis, the eree under the

curve et higher frequencies represents e signifi­cantly larger amount of noise then e similar eree

*t low frequencies.
the frequency response of the current measuring
system.

interesting et this point to compare this noise
spectrum with the frequency response of the voltage
amplifier in Figure 4 es shown in Figure 6a. A volt­age noise eouec.e et the input of the amplifier would
generate a noise spectrum according to the amplifier

frequency response as shown in Figure 6a. The noise

spectrum of such e system, then, is identical to the

noise spectrum of the feedback system as given in

Figure 9h.

that signal-to-noise performance of a measurement
cen**t be improved by feedback techniques. At the

high-frequency end the voltege noise is limited by

the frequency FA which is the high-frequency roll-

off point of the operational amplifier. It should

Because the logarithm of frequency is

Figure 9e ia identical to Figure 6b. It is

This illustrates the well-known fact

Inspection of Figure 6 will

For comparison, Figure 9a show

he noted that even though the useful bandwidth of

the system extends only t* Fl, there era noise com-

ponents of higher frequency present. To obtain
best widebend-noise performance, these high-frequency
noise components have to be removed. This ca* be

achieved by adding a low-pass filter section follow-

ing the feedback input stage. If the band-pass of

thin low-pass filter is made adjuetable this filter
can nerve the dual purpose of removing high-frequency

noise end of limiting the signal bandwidth of the

system.

2-2. THEORY OF OPERATION.

8. Current Feedback Technique. The basic circuit
configuration used in the current-feedback technique
is shown in Figure 7.
current-measuring resistor R is placed in the feedback
loop of e* inverting emplifier with a gain of A*. The
frequency response obtained with this circuit is iden­tical to thet s+nvn in Figure 6b. F* agein is the
frequency associated with the RC time constant:

F, =

The frequency response of the syetem is extended t* a

frequency fl where

F

, = AoF,

Note that the frequency rerponse is automatically flat
without heving to match break points. However, the
total bandwidth of the system (Fl) is still limited
by the value of the ahunt capacitance C across the

input.
back technique avoids the use of low values for R
which could generate exceesive current noise.

difficulty of the feedback system ariees from shunt
capacitence esaociated with the high-megohm resiaeor R

in the feedheck path.

the resistor is CFr then the bandwidth (FF) of the

system is determined by the time COnstent RCF:

This improved frequency response of the feed-

b. Refinements of the Feedback System. A major

In this configuration the

SE Eq. 6

Eq. 7

.

If the shunt cepacitence acroes

FIGURE 7.

6

Beslc circuit configuration for the feed­back method.

FIGURE 8.

The voltage noise associeted with the am­plifier input device is en important eourc~

of noise in the high-speed feedback syatew

0471

MODEL 427 OPERATION

FIGURE 9.

FIGURE 9b.

The bendwidth of the high-speed feedback

system (Fig. 9a) ten he limited by using
e filter with either e -6 dB/actave or a

-12 dB/octave roll-off. The effect of the
filter on the noise spectrum is showwin
Fig. 9h.

Effect of input capacitance on

noise is shown in Fig..9c.

Effect of filter on noise spectrum.

FIGURE 10.

Frequency compensation.

FF = 1

2 nllcp

Eq.

A slight modification of the feedback loop can correct

this problem es shown in Figure 10. If the time con­stant RlCl is made equal to the time constent R.CF,
it CB* be shown that the circuit within the dotted
line behaves exactly es a resistance R. The matching

of time constants in this cese does not become e draw­beck because the copscitances involved era all constant
and not effected by input impedance.

C. -12 dB/actave Filter.

1. Theory. To obtain optimum widehand noise per-

fomence e filter with e single high-frequency roll­off (i.e.,
dB/octeve is required.

-6 dB/octave) is not sufficient end -12
The effect of e -6 dB filter

is shown in Figure 9a end h. The filter is used to
limit the system bandwidth to a frequency F2, smaller
then Fl. The effect af this filter on the noise
spectrum is shown in Figure 9b. It ten be seen that

there ace egain high-frequency noise components above
F2, the useable bandwidth of the system. These can
he eliminated by using e filter with e -12 dB/octave

roll-off. The result of such .a filter on noise per-

formance is also shown in Figure 9b.

FIGURE 9c.

0471

Effect of input capacitance on noise.

2. Model 427. The input smplifier 18 followed by

en adjustable low-pass filter having e -12 dB/octeve

roll-off end a valtage gain of 10X. The voltage

gain in the low-pass filter avoids premature over-

loading in the input amplifier which ten be seen es
fallows. The maximum output voltage V,,t is $10

volts.

The maximum signal level et the input of the

low-pass filter is, therefore, +l volt. At this
point in the circuit, wide-band noise could still be
present end exceed the l-Volt signal level. The
voltege gain of 10 in the filter allows the total
pre-filter wide-hand noise to exceed the full scale

signal by e factor of 10 (20 dB). The frequency re-

sponse of this filter is edjustahle for variable

“damping” control.

7

OPERATION

2-3.

CONNECTIONS,

MODEL 427

2-5. OPERATING CONSIDERATIONS.

8.

Input.
type which metes with coaxial cables such es Keithley
Models 8201 end 8202.
high.

(5102 on the front, 5103 on the reer panel). These
era BNC type* where the inner contact is output high
end the outer shell is chassis ground.

rear panel is a 3-prong connector which metes with
Keithley pare number CO-6 line cord.

2-4. CONTROLS.

The outer shell is low or chassis ground.

b. Output.

C.

Power Input. The power receptacle (P305) on the

a. Front Panel.

1. Power Switch “PUSH ON” (5302). This switch

controls the line power to the instrument. ‘The

switch is a special pushbutton type with “Power On”
indicated by a self-contsined pilot lamp.

2. GAIN (VOLTS PER AMPERE) (S201). This switch
sets the overall gain in eight positions from 104
to loll.

ment of zero offsets.

3. RISE TIME Switch (5101). This switch sets

the lo-90% rise time in 10 positions from .Ol to
300 milliseconds(for the filter section only).

4. SUPPRESSION (MAX) Switch (S303). This switch

sets the maximum current suppression in eight pasi­tions from lo-10 to 10-3 A. When the switch
to “OFF” the current suppression circuit is disabled.

5. SUPPRESSION (FINE) control (~333). This con-

trol permits adjustment of suppression with 0.1%
resolution.

6. SUPPRESSION (POLARITY) Switch (S304). This

switch set* the polarity of the current suppression

(referred to the input).

7. ZERO ADJUST Control (R235). This control per-

mits adjustment of zero offset through the u*e of the
OVERLOAD indicator.

b. Rear Panel.

1. Line Voltage Switch (S301). Sets instrument

for either 117 or 234 V operation.

The input receptacle (5202) is e SNC

The inner contact is circuit

Two outp,ut receptacles *re provided

A “ZERO CHECK” position permits adjust-

Fuse RequirP.ment* 3AG, Sla-Slo

117V: 1/4A
234V: l/SA

Keithley No. F”17
Keithley No. F”-20

is

set

8. Gain. The gein of the Model 427 is defined in
terms of volts per *“pare. Since the output level is
10 volts for e full scale input, the gain could also

be expressed e* sensitivity in emperes referred to the

input BS in Table 2-1. Vout = - (Iin x GAIN)

Gain or Sensitivity Referred to the Input

GAIN

Setting Resistor

b. Rise Time. The rise time for each gain setting

is listed in the specifications a* “FAST RISE TIME”.

These rise times are obtained when the RISE TIME

switch is set to the positions indicated in Table 2-2.

GAIN

Setti”g

104
105
106
107

1 10;

1oy
1010
loll

c. Suppressian. Current suppression is provided in

the Model 427 for suppression of input currents up to

10e3 amperes.

variations in e larger signal can be observed. Currents

of either polarity can be suppressed. To suppress an
input current the SUPPRESSION should be *et to supply

e current of apposite polarity. The FINE control permits

adjustment up to 1.5 times the MAX setting.

d. Overloads,
en overload et two places in the circuit: before end
after the “RISE TIME” filter circuit. The OVERmAD

lamp (DS302) will indicate whenever the voltsge sensed
ia greater then full scale regardless of the RISE TIME

setting or the frequency.

a. Zero Adjust. The ZERO CHECK po*ition grounds the
the input of the instrument and co”“ert* the cuerent
amplifier to a high-gain voltage amplifier. The ampli-

fied offset voltage will turn on the OVERLOAD indicator
whenever the input voltage offset exceeds 5100 t0l.
Therefore the ZERO control should be adjusted so that

the OVERLOAD indicator is off when in ZERO CHECK mode,

yielding the specified input voltage drop.

Feedback

Switch Settings for “FAST RISE TIME”

Rise
Time

15 ps
15 us

15 I*8
40 us

60 ps .03 ms 800
220
400

1.5 1 ms 100

TABLE 2-l.

Full Scale
sensitivity Output

(Amperes)

103
104
105
106
107
108
109
1010

&Is

“S ps

By suppressing background currents, smell

The overload sensing circuit detects

1

x 10-3
1 x 10-4
1 x 10-5
I x 10-6
1 x 10‘7
1 x 10-g
1

x 10-9
1 x 10-10

TABLE 2-2.

RISE TIME
Setti”g*

.Ol “S
.Ol ms
.Ol “S
.03 “9

.l “S 400
.3 ms 200

Eq. 9

Full Scale

(Volts)

10
10
10
10
10
10
10
10

DYlWl”iC
Range

2000
2000

2000
2000

J

8

MODEL 427

APPLIChTIONS

SECTION 3. APPLICATIONS

3-1. CURRENT~MEASURING SYSTEM. The typical current
meaeuring system consists of a current source, a cur­rent amplifier, and a monitoring device, The current
source could include an ion chamber, photomultiplier,
or other high-impedance device,
such as the Model 427 provides sufficient gain to drive
a monitoring device such as a chart recorder or other
readout. The Model 427 in this case provides an out­put voltage which is calibrated in volts per ampere
a.3 in equation 10.

- (V,,t I GAIN)

Example :

3-2.
trates the trade-off between fast rise time and dynamic
range.
a.s the ratio of maximum peak-to-peak current to peak­to-peak current noise.
taken as 5-times the rms current noise. The maximum
peak-to-peak current is 2-times the maximum full scale
current.

When using current suppression the current-suppres-

sion resistor should be considered as an additional
current-noise generator. The values given in Table
3-l do not include the contribution of the suppres­sion resistor. Therefore the selected suppression
resistor Rs, should be as large as possible to min-

imize the contribution to current noise.

Iin =

GAIN = 106 voltslampe~e
” O”t = +500 In”
The input current Iin would be:
lin = - (5x10-~vo1ts/106vo1ts per ampere)
Iin = - 5 x 10-7

NOISE BANDWIDTH CONSIDERATIONS.

For this application dynamic range is defined

Peak-to-peak current noise is

NOTE

The current amplifier

Eq. 10

amperes

Table 3-l illus-

3-3. NOISE-IMPROVEMENT CONTOURS. The sensitivity
and speed of the Model 427 (for either d-c or a-c

measurements) can be compared to the best perfarm-

ante obtainable with the shunt method of measuring
current. The best “noise-risetime” product that
can be achieved for d-c measurements (with 100 pF
shunt capacitance) in a shunt system is 7 x lo-15
ampere-seconds.
2 x lo-l5 ampere-seconds (also with 100 pF shunt
capacitance).

(lock-in, etc.) the-degree of improvement is a func-

tion of shunt capacitance and operating frequency.

The achieveable imprownene over the shunt method

can be plotted in a graph similar to a set of noise

contours. Figure 11 shows the measured impravemene

(negative dB) that can be obtained with the Model

427 at a given frequency and shunt capacitance when

compared to an ideal (noiseless) amplifier Fn a shunt

system.

,

I

FIGURE 11. Plot of noise-improvement contours.

However the feedback system achieves

When used in a-c narrowband systems

FREOUENCY IHI1

RMS’Noise Current (Typical)1 as a Function of Gain and Rise-Time Setting

TABLE 3-l.

,vL.. I,.,“,.,

PULL SCALE

GAIN
VIA

104
105
106
107

108

109
1010
1011

1

With up to 100 pP input shunt capacitance.

KEY :

x = Filter Bandwidth is greater than current-amplifier bandwidth.
* = larger Rise Times are useful for increased filtering of the signal arid noise inherent in the source.

0471

CURRENT

AMPERES 300 100 30

10-3
10-4
10-5
m-6

10-7

104’

10-9

lo-1o *

They do not further improve the instrument noise contribution except when the input shunt capacitance
exceeds 100 DF.

* * * *
*

* * *
:

*

2x10-15 4x10-15 1x10-14 4x10-14 1x10-13 4x10-13 x

*
* *

*

1x1:-14

*

*

*x1:-14

Noise increases aa input shunt capacitance increases.

RISE TIME SETTING

10 3 1 .3

*
* * *
* *
* *
*

2do-13 2d0-13 5do-13 2~0-12 5x10-12 x x
5~0-14 2do-13 ~0-13 2do-12 x

MO-12 2x10-l2 5x10-12 1x10-11

lx&

*

1x10-8
1x10-9 1.2x10-9 4x10-9
1x10-10

1.5x10-11

.l .03 .Ol

1.2x10-8 4x10-8 1x10-7

1.2x10-10 4x10-10 1x10-9

2x10-11 1x10-10 x

4x10-11

x

x x
x x

1x10-B

x

9

ACCESSORIES

MODEL 427

SECTION 4.

4-l. GENERAL.
be used with the Model 427 eo provide additional con­venience and versatility.

Description:
The Model 1007 is a dual rack mounting kit with over-

all dimensions 3-l/2 in. (64 mm) high and 19 in. (483
mm) wide.
of two Angle Brackets, one Mounting Clamp, and extra
mounting BCreWS.

The following Keithley accesaaries can

Model 1007 Rack Mounring hit

The hardware included in this kit consists

ACCESSORIES

OPERATING INSTRUCTIONS. A separate Instruction

4-2.

Manual is supplied with each accessory giving complete

operating information.

Application:
The Model 1007 co""erts any half-rack, style "M"

instrument from bench mounting to rack mounting in
a standard 19-inch rack.
for rack mounting 19-inch full rack width insfru-

ltE"tS.
The Model 1007 Rack Mounting Kit can be used to m"u"t

instruments of 11 inch or 14 inch depth.
should decide the position af the i"~tr"me"ts to be

rack mounted.

instruments positioned as shown and identified as
instrument “A” and "B".

The Assembly Inaeructions refer co

The kit may also be used

The user

10

Parts List:
Item

NO.

DWCl+ltiO”
22 Angle Bracket
23 Screw, 16-32 x

Phillips Pa" Hd
24 Mounting Clamp
25 Screw, %6-32 x

Phillips Pa" Hd
26 Kep Nut 116-32

27 Screw, 116-32 X

Phillips Pa" Hd

28 Screw, 116-32 x

Phillips Pa" Hd

5/8,

1,

l/2,

718,

VY

Keq'd

2
6

1 247988
1

3

2

1

Keithley
Part No.

27410B

-_

__

_-

__

__

0877

MODEL 427

Assembly Instructions:

ACCESSORIES

Model 1007 Dual Rack MountinS Kit

1. Before assembling the rack kit, determine the

pasition of each instrwnent. Since the inserumenfs
can be mounted in either location, their position
should be determined by the user’s meas”rement. The

following instructions refer to instruments “n” and

UB”

positio”ed as shorvn. For mounfinS 19-inch full

rack Width instruments, disregard steps 2 through 5.

2. Once the position of each instrument has been
determined, ebe “side dress” panels on both sides of
each instrument should be removed. Renxwal is
accomplished by looseninS the screwy (Item 8, oriS­inal hardware) in two places. Slide the “side dress”
panels co the rear of the instrument to remove.

3. The mountinS clamp is installed on instrument
“A” using the oriSina1 hardware (Item 8). With the
screws removed, insert the “mounting clamp” behind
the “corner bracket” (Item 7) and replace the screws
to hold the mounting clamp in place.

4. Tighten the screwy (Item 8) on instrument “B”.

Insert the “mounting clamp” behind the “corner

bracket” (Item 7) an instrument “8” a8 shown.

5. When mounting instruments having the same depth,
a screw (Item 25) and kep nut (Item 26) are required
to secure the two instruments together. When ,,,oune­ing instruments of different depth, da not use kep

nut (Item 26) but substitute shorter screw (Item 28).

6. Attach an “anSle bracket” (Item 22) on each
instrument using hardware (Item 23) in place of the
original hardware (Item 8). For 14 in. long instru-

ments use 116-32 x 518 Phillips screw (Item 23) with

116-32 kep “uf (Item 26).

7. The bottom cover feet and tilt bail assemblies

may be removed if necessary.

8. The original hardware, side dress panels, feet
and tilt bail assemblies should be retained for fut-

ure conversion back to bench mountine.

0777

11