INSTRUCTION MANUAL
MODEL 427
CURRENT AMPLIFIER
0 1975, KEITHLEY INSTRUMENTS, INC.
CLEVELAND, OHIO, U.S.A.
DOCUMENT NUMBER 29103
CONTENTS
CONTENTS
1. GF,Ng3,&.L DESCRIPT~ON-------------------------------.-------- 1
2.
OPERATION-------------------------------------------------- 4
MODEL 427
3. APPLICATIONS----------------------------------------------- 9
ACCESSORIES------------------------------------------------ 10
4.
5. ClRC"IT DESCRIPTION---------------------------------------- 12
6. REp~Cp‘&LE PARTS------------------------------------------ 17
7. CALIBRATION------------------------------------------------ 26
SCNEMATICS---------------------------------------------------- 33
1074
MODEL 427
ILLUSTRATIONS
ILLUSTRATIONS
Figure No. Title
1 Front panel. __-___---___-_____-___________________ 1
2 Front pane1 Controls. ________--______-____________ 3
3 Rear Panel Controls. - 3
4 shunt Method Measurement, -----------------__------ 4
5 Effect of Shunt capacitance. ---------------------- 4
6 Compensation for Shunt Capacitance. --------------- 5
6b Extended Frequency Response. ---------------------- 5
7 Fee&a& Method. _--__-_-__________-_______________ 6
8 Input Voltage N&se. -------------___-------------- 6
9 Bandwidrh of Feedback System. --------------------- 7
9b Effect of filter on Noise spectrum. -------------- 7
9, Effect of Input Capacitance on Noise. ------------- 7
10 Frequency Compensafian. ---------_____--_-__------- 7
11 Plot of Noise-Improvement Contours.
12 Block Diagram of a High-Speed Current Amplifier, -- 12
__- _____ -__-__
Page
9
13 Filter circuit. ------_______--_-__-_______________ 13
14 power s”pp~y Regulator. --------------------------- 13
15 current suppression. ------------__---------------- 13
lb COmpOnent Layout - PC-291. ------------------------ 13
17 component Layout - pc-290, -----------------___---- 14
18 component Layout 19 COmpOnenf Layout 20 chassis - Top vie”. ------------___--------------- 16
21 Chassis Assembly - Exploded “few. ----------------- 19
22 Bottom CO”er Assembly. -------------------------- 19
23 Measurement of Input Voltage Drop. ---------------- 27
24 Measurement of Rise Time. -------_--_____-___-_____ 28
25 Measurement of Filter Rise Time. ------------------ 29
pc-2?39.
P&292.
---------_------___-____ 15
---------------__------ 15
1074
iii
SPECIFICATIONS
SPECIFICATIONS
RANGE: 10’ to 10” volts/ampere in eight decade ranges.
(10-13 ampere resolution to 10.” ampere full output).
OUTPUT: *10 volts at up to 3 milliamperes.
OUTPUT RESISTANCE: Less than 10 ohms dc to 30 kHz.
OUTPUT ACCURACY: 12% of reading to the lo9 vattsl
ampere range, +4% of reading on the 1O’O and 10”
volts/ampere ranges exclusive of noise. drift and current
offset.
RISE TIME (10% to 90%): Adjustable in lx and 3.3x steps
from “Fast Rise Time” listed below to 330 rnsec.
NOISE VS. RISE TIME’:
I
FAST RISE TlML
MODEL 427
STABILITY: Current offset doubles per 10°C above 25°C.
Voltage drift is less than 0.005% per “C and less than
0.005% ,,er da” of full outDut after l.hour warmur,.
OFFSET CURRENT: Less than lQLz am&e at 25”C’and up
CURRENT SUPPRESSION: lo-lo ampere to 10-a ampere in
OVERLOAD INDICATION: Lamp indicates pre-filter or post.
CONNECTORS: Input: (Front) ENC. Output: (Front and
DIMENSIONS; WEIGHT: Style M 3%” half.rack, overall bench
,-
to 7001, relative humidity.
eight decade ranges with 0.1% resolution (lO.turn poten.
tiometer). Stability is +O.Zo/. of suppressed value per “C
bO.Z% per day.
INPUT VOLTAGE DROP: Less than 400 /.IV for fullaxle
output on the 10” to 10” volts/ampere ranges when
properly zeroed.
EFFECTIVE INPUT RESISTANCE: Less than 15 ohms on the
10” and 105 volts/ampere ranges. increasing to less than
4 megohms on the 10” volts/ampere range.
MAXIMUM INPUT OVERLOAD: Transient: 1000 volts on any
range for up to 3 seconds using a Keithley (or other 10
mA-limited) highaoltags supply. Continuous: 500 volts
on the 10” to 1O’voltsjampere ranges, decreasing to ‘ZOO
an the 10”. 70 on the lo5 and 20 volts on the 10’ volts,
ampere ranges.
filter overload.
DYNAMIC RESERVE: 10 (20 dB).
Rear) BNC.
POWER: 90.125 or 180.250volts (switch selected), 50.60 Hr.
5 watts.
size4’~highx8%‘widex12L/a”deep(100x217 x310 mm).
Net weight, 7 Ibs. (3.0 kg).
i”
1074
MODEL 427
GENERAL DESCRIPTION
SECTION 1.
l-1. GENERAL; The Model 427 Current Amplifier is a
high-speed, feedback-type amplifier with particular
features useful for automated semiconductor testing,
mass spectrometry, and gas chromatography applications.
l-2. FEATURES.
a. Wide Dynamic Range. Selectable rise times permit
low-noise operation important when resolving small
current levels.
b. High Speed.
out of a 10sSZpere signal with a 100 microsecond
rise time.
Typical resolution is 20 picoamperes
GENERAL DESCRIPTION
C. Built-in Current Suppression.
the signal level can be measured since large ambient
current levels can be easily suppressed.
d. Overload Indication.
assured since overloads are automatically indicated.
e. Variable Cain. The GAIN Switch is designated
in eight gain positions from lo4 to 1011 volts per
ampere - therefore gain adjustment is straight forward.
f. Variable Kise Time. Optimum response can be
selected for each gain setting since a separate RISE
TIME switch is provided on the front panel.
Small changes in
Accurate measurements are
0471
1
GENERAL DESCRIPTION
MODEL 427
TABLE 1-l.
Front Panel Controls and Terminals
PUSH Power Switch (S302)
GAIN Switch (5201)
RISE TIME (5101)
SUPPRESSION
MAX AMPERES Switch (S303)
FINE Control (R333)
POLARITY Switch (5304)
ZERO ADJ Control (R235)
INPUT
Receptacle
OUTPUT Receptacle (5102)
~ OVERLOAD Indicator (DS302)
(5202)
Functional Description
Controls power to instrument.
sets gain in Volts per ampere.
Sets
rise time Fn milliseconds.
Sets maximum suppression.
Adjusts suppression.
Sets polarity of suppression.
Adjusts output zero.
Input source connection.
Output connectFon.
Indicates overload condition.
Rear Panel Controls and Terminals
TABLE 1-2.
Paragraph
2-4, al
2-4, a2
2-4, a3
2-4, a4
2-4, a5
2-4, a6
2-4, a7
2-3, a
2-3, b
2-5, d
Control or Terminal Functional Description Paragraph
Line Switch (5301)
Power Receptacle
Fuse (F301)
0lJTPuT Receptacle
(P305)
(5103) Output connection.
Sets instrument for 117V or 23411.
Connection to line power.
Type 3AG Slow-Blow, 117V @ l/4 A (w-17)
234V @ l/S A (w-20)
WARNING
Using a Line Power Cord other than the one supplied
with your instrument may result in an electrical
shock hazard. If the Line Power cord is lost or
damaged,
replace only with Keithley Part No. CO-7.
2-4, b
2-3, c
2-3, b
0878
MODEL 427
GAIN
SWITC
S20
,----SUPPRESSION-,
FINE
ADJUST
GENERAL DESCRIPTION
POLARITY
SWITCH
INPUT
5202
ZERO ADJ
R235
FIG"RF 2.
OVERIDAD Power
.
DS302 S302
want Panel Controls.
P
OUTFW
5102
I
0471
FIGURE 3.
Rear Panel Controls.
OPERATION
MODEL 427
SECTION 2.
MEASUREMENT CONSIDERATIONS.
2-1.
a. Current-Detection Devices.
‘small electrical c~rrent8 has been the basis for a
number of instrumental methods used by the analyst.
Ion chambers, high-impedance electrodes, many forms
of ch=“metog=aphic detectors, phototubes and multipliers are coaronly-used t=ansduce=a which eequire the
measurement of small currents. Devices used for this
measurement a=e often called electraaeters.
b.. In any measure-
ment, if e”“=ce noise greatly exceeds that added by
the inst=“mentatian, optimization of instrumenteti””
is unimportant.
ical minimum, optimization of instrumentation charac-
teristics becomes imperative. TO determine the category into which this meas”=ement falls. the researcher
needs t” be familiar with the characteristics which
impoee theoretical and practical limitations on his
me*surement .
theoretical limitations present in voltage measueements
The noise inceeases with 8”“=ce realstance, and the
familiar equation for the mean-square noise voltage is
q = 4kTRAf Eq. 1
When source noise approaches theoret-
Most researchers a=e familiar with the
The DleasUrement of
OPERATION
Prom this equation it is irmnediately apparent that the
m%QB”rement of emall current =equi=es large values of
R, i.e., high impedance levels. Howwee, thfe gives
difficulties for meas”=ements requiring wide bandwidths
because of the RC time constant associated with a
high-megohm resistor and even a few picofarads of cir-
cuit capacitance.
generating a voltage across a parallel RC. The frequency response of this current measurement is limited
by the RC time constant.
end the -3 dB p”int “CC”=B at a frequency
Lor* noise and high
requirements.
techniques must be used which obtain high speed “sing
high-impedance devices.
C. Hiah Speed Methods.
1. High epeed can, af c”“=se, be obtained in .
shunt-type meae”=eme”t by “sing a low value for the
shunt resist”=.
resistor value int=ad”ces excessive noise into the
meQ8”rement.
Figure 4
speed,
TO optimize a current-measuring system,
As pointed ““t above, such a srmll
shows a c”==ent s”“=ce
Figure 5
therefore, a=e contradictory
shows this response
whe=e k is the Boltzma”” Constant, T ia the absolute
temperature of the s”“=ce resistance R, and
noise bendwidth( 3
single RC rolloff.) In the case of cureent measure-
ments it Is more appropriate to consider the noise
current generated by the ~l”“=ce and load resistances.
The mean squaee noise c”==ent generated by a resistor
is given by Eq. 2.
FIGURE 4.
In the shunt method c”=re”t is measured by
the voltage drop ac=“ee a resistor.
times the 3 dB bandwidth for a
Af
is the
2. A second method to achieve bandwidth is to
keep R large, to accept the frequency roll-off
starting at F”, and t” change the frequency eesponae
of the voltage amplifier a8 ehown in Figure 6a. The
combined effects of the RC time c”“sta”c folloved
by this amplifier is shown in Pigure 6b and it is
seen that the frequency response of the c”==ent
measurement has been extended to Pl. The frequency
at which the amplifier gain sta=‘ts to increase must
be exactly equal t” the frequency F” determined by
the RC time c”nstant in order for this approach t”
result in a flat frequency respanse. Therefore,
FO
FIGURE 5. The frequency respanse of the shunt method
1s limited by omnipresent ahunt capacitance.
LOG FREQUENCY
I
0471
MODEL 427
OPERATION
this oethad is ueeful only far application* where
the shunt capacitance C is constant. Aaide from
thin drawback this is * 1eSitimste approach which
is being wed in low-noise, high-speed current-
meesuring applicatians. In addition to current noise
in the *hunt and in the amplifier input stage, B
maJor source of noise in this system *ri*** from
the voltage-noise generator ssaociated with thb input atage (reflected a* current noise in the shunt
resistor) caused by the high-frequency peeking in
the following stages of amplification. More will be
said about this in the discussion on noise behavior.
3. A third method used for speeding up * current
measurement asas guarding techniques to eliminate
the effects of capacit*nces. Unfortunately only
certain type* of capacitance*, such ** cable cap=-
itances, can be conveniently eliminated in this
manner.
itences associated with the *ource itself became*
very cumbersome and m*y not be feasible in many in-
stsnces.
*re identical to those mentioned in the second
system.
4. A fourth circuit configuration combines the
capability of low-noise and high-speed performance
with tolerance for varying input C and eliminate*
need for separate guard by making the ground plane
*n effective guard. This is the current-feedback
technique.
ment of 3 over shunt technique*. Again, the major
sources of noise are identical to those mentioned
in the second system.
d. Noise in Current Measurements. Noise forms *
b*aic limitation in *nv hinh-speed current-measurinn
system. The shunt *y&m give; the simplest curren;
measurement but does not give low-noise performance.
A properly designed feedback *y*tem gives superior
noise - bandwidth performance. Noise in these two
systems will be discussed next.
1. Noise Behavior of the Shunt System. High
speed end low noise *r* contradictory requirements
in any current meesurement because *orw capacitance
is always present. The theoretical performance
limftetion of the shunt *yetem c*n be calculated **
To eli,r,inate the effect of parasitic c*p*c-
The major *ourc** of noise in this *“*tern
This technique gives * typical improve-
The rms thermal noise current (in) generated by *
resistance R is given by
Eq. 4
The equivalent noise bandwidth (.f) of * parallel SC
combination is Af = 1/(4RC) snd the eignal handwidth (3 dB bandwidth) F, = 1/(2nRC). For practical
purposes peak-to-peak noise is taken 88 5 times the
ml* value. The peak-to-peak noise current can now
be written a*
i
UPP =
In practice, e typical value for shunt cape.cit*nce
is 100 picofarads.
rule-of-thumb is obtained. The lowest ratio of
detectable current divided by signal bandwidth using
*hunt-techniques is 2-10-14 ampere/Hertz for B peakto-peak signal-to-noise ratio equal to 1. A coroll-
ary far this rule-of-thumb expresses the noise cur-
rent in term* of obtainable risetime (lo-SO% risetime tr = 2.2 RC). The lowest product of detectable
current and risetime using shunt technique* is 7 x
lo-l5 ampere seconds.
assumed that the voltage amplifier does not contribute noise to the measurement.
2. Noise Behavior of the Feedback System. There
are three *ource* of noise in the feedback system
that have to be looked at closely. The firat two,
input-stage shot noise and current noise from the
mea*urinS resistor, are rather straight-forward. The
third, voltage noise from the input device of the
amplifier, cau*e* *ome peculiar difficulties in the
measurement. Any resistor connected to the input
injects white current noise (Eq. 4). In the circuit
of Figure 7 the only resistor that is connected to
the input is the feedback resistor R. As in the
shunt system, R must be made large for lowest noi*e.
Beceuse this noise is white, the total contribution
can be calculated by equ.,ting Af to the equivalent
noise bandwidth of the system. The second *ource of
noise is the current noise from the amplifier input.
This component is essentially the shot noise asaociared with the gate leakage current (io) of the input
device. Its rms value equals . . .
2 x 10-9 F,
With this value the following
F
In this derivation it has been
Eq. 5
FIGURE 6.
0471
LOG FREWENCY
FO
ny tailoring the frequency response of the
amplifier (Pig. 6a) the frequency response
of the shunt method c*n be extended.
F>
FIGURE 6b.
FO
Extended frequency response.
LOG FREQUENCY
F,
OPERATION
MODEL 427
T;; = J-zTp-
where e is the electronic charge. The contribution
of this noise generator is also white. N*t only do
these two noise sources generate white current noise,
the noise in a given bandwidth is also independent
of the input capecitence C. The mejor source of
noise in e feedback current meesurement is the noise
contribution aseocisted with the voltage noise of
the input amplifier. The voltage noise ten be rep-
resented by a VOltage noise generator (0,) et the
emplifier input es shown in Figure 8. This wise
generator itself is assumed to be white. However,
its total noise contribution to the current-measuring
system is not white.
reveal that et low frequencies P large em*u*t of feedbeck ie applied around the voltage noise source {en).
However, the SC combination ettenuetes the high-
frequency components of V,,t so that no feedback is
present et high frequencies. Thus, the noise con-
tribution to the output voltage V,,t from the valtage noise source a* is no longer independent of
frequency. The noise is “colored” and increases in
intensity for ell frequencies higher than F,. The
resulting noise spectrum is shown in Figure 9b. The
tote1 system noise is related to the are* under
this curve.
plotted on the horizontal axis, the eree under the
curve et higher frequencies represents e significantly larger amount of noise then e similar eree
*t low frequencies.
the frequency response of the current measuring
system.
interesting et this point to compare this noise
spectrum with the frequency response of the voltage
amplifier in Figure 4 es shown in Figure 6a. A voltage noise eouec.e et the input of the amplifier would
generate a noise spectrum according to the amplifier
frequency response as shown in Figure 6a. The noise
spectrum of such e system, then, is identical to the
noise spectrum of the feedback system as given in
Figure 9h.
that signal-to-noise performance of a measurement
cen**t be improved by feedback techniques. At the
high-frequency end the voltege noise is limited by
the frequency FA which is the high-frequency roll-
off point of the operational amplifier. It should
Because the logarithm of frequency is
Figure 9e ia identical to Figure 6b. It is
This illustrates the well-known fact
Inspection of Figure 6 will
For comparison, Figure 9a show
he noted that even though the useful bandwidth of
the system extends only t* Fl, there era noise com-
ponents of higher frequency present. To obtain
best widebend-noise performance, these high-frequency
noise components have to be removed. This ca* be
achieved by adding a low-pass filter section follow-
ing the feedback input stage. If the band-pass of
thin low-pass filter is made adjuetable this filter
can nerve the dual purpose of removing high-frequency
noise end of limiting the signal bandwidth of the
system.
2-2. THEORY OF OPERATION.
8. Current Feedback Technique. The basic circuit
configuration used in the current-feedback technique
is shown in Figure 7.
current-measuring resistor R is placed in the feedback
loop of e* inverting emplifier with a gain of A*. The
frequency response obtained with this circuit is identical to thet s+nvn in Figure 6b. F* agein is the
frequency associated with the RC time constant:
F, =
The frequency response of the syetem is extended t* a
frequency fl where
F
, = AoF,
Note that the frequency rerponse is automatically flat
without heving to match break points. However, the
total bandwidth of the system (Fl) is still limited
by the value of the ahunt capacitance C across the
input.
back technique avoids the use of low values for R
which could generate exceesive current noise.
difficulty of the feedback system ariees from shunt
capacitence esaociated with the high-megohm resiaeor R
in the feedheck path.
the resistor is CFr then the bandwidth (FF) of the
system is determined by the time COnstent RCF:
This improved frequency response of the feed-
b. Refinements of the Feedback System. A major
In this configuration the
SE Eq. 6
Eq. 7
.
If the shunt cepacitence acroes
FIGURE 7.
6
Beslc circuit configuration for the feedback method.
FIGURE 8.
The voltage noise associeted with the amplifier input device is en important eourc~
of noise in the high-speed feedback syatew
0471
MODEL 427 OPERATION
FIGURE 9.
FIGURE 9b.
The bendwidth of the high-speed feedback
system (Fig. 9a) ten he limited by using
e filter with either e -6 dB/actave or a
-12 dB/octave roll-off. The effect of the
filter on the noise spectrum is showwin
Fig. 9h.
Effect of input capacitance on
noise is shown in Fig..9c.
Effect of filter on noise spectrum.
FIGURE 10.
Frequency compensation.
FF = 1
2 nllcp
Eq.
A slight modification of the feedback loop can correct
this problem es shown in Figure 10. If the time constant RlCl is made equal to the time constent R.CF,
it CB* be shown that the circuit within the dotted
line behaves exactly es a resistance R. The matching
of time constants in this cese does not become e drawbeck because the copscitances involved era all constant
and not effected by input impedance.
C. -12 dB/actave Filter.
1. Theory. To obtain optimum widehand noise per-
fomence e filter with e single high-frequency rolloff (i.e.,
dB/octeve is required.
-6 dB/octave) is not sufficient end -12
The effect of e -6 dB filter
is shown in Figure 9a end h. The filter is used to
limit the system bandwidth to a frequency F2, smaller
then Fl. The effect af this filter on the noise
spectrum is shown in Figure 9b. It ten be seen that
there ace egain high-frequency noise components above
F2, the useable bandwidth of the system. These can
he eliminated by using e filter with e -12 dB/octave
roll-off. The result of such .a filter on noise per-
formance is also shown in Figure 9b.
FIGURE 9c.
0471
Effect of input capacitance on noise.
2. Model 427. The input smplifier 18 followed by
en adjustable low-pass filter having e -12 dB/octeve
roll-off end a valtage gain of 10X. The voltage
gain in the low-pass filter avoids premature over-
loading in the input amplifier which ten be seen es
fallows. The maximum output voltage V,,t is $10
volts.
The maximum signal level et the input of the
low-pass filter is, therefore, +l volt. At this
point in the circuit, wide-band noise could still be
present end exceed the l-Volt signal level. The
voltege gain of 10 in the filter allows the total
pre-filter wide-hand noise to exceed the full scale
signal by e factor of 10 (20 dB). The frequency re-
sponse of this filter is edjustahle for variable
“damping” control.
7
OPERATION
2-3.
CONNECTIONS,
MODEL 427
2-5. OPERATING CONSIDERATIONS.
8.
Input.
type which metes with coaxial cables such es Keithley
Models 8201 end 8202.
high.
(5102 on the front, 5103 on the reer panel). These
era BNC type* where the inner contact is output high
end the outer shell is chassis ground.
rear panel is a 3-prong connector which metes with
Keithley pare number CO-6 line cord.
2-4. CONTROLS.
The outer shell is low or chassis ground.
b. Output.
C.
Power Input. The power receptacle (P305) on the
a. Front Panel.
1. Power Switch “PUSH ON” (5302). This switch
controls the line power to the instrument. ‘The
switch is a special pushbutton type with “Power On”
indicated by a self-contsined pilot lamp.
2. GAIN (VOLTS PER AMPERE) (S201). This switch
sets the overall gain in eight positions from 104
to loll.
ment of zero offsets.
3. RISE TIME Switch (5101). This switch sets
the lo-90% rise time in 10 positions from .Ol to
300 milliseconds(for the filter section only).
4. SUPPRESSION (MAX) Switch (S303). This switch
sets the maximum current suppression in eight pasitions from lo-10 to 10-3 A. When the switch
to “OFF” the current suppression circuit is disabled.
5. SUPPRESSION (FINE) control (~333). This con-
trol permits adjustment of suppression with 0.1%
resolution.
6. SUPPRESSION (POLARITY) Switch (S304). This
switch set* the polarity of the current suppression
(referred to the input).
7. ZERO ADJUST Control (R235). This control per-
mits adjustment of zero offset through the u*e of the
OVERLOAD indicator.
b. Rear Panel.
1. Line Voltage Switch (S301). Sets instrument
for either 117 or 234 V operation.
The input receptacle (5202) is e SNC
The inner contact is circuit
Two outp,ut receptacles *re provided
A “ZERO CHECK” position permits adjust-
Fuse RequirP.ment* 3AG, Sla-Slo
117V: 1/4A
234V: l/SA
Keithley No. F”17
Keithley No. F”-20
is
set
8. Gain. The gein of the Model 427 is defined in
terms of volts per *“pare. Since the output level is
10 volts for e full scale input, the gain could also
be expressed e* sensitivity in emperes referred to the
input BS in Table 2-1. Vout = - (Iin x GAIN)
Gain or Sensitivity Referred to the Input
GAIN
Setting Resistor
b. Rise Time. The rise time for each gain setting
is listed in the specifications a* “FAST RISE TIME”.
These rise times are obtained when the RISE TIME
switch is set to the positions indicated in Table 2-2.
GAIN
Setti”g
104
105
106
107
1 10;
1oy
1010
loll
c. Suppressian. Current suppression is provided in
the Model 427 for suppression of input currents up to
10e3 amperes.
variations in e larger signal can be observed. Currents
of either polarity can be suppressed. To suppress an
input current the SUPPRESSION should be *et to supply
e current of apposite polarity. The FINE control permits
adjustment up to 1.5 times the MAX setting.
d. Overloads,
en overload et two places in the circuit: before end
after the “RISE TIME” filter circuit. The OVERmAD
lamp (DS302) will indicate whenever the voltsge sensed
ia greater then full scale regardless of the RISE TIME
setting or the frequency.
a. Zero Adjust. The ZERO CHECK po*ition grounds the
the input of the instrument and co”“ert* the cuerent
amplifier to a high-gain voltage amplifier. The ampli-
fied offset voltage will turn on the OVERLOAD indicator
whenever the input voltage offset exceeds 5100 t0l.
Therefore the ZERO control should be adjusted so that
the OVERLOAD indicator is off when in ZERO CHECK mode,
yielding the specified input voltage drop.
Feedback
Switch Settings for “FAST RISE TIME”
Rise
Time
15 ps
15 us
15 I*8
40 us
60 ps .03 ms 800
220
400
1.5 1 ms 100
TABLE 2-l.
Full Scale
sensitivity Output
(Amperes)
103
104
105
106
107
108
109
1010
&Is
“S ps
By suppressing background currents, smell
The overload sensing circuit detects
1
x 10-3
1 x 10-4
1 x 10-5
I x 10-6
1 x 10‘7
1 x 10-g
1
x 10-9
1 x 10-10
TABLE 2-2.
RISE TIME
Setti”g*
.Ol “S
.Ol ms
.Ol “S
.03 “9
.l “S 400
.3 ms 200
Eq. 9
Full Scale
(Volts)
10
10
10
10
10
10
10
10
DYlWl”iC
Range
2000
2000
2000
2000
J
8
MODEL 427
APPLIChTIONS
SECTION 3. APPLICATIONS
3-1. CURRENT~MEASURING SYSTEM. The typical current
meaeuring system consists of a current source, a current amplifier, and a monitoring device, The current
source could include an ion chamber, photomultiplier,
or other high-impedance device,
such as the Model 427 provides sufficient gain to drive
a monitoring device such as a chart recorder or other
readout. The Model 427 in this case provides an output voltage which is calibrated in volts per ampere
a.3 in equation 10.
- (V,,t I GAIN)
Example :
3-2.
trates the trade-off between fast rise time and dynamic
range.
a.s the ratio of maximum peak-to-peak current to peakto-peak current noise.
taken as 5-times the rms current noise. The maximum
peak-to-peak current is 2-times the maximum full scale
current.
When using current suppression the current-suppres-
sion resistor should be considered as an additional
current-noise generator. The values given in Table
3-l do not include the contribution of the suppression resistor. Therefore the selected suppression
resistor Rs, should be as large as possible to min-
imize the contribution to current noise.
Iin =
GAIN = 106 voltslampe~e
” O”t = +500 In”
The input current Iin would be:
lin = - (5x10-~vo1ts/106vo1ts per ampere)
Iin = - 5 x 10-7
NOISE BANDWIDTH CONSIDERATIONS.
For this application dynamic range is defined
Peak-to-peak current noise is
NOTE
The current amplifier
Eq. 10
amperes
Table 3-l illus-
3-3. NOISE-IMPROVEMENT CONTOURS. The sensitivity
and speed of the Model 427 (for either d-c or a-c
measurements) can be compared to the best perfarm-
ante obtainable with the shunt method of measuring
current. The best “noise-risetime” product that
can be achieved for d-c measurements (with 100 pF
shunt capacitance) in a shunt system is 7 x lo-15
ampere-seconds.
2 x lo-l5 ampere-seconds (also with 100 pF shunt
capacitance).
(lock-in, etc.) the-degree of improvement is a func-
tion of shunt capacitance and operating frequency.
The achieveable imprownene over the shunt method
can be plotted in a graph similar to a set of noise
contours. Figure 11 shows the measured impravemene
(negative dB) that can be obtained with the Model
427 at a given frequency and shunt capacitance when
compared to an ideal (noiseless) amplifier Fn a shunt
system.
,
I
FIGURE 11. Plot of noise-improvement contours.
However the feedback system achieves
When used in a-c narrowband systems
FREOUENCY IHI1
RMS’Noise Current (Typical)1 as a Function of Gain and Rise-Time Setting
TABLE 3-l.
,vL.. I,.,“,.,
PULL SCALE
GAIN
VIA
104
105
106
107
108
109
1010
1011
1
With up to 100 pP input shunt capacitance.
KEY :
x = Filter Bandwidth is greater than current-amplifier bandwidth.
* = larger Rise Times are useful for increased filtering of the signal arid noise inherent in the source.
0471
CURRENT
AMPERES 300 100 30
10-3
10-4
10-5
m-6
10-7
104’
10-9
lo-1o *
They do not further improve the instrument noise contribution except when the input shunt capacitance
exceeds 100 DF.
* * * *
*
* * *
:
*
2x10-15 4x10-15 1x10-14 4x10-14 1x10-13 4x10-13 x
*
* *
*
1x1:-14
*
*
*x1:-14
Noise increases aa input shunt capacitance increases.
RISE TIME SETTING
10 3 1 .3
*
* * *
* *
* *
*
2do-13 2d0-13 5do-13 2~0-12 5x10-12 x x
5~0-14 2do-13 ~0-13 2do-12 x
MO-12 2x10-l2 5x10-12 1x10-11
lx&
*
1x10-8
1x10-9 1.2x10-9 4x10-9
1x10-10
1.5x10-11
.l .03 .Ol
1.2x10-8 4x10-8 1x10-7
1.2x10-10 4x10-10 1x10-9
2x10-11 1x10-10 x
4x10-11
x
x x
x x
1x10-B
x
9
ACCESSORIES
MODEL 427
SECTION 4.
4-l. GENERAL.
be used with the Model 427 eo provide additional convenience and versatility.
Description:
The Model 1007 is a dual rack mounting kit with over-
all dimensions 3-l/2 in. (64 mm) high and 19 in. (483
mm) wide.
of two Angle Brackets, one Mounting Clamp, and extra
mounting BCreWS.
The following Keithley accesaaries can
Model 1007 Rack Mounring hit
The hardware included in this kit consists
ACCESSORIES
OPERATING INSTRUCTIONS. A separate Instruction
4-2.
Manual is supplied with each accessory giving complete
operating information.
Application:
The Model 1007 co""erts any half-rack, style "M"
instrument from bench mounting to rack mounting in
a standard 19-inch rack.
for rack mounting 19-inch full rack width insfru-
ltE"tS.
The Model 1007 Rack Mounting Kit can be used to m"u"t
instruments of 11 inch or 14 inch depth.
should decide the position af the i"~tr"me"ts to be
rack mounted.
instruments positioned as shown and identified as
instrument “A” and "B".
The Assembly Inaeructions refer co
The kit may also be used
The user
10
Parts List:
Item
NO.
DWCl+ltiO”
22 Angle Bracket
23 Screw, 16-32 x
Phillips Pa" Hd
24 Mounting Clamp
25 Screw, %6-32 x
Phillips Pa" Hd
26 Kep Nut 116-32
27 Screw, 116-32 X
Phillips Pa" Hd
28 Screw, 116-32 x
Phillips Pa" Hd
5/8,
1,
l/2,
718,
VY
Keq'd
2
6
1 247988
1
3
2
1
Keithley
Part No.
27410B
-_
__
_-
__
__
0877
MODEL 427
Assembly Instructions:
ACCESSORIES
Model 1007 Dual Rack MountinS Kit
1. Before assembling the rack kit, determine the
pasition of each instrwnent. Since the inserumenfs
can be mounted in either location, their position
should be determined by the user’s meas”rement. The
following instructions refer to instruments “n” and
UB”
positio”ed as shorvn. For mounfinS 19-inch full
rack Width instruments, disregard steps 2 through 5.
2. Once the position of each instrument has been
determined, ebe “side dress” panels on both sides of
each instrument should be removed. Renxwal is
accomplished by looseninS the screwy (Item 8, oriSinal hardware) in two places. Slide the “side dress”
panels co the rear of the instrument to remove.
3. The mountinS clamp is installed on instrument
“A” using the oriSina1 hardware (Item 8). With the
screws removed, insert the “mounting clamp” behind
the “corner bracket” (Item 7) and replace the screws
to hold the mounting clamp in place.
4. Tighten the screwy (Item 8) on instrument “B”.
Insert the “mounting clamp” behind the “corner
bracket” (Item 7) an instrument “8” a8 shown.
5. When mounting instruments having the same depth,
a screw (Item 25) and kep nut (Item 26) are required
to secure the two instruments together. When ,,,ouneing instruments of different depth, da not use kep
nut (Item 26) but substitute shorter screw (Item 28).
6. Attach an “anSle bracket” (Item 22) on each
instrument using hardware (Item 23) in place of the
original hardware (Item 8). For 14 in. long instru-
ments use 116-32 x 518 Phillips screw (Item 23) with
116-32 kep “uf (Item 26).
7. The bottom cover feet and tilt bail assemblies
may be removed if necessary.
8. The original hardware, side dress panels, feet
and tilt bail assemblies should be retained for fut-
ure conversion back to bench mountine.
0777
11