Page 1
THAT Corporation
Low-voltage, Low-power
Analog Engine
®
Dynamics Processor
THAT 4311
FEATURES
High Performance VCA, RMS-Level
·
Detector, and three 0pamps in one
package
Wide Dynamic Range: >105 dB
·
Low THD: <0.09%
·
Low Power: 7 mA typ.
·
Surface-Mount Package
·
5 VDC Operation
·
Description
The THAT 4311 Low Power Dynamics Proces
sor combines in a single IC all the active circuitry
needed to construct a wide range of dynamics
processors. The 4311 includes a high performance, voltage controlled amplifier, a log responding RMS-level sensor and three opamps,
one of which is dedicated to the VCA, while the
other two may be used for the signal path or control voltage processing.
The exponentially-controlled VCA provides
two opposing-polarity, voltage sensitive control
ports. Dynamic range exceeds 105 dB, and THD
is typically 0.09% at 0dB gain. The RMS detector
provides accurate RMS to DC conversion over an
APPLICATIONS
Wireless microphone systems
·
Wireless in-ear monitors
·
Compressors and Limiters
·
Gates
·
De-Essers
·
Duckers
·
-
80 dB dynamic range.
Though originally designed for use in micro
phone noise reduction systems, the 4311 is a use
ful building block in a number of analog signal
processing applications. The combination of exponential VCA gain control and logarithmic detector response - “decibel-linear” response simplifies the mathematics of designing the control paths of dynamics processors, making it easy
to develop audio compressors, limiters, gates, expanders, de-essers, duckers, and the like. The
high level of integration ensures excellent temperature tracking between the VCA and the detector,
while minimizing the external parts count.
-
-
Pin Name DMP20
RMS IN 1
)2
IT (I
18
19
OA1
20
17
16 15
EC-
IN
THAT4311
1
IN
RMS
IT
2
CT
OUT
3
5
4
14 13 12
SYM
EC+
OUT
VCA
OA2
11
VCC
OA3
VREF
VREF
VEE
6
7
9
8
10
Figure 1. THAT 4311 equivalent block diagram
TIME
OA2 -IN 3
RMS OUT 4
)5
CT (C
TIME
CLIP 6
OA2 OUT 7
CAP 8
VREF 9
VEE 10
VCC 11
OA3 OUT 12
VCA OUT 13
SYM 14
EC+ 15
EC- 16
VCA IN 17
OA1 OUT 18
OA1 -IN 19
OA1 +IN 20
Table 1. THAT 4311 pin assignments
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA
Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com
Page 2
Page 2 Low-voltage, Analog Engine
Preliminary Information
®
Dynamics Processor
SPECIFICATIONS
1
Absolute Maximum Ratings (TA= 25°C)
Positive Supply Voltage (VCC) +15 V Power Dissipation (PD)(TA= 75°C) 700 mW
Operating Temperature Range (T
Max DE
-(EC-) ± 1V
CEC+
) -20 to +70°C Storage Temperature Range (TST) -40 to +125°C
OP
Recommended Operating Conditions
Parameter Symbol Conditions Min Typ Max Units
Positive Supply Voltage V
CC
Electrical Characteristics
Parameter Symbol Conditions Min Typ Max Units
Supply Current I
Reference Voltage V
CC
REF
No signal; VCC=+7 VDC — 7.0 9.0 mA
+5 +15 V
2
1.8 1.95 2.1 V
Encode and Decode – Companding Noise Reduction (VCC= +7V encoder, +15V decoder)
Encode Level Match LMe Encode mode; f = 1kHz -25.3 -23.0 -20.7 dBV
Encode Gain Accuracy Encode mode, f = 1kHz
Decode Level Match LMd Decode mode; f = 1kHz -18.3 -16.0 -13.7 dB
Decode Gain Accuracy Decode mode; f=1kHz
Max Input Voltage Vime Encode mode; THD = 3%; f = 1kHz 3 5 — dBV
Max Output Voltage Vomd Decode mode; THD = 3%; f = 1kHz 10.7 13.7 — dBV
Total Harmonic Distortion THDtrim End-to-end; Vin = LMe; f = 1kHz — 0.025 — %
(with trim)
Total Harmonic Distortion THDnotrim End-to-end; Vin = LMd; f = 1kHz — 0.15 0.7 %
(no trim)
Output Noise Vnod End-to-end ; Vin = short; A-weighted — 7 — µVrms
GAe1 Vin = LMe + 10dB +3.5 +5 +6.5 dB
GAe2 Vin = LMe - 40dB -23 -20 -17 dB
GAd1 Vin = LMd + 5dB +8.5 +10 +11.5 dB
GAd2 Vin = LMd - 20dB -43 -40 -37 dB
1. All specifications are subject to change without notice.
2. Unless otherwise noted, T
=25°C, test circuit as shown in Fig 2.
A
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA
Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com
Page 3
Rev. 08/30/01 Page 3
Preliminary Information
Electrical Characteristics (con’t)
Parameter Symbol Conditions Min Typ Max Units
Op amp OA1
Offset Voltage V
Equivalent Input Noise V
nOA1
Total Harmonic Distortion THD
Open Loop Gain A
VO-OA1
Gain Bandwidth Product GBW
Slew Rate SR
Op amp OA2
Offset Voltage V
Equivalent Input Noise V
Total Harmonic Distortion THD
Open Loop Gain A
Gain Bandwidth Product GBW
Slew Rate SR
nOA1
VO-OA1
IO
OA1
IO
OA1
OA1
OA1
OA1
OA1
RL=2kW — ±0.5 ±6 mV
A-weighted 6.5 10
nV
1kHz, AV=1; RL= 10kW — 0.0007 0.003 %
RL= 10kW--115 --
at 50kHz — 5 —
—2—
RL=2kW — ±0.5 ±6 mV
A-weighted 7.5 12
nV
1kHz, AV=1; RL= 10kW — 0.0007 0.003 %
RL= 10kW--110 --
at 50kHz — 5 —
—2—
Hz
Hz
Input
XLR1
XLR-F
2
5
4
TP1
RMS Input
+40dB +60dB
+20dB
+
R10
100R 5%
Bypass
Capacitors
D3
1N4004
D4
1N4004
C15
1000u
+
0dB
V+
+
C10
22u
+
C13
22u
V+
Sym
1
3
3
1
2
R1
100k
External
Control
Input
C6
+
47u
R11
23k2
3
2
1
CN1
Iset2TC
THAT4311
R12
261k
V-
R2
10k0
R21
10k0
U1B
IN1OUT
RMS
R3
10k0
C12
3u3
R20
10k0
C9
3u3
R26
4
5
100R
5%
+
C7
10u
VREF
50k
R7
C4
47u
R8
cw
300k
V-
VREF
R5
17
20k0
R4
1k33
2
3
81
R29
1k33
TP2
RMS Output
RMS Output
3
2
1
CN1
5%
R9
51R
5%
1415
EC+ SYM
OUT
IN
EC-
16
VCA
VREF
CONTROL-VOLTAGE
6
U1
OP-27
VREF
Power
Input
3
2
1
CN2
13
C5
47p
R6
20k0
U1A
OA3
THAT4311
NPO
12
C11
100n
(U1)
V-
R24
2k80
SW1F
C19
1000u
SW1C
16
R15
31k6
C14
100n
(U1)
611
C16
47u
R23
280R
512
SW1E
R19
100k
+
R18
10k0
1
+40dB
R16
280R
C1
10u
R22
30R1
V+
10k0
3
VREF
V+
+
R27
19
20
+
C18
47u
R17
31k6
C8
OA2
U1C
THAT4311
U1E
11
VCC
VEE
10
THAT4311
R14
31k6
OA1
U1D
THAT4311
47p
SW1B
2
6
7
VREF
C17
47p
VREF
CAP
15
10u
Output
SW1A
XLR2
XLR-M
1
3
3
1
2
5
2
4
R25
18
100R
5%
R28
100R 5%
VREF
9
8
+
+
C3
C2
22u
Fig 2. THAT 4311 test circuit
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA
Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com
Page 4
Page 4 Low-voltage Analog Engine
Preliminary Information
Representative Data (Stand-alone)
®
Dynamics Processor
Fig 3. VCA Gain vs. Control Voltage (Ec-) at 25°C
Fig 5. VCA 1kHz THD+Noise vs. Input, +15 dB Gain Fig 6. VCA 1kHz THD+Noise vs. Input, 0 dB Gain
Fig 4. VCA 1kHz THD+Noise vs. Input, -15 dB Gain
Fig 7. VCA THD vs. Frequency, 0 dB gain, 1 Vrms Input
Fig 9. Departure from Ideal Detector Law vs. Level
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA
Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com
Fig 8. RMS Output vs. Input Level, 1 kHz & 10 kHz
Fig 10. Detector Output vs. Frequency at Various Levels
Page 5
Rev. 08/30/01 Page 5
Preliminary Information
Representative Data (Companding Noise Reduction)
Fig 11. End-to-End Transfer Function, 1kHz Fig 12. End-to-End THD+N
Fig 13. Encoder Transfer Function, 1kHz Fig 14. Encoder Frequency Response 20-20kHz
+5
Encoder
In
C16
10u
C2
3u3
R2
200k
+
+
V+
V+
R1
200k
11
Vcc
10
Vee
THAT4311
20
19
R13
2k
C1
10n
U1E
Vref
U1D
THAT4311
+
_
18
R3
8k06
9
Vref
8
Cap
+
C3
10u
+
3u3
Vref
C4
R5
15k
+
Vref
C8
22u
17
R9
51R
15
EC+ SYM
VCA
IN
EC-
C7
10u
16
14
OUT
Vref
4
5
+
R8
50k
C5
270p
R6
13
30k
___
OA3
+
U1A
THAT4311
U1B
OUT
RMS
Iset
TC
THAT4311
R7
20k
22kHz 3 pole BW filter
R10
6k19
C9
570p
R4
12
6k19
C10
4n7
R31
4k32
3
Vref
R11A
C6
1
+
IN
2
10u
20k
6
7
OA2
+
U1C
THAT4311
R11B
10k
optional
VR1
20k
R12
261k
Fig 15. THAT 4311 Noise Reduction Encoder Schematic
Encoder
Out
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA
Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com
Page 6
Page 6 Low-voltage, Analog Engine
Preliminary Information
+15
®
Dynamics Processor
Decoder In
C6
10u
C16
R9
51R
+
10u
R11
15k
R12
261k
+
U1B
1
IN
RMS
Iset2TC
THAT4311
OUT
4
5
+
C5
10u
C4
3u3
U1E
Vref
R5
24k3
Vref
Cap
V+
+
11
Vcc
10
Vee
THAT4311
Fig 16. THAT 4311 Noise Reduction Decoder Schematic
EC+ SYM
17
IN
9
8
+
C2
10u
VCA
EC-
16
OUT
+
Theory of Operation
150k
131415
Vref
C3
22u
R8
R2
56k
U1A
_
OA3
+
THAT4311
R3
8k87
12
R4
7k5
R7
20k
R1
6k04
R6
24k3
19
_
OA1
20
+
U1D
THAT4311
+
C7
10u
C1
3n3
18
Decoder Out
The THAT 4311 Analog EngineâDynamics Processor combines THAT,s proven Voltage-Controlled
Amplifier (VCA) and RMS-Level Detector designs with
three opamps to produce a multipurpose dynamics
processor useful in a variety of applications. For de
tails of the theory of operation of the VCA and RMS
Detector building blocks, the interested reader is re
ferred to THAT Corporation’s data sheets on the
218x Series VCAs and the 2252 RMS-Level Detector.
Theory of the interconnection of exponen
tially-controlled VCAs and log-responding level detec
tors is covered in THAT Corporation’s application
note AN101, The Mathematics of Log-Based Dynamic
Processors.
The VCA - in Brief
The THAT 4311 VCA is based on THAT Corpora
tion’s highly successful complementary log/anti-log
gain cell topology, as used in THAT’s 218x and
215x-Series IC VCAs. The THAT 4311 is integrated
using a fully complementary, BiFET process. The
combination of FETs with high-quality, complemen
tary bipolar transistors (NPNs and PNPS) allows ad
ditional flexibility in the design of the VCA over
previous efforts.
Input signals are currents to the VCA IN pin.
This pin is a virtual ground biased at VREF, so in
normal operation an input voltage is converted to input current via an appropriately sized resistor (R5 in
-
Fig 2, Page 3). Because the current associated with
DC offsets relative to VREF present at the input pin
-
and any DC offset in preceding stages will be modu
lated by gain changes (thereby becoming audible as
thumps), the input pin is normally AC-coupled (C4 in
-
Fig 2).
The VCA output signal is also a current, inverted
with respect to the input current. In normal opera
tion, the output current is converted to a voltage via
inverter OA3, where the ratio of the conversion is de
termined by the feedback resistor (R6, Fig 2) con
-
nected between OA3’s output and its inverting input.
The signal path through the VCA and OA3 is
non-inverting.
The gain of the VCA is controlled by the voltage
applied between EC- and the combination of EC+
and SYM. Gain (in decibels) is proportional to EC-,
-
-
provided that EC+ and SYM are at VREF. The con
stant of proportionality is -6.1mV/dB (for 5V sup
-
plies) for the voltage at EC-, and 6.1mV/dB for the
voltage at EC+, and SYM
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA
Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com
Page 7
Rev. 08/30/01 Page 7
Preliminary Information
+5
R5
20k
Signal In
C3
+
47u
264k
R6
10k
R7
U1B
1
IN
RMS
2
Iset
THAT 4311
+5
+
C6
OUT
TC
RMS Out
4
5
+
C4
10u
10u
C7
100n
U1E
11
Vcc
Vref
10
Vee
Vref
Cap
9
8
THAT 4311
Fig 17. Circuit showing gain control at E
As mentioned, for proper operation, the same
voltage must be applied to EC+ and SYM, except for
a small (±2.5 mV) DC bias applied between these
pins. This bias voltage adjusts for internal mismatches in the VCA gain cell which would otherwise
cause small differences between the gain of positive
and negative half-cycles of the signal. The voltage is
usually applied via an external trim potentiometer
(R7 in Fig 2), which is adjusted for minimum signal
distortion at unity (zero dB) gain.
The VCA may be controlled via EC-, as shown in
Fig 17, or via the combination of EC+ and SYM.
This connection is illustrated in Fig 18. Note that
this latter figure shows only that portion of the cir
cuitry needed to drive the positive VCA control port;
circuitry associated with OA1, OA2 and the RMS de
tector has been omitted.
While the 4311’s VCA circuitry is very similar to
that of the THAT 2180 Series VCAs, there are several
important differences, as follows:
1. Supply current for the VCA is fixed internally.
Approximately 500mA is available for the sum of in
put and output signal currents.
2. The signal current output of the VCA is inter
nally connected to the inverting input of an on-chip
C1
+
47u
R1
20k
Vref
Control Port Drive
+
C5
10u
+
C8
22u
opamp. In order to provide external feedback
around this opamp, this node is brought out to a pin.
3. The input stage of the 4311 VCA uses integrated P-channel FETs rather than a bias-current
corrected bipolar differential amplifier. Input bias
currents have therefore been reduced.
The RMS Detector - in Brief
The THAT 4311’s detector computes RMS level
by rectifying input current signals, converting the rec
tified current to a logarithmic voltage, and applying
that voltage to a log-domain filter. The output signal
is a DC voltage proportional to the decibel-level of the
-
-
-
-
RMS value of the input signal current. Some AC
component (at twice the input frequency) remains su
perimposed on the DC output. The AC signal is at
tenuated by a log-domain filter, which constitutes a
single-pole roll-off with cutoff determined by an ex
ternal capacitor and a programmable DC current.
As in the VCA, input signals are currents to the
RMS IN pin. This input is a virtual ground biased at
VREF, so a resistor (R11 in Fig 2) is normally used to
convert input voltages to the desired current. The
level detector is capable of accurately resolving sig
nals well below 10mV (with a 10kW input resistor).
However, if the detector is to accurately track such
low-level signals, AC coupling is normally required.
17
U1C
3
Vref
R4
51k
R3
51R
15
EC+ SYM
VCA
IN
EC-
OA2
THAT 4311
C-
R2
20k
C2
47p
13
14
OUT
16
U1A
OA3
THAT 4311
Signal Out
12
Vref
U1D
6
7
19
OA1
20
THAT 4311
18
Vref
-
-
-
-
-
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA
Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com
Page 8
Page 8 Low-voltage, Analog Engine
Preliminary Information
®
Dynamics Processor
U1B
1
2
IN
Iset
RMS
N/C
N/C N/C
OUT
TC
4
N/C
5
THAT 4311
Control Port Drive
Signal In
+5
C7
100n
11
10
Vcc
Vee
C1
+
47u
U1E
Vref
R1
20k
Vref
Cap
THAT 4311
+5
R5
50k
R4
51k
R3
R2
20k
C2
51R
47p
16
OUT
13
U1A
OA3
THAT 4311
Vref
17
IN
15 14
EC+ SYM
VCA
EC-
Vref
U1C
9
8
++
C5
10u
C8
22u
N/C
3
Vref
Fig 18. Circuit showing gain control at E
12
6
7
OA2
THAT 4311
C+
Signal Out
U1D
N/C
19
N/C
N/C
20
18
OA1
THAT 4311
N/C
The log-domain filter cutoff frequency is usually
placed well below the frequency range of interest.
For an audio-band detector, a typical value would be
5Hz, or a 32ms time constant (t). The filter’s time
constant is determined by an external capacitor attached to the CT pin, and an internal current source
(I
) connected to CT. The current source is pro
TIME
grammed via the IT pin: current in IT is mirrored to
with a gain of approximately one. The resulting
I
TIME
time constant t is approximately equal to
(0.026 ´ CT) / I
matics of rms detection, the attack and release time
constants are fixed in their relationship to each other.
The DC output of the detector is scaled with the
same constant of proportionality as the VCA gain
control: 6.1mV/dB. The detector’s zero dB reference
(Iin0, the input current which causes zero volts out
put), is determined by IT as follows: Iin0=IT. The
detector output stage is capable of sinking or sour
cing l00mA.
Differences between the 4311’s RMS-Level Detec
tor circuitry and that of the THAT 2252 RMS Detec
tor are as follows:
1. The rectifier in the 4311 RMS Detector is inter
nally balanced by design, and cannot be balanced via
an external control. The 4311 will typically balance
positive and negative halves of the input signal within
. Note that, as a result of the mathe
T
±1.5%, but in extreme cases the mismatch may
reach +20%. However, a 20% mismatch will not significantly increase ripple-induced distortion in dynamics processors over that caused by signal ripple
alone.
2. The time constant of the 4311’s RMS detector
-
-
-
-
-
-
-
is determined by the combination of an external ca
pacitor (connected to the CT pin) and an internal,
programmable current source. The current source is
equal to IT. Normally, a resistor is not connected di
rectly to the CT pin on the 4311.
3. The zero dB reference point, or level match, is
not adjustable via an external current source. How
ever, as in the 2252, the level match is affected by the
timing current, which, in this case, is drawn from the
IT pin and mirrored internally to CT.
4. The input stage of the 4311 RMS detector uses
integrated P-channel FETs rather than a bias-current
corrected bipolar differential amplifier. Input bias
currents are therefore negligible, improving perfor
mance at low signal levels.
The Opamps - in Brief
The three opamps in the 4311 are intended for
general purpose applications. All are 5MHz opamps
with slew rates of approximately 2V/ms. All use bipo
lar PNP input stages. However, the design of each is
optimized for its expected use. Therefore, to get the
-
-
-
-
-
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA
Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com
Page 9
Rev. 08/30/01 Page 9
Preliminary Information
+5
In
Threshold
Vin0=-10dBu
C3
R6
+
28k7
47u
R7
267k
+5
R12
20k
U1B
1
IN
RMS
2
Iset
THAT 4311
OUT
R5
20k
C1
+
R10
82k
R9
10k
C7
R11
51k
R8
4
4k99
5
TC
+
C4
10u
CR1
1N4148
3
OA2
Vref
C6
+
10u
100n
6
7
U1C
THAT4311
CR2
1N4148
+5
Gain
R14
1k43
R13
10k
Vref
U1E
11
Vcc
Vref
10
Vee
THAT 4311
47u
+5
R18
50k
Compression
Ratio
9
Vref
8
Cap
++
10u
20k
R17
82k
R15
10k
Vref
C8
C9
22u
R1
19
20
Vref
17
IN
R16
4k99
C5
100n
U1D
OA1
THAT4311
R4
33k
R3
51R
15 14 13
SYM
EC+
VCA
EC-
16
18
OUT
Vref
R2
20k
C2
47p
U1A
12
OA3
THAT4311
Out
Fig 19. Simple compressor / limiter using the THAT 4311
most out of the 4311, it is useful to know the major
differences among these opamps.
OA3, being internally connected to the output of
the VCA. is intended for current-to-voltage conversion. Its input noise performance, at
75./nV Hz
complements that of the VCA, adding negligible noise
at unity gain. Its output section is capable of driving
1mA into a 2kW load.
OA1 is the quietest opamp of the three, and with
its typical input referred noise of
65./nV Hz
,isthe
opamp of choice for input stages. Its output section
is nominally capable of driving 3mA into a 5kW load.
OA2 is best suited for control voltage processing,
though is does have anti-paralleled diodes that can
Application Information
As noted previously, the THAT 4311 was origi
nally designed for noise reduction systems, hence the
emphasis on those parameters in the specifications.
Its low power consumption, integration, and similar
ity to the THAT 4301, however, extend its utility to a
variety of other products and applications. The cir
cuit of Fig 19, shows a typical application for the
THAT 4311. This simple compressor/limiter design
features adjustable hard-knee threshold, compres
be used to fashion it into a clipper. (However in most
applications where a clipper is needed, it’s preferable
to place it around OA3). OA2’s input noise is comparable to OA3 and its output drive is comparably to
,
OA1.
The Reference Voltage - In Brief
THAT Corporation’s log/anti-log VCAs and RMS`
detectors require a reference voltage between the pos
itive and negative power supplies, and to supply this,
the THAT 4311 provides an on-chip, 2V reference
about which the VCA, the RMS detector, and OA2 are
biased. This reference is a buffered band-gap refer
ence that is amplified to 2V. Pins are provided for fil
ter capacitors at both the input and the output of the
buffer, which are labeled CAP and VREF respectively.
-
sion ratio, and static gain. The applications discus
sion in this data sheet will center on this circuit for
the purpose of illustrating important design issues.
-
-
Signal Path
As mentioned in the section on theory, the VCA
input pin is a virtual ground with negative feedback
provided internally. An input resistor (R1, 20kW)is
-
required to convert the AC input voltage to a current
-
-
-
-
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA
Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com
Page 10
Page 10 Low-voltage, Analog Engine
Preliminary Information
®
Dynamics Processor
within the linear range of the THAT 4311. (Peak VCA
input currents should be kept under 250 mA for best
distortion performance.) The coupling capacitor (Cl,
47 mF) is strongly recommended to block DC current
from preceding stages (and from offset voltage at the
input of the VCA). Any DC current into the VCA will
be modulated by varying gain in the VCA, showing up
in the output as “thumps”. Note that Cl, in conjunc
tion with R1, will set the low frequency limit of the
circuit.
The VCA output is connected to OA3, configured
as an inverting current-to-voltage converter. OA3’s
feedback components (R2, 20 kW, and C2, 47 pF) de
termine the constant of current-to-voltage conversion.
The simplest way to deal with this is to recognize that
when the VCA is set for unity (zero dB) gain, the in
put to output voltage gain is simply R2/R1, much like
the case of a single inverting stage. If, for some rea
son, more than zero dB gain is required when the
VCA is set to unity, then the resistors may be skewed
to provide it. Note that the feedback capacitor (C2) is
required for stability. The VCA output has approximately 45pF of capacitance to ground, which must be
neutralized via the 47pF feedback capacitor across
R2.
The VCA gain is controlled via the EC- terminal,
whereby gain in dB will be proportional to the negative of the voltage at EC-. In this application the EC+
terminal is tied to VREF, though it could be the
driven port, or the control ports could be driven differentially. The SYM terminal is returned nearly to
the EC+ terminal (which is in this case VREF) via a
small resistor (R3, 51W). The VCA SYM trim (R5,
20kW) allows a small voltage to be applied to the
SYM terminal via R4 (33kW). This voltage adjusts for
small mismatches within the VCA gain cell, thereby
reducing even-order distortion products. To adjust
the trim, apply to the input a middle-level, mid
dle-frequency signal (1kHz at 200mV
rms
is a good
choice with this circuit) and observe THD at the sig
nal output. Adjust the trim for minimum THD.
RMS-Level Detector
The RMS detector’s input is similar to that of the
VCA. An input resistor (R6, 28.7kW) converts the AC
input voltage to a current within the linear range of
the THAT 4311. The coupling capacitor (C3, 47mF)
is recommended to block the current from preceding
stages (and from offset voltage at the input of the de
tector). Any DC current into the detector will limit
the low-level resolution of the detector, and will upset
the rectifier balance at low levels. Note that, as with
the VCA input circuitry, C3 in conjunction with R6
will set the lower frequency limit of the detector.
The time response of the RMS detector is deter
mined by the capacitor attached to CT (C4, 10 uF)
and the size of the current in pin IT (determined by
R7, 267 kW and VREF, 2V). Since the voltage at IT is
approximately 2V, the circuit of Figure 19 produces
-
7.5 mA in IT, The current in IT is mirrored to the CT
pin, where it is available to discharge the timing ca
pacitor (C4). The combination produces a log filter
with time constant equal to approximately 0.026
CT/IT (~35 ms in the circuit shown).
-
-
-
The waveform at CT will follow the logged (deci
bel) value of the input signal envelope, plus a DC off
set of about 2V
plus VREF or about 3.3V. The
BE
capacitor used should be a low-leakage, electrolytic
type in order not to add significantly to the timing
current.
The output stage of the RMS detector serves to
buffer the voltage at CT and removes the 1.3 V
-
-
DC
(2 VBE) offset, resulting in an output centered around
VREF for input signals of about 245 mV
rms
,or
-10 dBu. The output voltage increases 6.1 mV for every 1 dB increase in input signal level. This relationship holds over more than a 60 dB range in input
currents.
Control Path
The primary function of an audio compressor is
to reduce a signal's dynamic range. A 2:1 compressor reduces a 100 dB dynamic range to 50 dB. A
limiter, or infinite compressor, is a special case of
compressor where the dynamic range is reduced to
the point where the rms level of the signal is con
stant. This reduction in dynamic range is accom
plished by a) raising the gain when the signal is
below some particular level -- often referred to as the
'zero dB reference level' -- and b) reducing the gain
-
when the signal is above that level. In addition, these
devices often have a threshold, below which the sig
-
nal is passed unprocessed and above which compres
sion takes place. This feature keeps the noise floor
from rising to noticeable levels in the absence of sig
nal.
We previously established that the zero dB refer
ence level of the detector is -10 dBu (zero dB refer
ence level = 7.5 , R6 = 28.7 kilohms). Neglecting the
effect of the threshold control (R11 and R12), when
the output is below this level the output of OA2 is
driven high, forward biasing CR1 and reverse biasing
-
CR2. Since CR2 is not conducting, no signal is
passed to the VCA's control port by OA1. When the
signal level exceeds -10 dBu, the output of the RMS
detector goes positive, and CR2 begins to conduct. In
-
-
-
-
-
-
-
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA
Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com
Page 11
Rev. 08/30/01 Page 11
Preliminary Information
this case, OA2's feedback is provided through R9,
and the sensitivity at this point is 12.2 mV/dB, since
the output of the RMS detector is multiplied by
-R9/R8, or a gain of -2.
A threshold control is provided to vary the
threshold above or below -10 dBu. The output sensi
tivity of the RMS detector is 6.1 mV/dB. This is con
verted to a current by R8, and the sensitivity at the
summing node of OA2 is
mV
61
.
dB
12
.
=m
kAW
499
.
The wiper of R12 can swing between -2V and +3V
relative to the summing node of OA2 which is at
VREF. If we want the threshold to swing as high as
+30 dB, then the value required for R11 can be cal
culated as
V
Rk
11 51
2
=»
A
m
12 30
dB
´.
dB
W
when rounded to the nearest 5% resistor value.
Using this value and knowing that the pot's swing in
the other direction is 3V, we can calculate the thresh
old swing in the negative direction to be
3
V
51
k
W
49
dB
»-
A
m
12
.
dB
Since the zero dB reference level of the detector is
-10 dBu, the threshold can be adjusted from 20 dBu
to -59 dBu.
The output of the threshold detector represents
the signal level above the determined threshold, at a
constant of about 13mV/dB (from [R9/R8]
6.1mV/dB). This signal is passed on to the COM
PRESSION control (R13), which variably attenuates
the signal passed on to OA1. Note that the gain of
OA1, from the wiper of the COMPRESSION control to
OA1’s output is R16/R15 (0.5), precisely the inverse
of the gain of OA2. Therefore, the COMPRESSION
control lets the user vary the above threshold gain be
tween the RMS detector output and the output of
OA1, from zero to a maximum of unity.
The gain control constant of the VCA (6.1mV/dB)
is exactly equal to the output scaling constant of the
RMS detector. Therefore, at maximum COMPRES
SION, above threshold, every dB increase in input
signal level causes a 6.1mV increase in the output of
OA1, which in turn causes a 1dB decrease in the VCA
gain. With this setting, the output will not increase
despite large increases in input level above threshold.
This is infinite compression. For intermediate set
tings of COMPRESSION, a 1dB increase in input sig
nal level will cause less than a 1dB decrease in gain,
thereby varying the compression ratio.
The resistor R14 is included to alter the taper of
the COMPRESSION pot to better suit common usage.
If a linear taper pot is used for R13, the compression
ratio will be 1:2 at the middle of the rotation. How
ever, 1:2 compression in an above-threshold com
-
pressor is not very strong processing, so 1:4 is often
preferred at the midpoint. R14 warps the taper of
-
-
R13 so that 1:4 compression occurs at approximately
the midpoint of R13’s rotation,
The GAIN control (R18) is used to provide static
gain or attenuation in the signal path. This control
adds between 120mV and -180mV of offset to the
output of OA1, which is approximately a -20dB to
+30dB change in gain of the VCA. The gain control
signal is passed into OA1 via R17, but this signal is
also passed back to the threshold amplifier (OA2) via
-
R10. This arrangement results in the threshold be
ing fixed relative to the output. In other words, as
the gain is increased, the threshold is lowered to
keep the threshold of compression or limiting at the
same output level. This is particularly important in
limiters, since it keeps the gain control from interact
-
ing with the threshold.
-
C5 is used to attenuate the noise of OA1, OA2,
and the resistors R8 through R16 used in the control
path. All these active and passive components produce noise which is passed on to the control port of
the VCA, causing modulation of the signal. By itself,
the THAT 4311 VCA produces very little noise modulation, and its performance can be significantly degraded by the use of noisy components in the control
voltage path.
-
Overall Result
The resulting compressor circuit provides
hard-knee compression above threshold with three
essential user adjustable controls. The threshold of
compression may be varied over a range from about
-
-58dBu to +20dBu. The compression ratio may be
varied from 1:1 (no compression) to ¥:1. And, static
gain may be added between -20 and +30dB. Audio
performance is excellent, with THD running below
0.1% at middle frequencies even with 10 dB of com
-
pression, and an input dynamic range of over 105dB.
-
Perhaps most important, this example design
only scratches the surface of the large body of appli
cations circuits which may be constructed with the
THAT 4311. The combination of an accurate, wide
dynamic range, log-responding level detector with a
-
-
high-quality, exponentially-responding VCA produces
a versatile and powerful analog engine. These, along
with its on-board opamps, allow a designer to con
struct, with a single IC and a handful of external
components, gates, expanders, de-essers, noise re
duction systems and the like.
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA
Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com
Page 12
Page 12 Low-voltage, Analog Engine
Preliminary Information
Package Information
The THAT 4311 is available in a 20-pin surface
mount package. The package dimensions are shown
in Fig 20 while the pinout is given in Table 1.
0-10º
C
B
1
D
F
A
E
J
H
G
I
®
Dynamics Processor
Item Millimeters Inches
10.0 ± 0.3 0.394 ± 0.012
A
5.0 ± 0.2
B
6.8 ± 0.4
C
0.35 ± 0.1
D
0.95
E
0.87 MAX
F
1.6±0.015
G
0.15 ± 0.1
H
0.5 ± 0.2
I
0.15 +0.1 -0.05
J
0.197 ± 0.008
0.268 ± 0.016
0.014 ± 0.004
0.037
0.034 MAX
0.063 ± 0.006
0.006 ± 0.004
0.02 ± 0.008
0.006 +0.004 -0.002
Figure 20. -S (DMP20 surface mount) package drawing
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA
Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com
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