TEXAS INSTRUMENTS TPA741 Technical data

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1 2 3 4

8 7 6 5

SHUTDOWN

BYPASS

IN+ IN–

VO– GND V

VO+

D OR DGN PACKAGE

(TOP VIEW)

Audio

Input

Bias

Control

700 mW

VO+

24BYPASS

IN–

VDD/2

SHUTDOWN

VO– 8

GND

From System Control

3 IN+

–

700-mW MONO LOW-VOLTAGE AUDIO POWER

AMPLIFIER WITH DIFFERENTIAL INPUTS

FEATURES DESCRIPTION

• Fully Specified for 3.3-V and 5-V Operation

• Wide Power Supply Compatibility 2.5 V - 5.5 V

• Output Power for R

– 700 mW at V – 250 mW at V

DD DD

• Integrated Depop Circuitry

• Thermal and Short-Circuit Protection

• Surface-Mount Packaging

– SOIC – PowerPAD™ MSOP

= 8 Ω

= 5 V = 3.3 V load at less than 0.6% THD+N throughout voice band

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

The TPA741 is a bridge-tied load (BTL) audio power amplifier developed especially for low-voltage applications where internal speakers are required. Operating with a 3.3-V supply, the TPA741 can deliver 250-mW of continuous power into a BTL 8-Ω

frequencies. Although this device is characterized out to 20 kHz, its operation is optimized for narrower band applications such as wireless communications. The BTL configuration eliminates the need for external coupling capacitors on the output in most applications, which is particularly important for small battery-powered equipment. This device features a shutdown mode for power-sensitive applications with a supply current of 7 µA during shutdown. The TPA741 is available in an 8-pin SOIC surface-mount package and the surface-mount PowerPAD™ MSOP, which reduces board space by 50% and height by 40%.

PowerPAD is a trademark of Texas Instruments.

PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters.

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

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TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates.

AVAILABLE OPTIONS

PACKAGED DEVICES

SMALL OUTLINE

(1)

(D) (DGN)

–40°C to 85°C TPA741D TPA741DGN AJD

(1) In the D package, the maximum output power is thermally limited to 350 mW; 700-mW peaks can be driven, as long as the RMS value

is less than 350 mW.

(2) The D and DGN packages are available taped and reeled. To order a taped and reeled part, add the suffix R to the part number (e.g.,

TPA741DR).

Terminal Functions

TERMINAL

NAME NO.

BYPASS 2 I GND 7 GND is the ground connection.

IN- 4 I IN- is the inverting input. IN- is typically used as the audio input terminal. IN+ 3 I IN+ is the noninverting input. IN+ is typically tied to the BYPASS terminal for SE operations. SHUTDOWN 1 I SHUTDOWN places the entire device in shutdown mode when held high. V

O+

O-

I/O DESCRIPTION

BYPASS is the tap to the voltage divider for internal mid-supply bias. This terminal should be connected to a 0.1-µF to 2.2-µF capacitor when used as an audio amplifier.

6 V 5 O V

is the supply voltage terminal.

is the positive BTL output.

O+

8 O VO-is the negative BTL output.

(2)

MSOP

MSOP SYMBOLIZATION

ABSOLUTE MAXIMUM RATINGS

over operating free-air temperature range (unless otherwise noted)

V V

T T T

(1) Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings

Supply voltage 6 v

Input voltage –0.3 V to V

Continuous total power dissipation Internally limited (see Dissipation Rating Table) Operating free-air temperature range –40°C to 85°C

Operating junction temperature range –40°C to 150°C

Storage temperature range –65°C to 150°C

stg

Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds 260°C

only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

(1)

UNIT

+0.3 V

DISSIPATION RATING TABLE

PACKAGE TA≤ 25°C DERATING FACTOR TA= 70°C TA= 85°C

D 725 mW 5.8 mW/°C 464 mW 377 mW

DGN 2.14 W

(1) See the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report

(SLMA002), for more information on the PowerPAD package. The thermal data was measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended Board for PowerPAD of that document.

(1)

17.1 mW/°C 1.37 W 1.11 W

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SLOS316C – JUNE 2000 – REVISED JUNE 2004

RECOMMENDED OPERATING CONDITIONS

MIN MAX UNIT

Supply voltage, 2.5 5.5 V High-level voltage (SHUTDOWN) 0.9V

Low-level voltage (SHUTDOWN) 0.1V Operating free-air temperature –40 85 °C

ELECTRICAL CHARACTERISTICS

at specified free-air temperature, V

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

| Output offset voltage (measured differentially) SHUTDOWN = 0 V, RL= 8 Ω, RF = 10 kΩ 20 mV

PSRR Power supply rejection ratio V I

I |IIH| SHUTDOWN, V

|IIL| SHUTDOWN, V

Supply current SHUTDOWN = 0 V, RF = 10 kΩ 1.35 2.5 mA

Supply current, shutdown mode

DD(SD)

(see Figure 6 )

= 3.3 V, TA= 25°C (unless otherwise noted)

= 3.2 V to 3.4 V 85 dB

SHUTDOWN = VDD, RF = 10 kΩ 7 50 µA

= 3.3 V, Vi= 3.3 V 1 µA

= 3.3 V, Vi= 0 V 1 µA

OPERATING CONDITIONS

= 3.3 V, TA= 25°C, RL= 8 Ω

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

Output power, See THD + N Total harmonic distortion plus noise PO= 250 mW, f = 200 Hz to 4 kHz, See Figure 7 0.55% B

SVR

Maximum output power bandwidth AV= -2 V/V, THD = 2%, See Figure 7 20 kHz

Unity-gain bandwidth Open loop, See Figure 15 1.4 MHz

Supply ripple rejection ratio f = 1 kHz, CB= 1 µF, See Figure 2 79 dB

Noise output voltage AV= -1 V/V, CB= 0.1 µF, See Figure 19 17 µV(rms)

(1) Output power is measured at the output terminals of the device at f = 1 kHz.

(1)

THD = 0.5%, See Figure 9 250 mW

TPA741

V V

ELECTRICAL CHARACTERISTICS

at specified free-air temperature, V

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

| Output offset voltage (measured differentially) SHUTDOWN = 0 V, RL= 8 Ω, RF = 10 kΩ 20 mV

PSRR Power supply rejection ratio V I

I |IIH| SHUTDOWN, V

|IIL| SHUTDOWN, V

Supply current SHUTDOWN = 0 V, RF = 10 kΩ 1.45 2.5 mA

Supply current, shutdown mode (see Fig-

DD(SD)

ure 4 )

= 5 V, TA= 25°C (unless otherwise noted)

= 4.9 V to 5.1 V 78 dB

SHUTDOWN = VDD, RF = 10 kΩ 50 100 µA

= 5.5 V, Vi= V

= 5.5 V, Vi= 0 V 1 µA

1 µA

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Audio

Input

Bias

Control

VO+

24BYPASS

IN–

VDD/2

SHUTDOWN

VO– 8

L = 8

Ω

GND

3 IN+

–

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

OPERATING CHARACTERISTICS

= 5 V, TA= 25°C, RL= 8Ω

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

THD + N Total harmonic distortion plus noise 0.5% B

SVR

Output power THD = 0.5%, See Figure 13 700

PO= 250 mW, f = 200 Hz to 4 kHz,

See Figure 11 Maximum output power bandwidth AV= -2 V/V, THD = 2%, See Figure 11 20 kHz Unity-gain bandwidth Open loop, See Figure 16 1.4 MHz Supply ripple rejection ratio f = 1 kHz, CB= 1 µF, See Figure 2 80 dB Noise output voltage AV= -1 V/V, CB= 0.1 µF, See Figure 20 17 µV(rms)

(1) The DGN package, properly mounted, can conduct 700-mW RMS power continuously. The D package can only conduct 350-mW RMS

power continuously with peaks to 700 mW.

PARAMETER MEASUREMENT INFORMATION

(1)

Figure 1. BTL Mode Test Circuit

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−50

−60

−80

−100 20 100 1k

−30

−20

f − Frequency − Hz

10k 20k

−10

−40

−70

−90

VDD = 5 V

VDD = 3.3 V

RL = 8 Ω CB = 1 µF

SVR

− Supply Ripple Rejection Ratio − dB

VDD − Supply Voltage − V

1.8

0.8

0.6

3 4

5.5

− Supply Current − mA

2.5 3.5 4.5

1.6

1.2

1.4

SHUTDOWN = 0 V RF = 10 kΩ

SVR

Supply ripple rejection ratio vs Frequency 2 Supply current vs Supply voltage 3, 4

Output power

THD+N Total harmonic distortion plus noise

Open-loop gain and phase vs Frequency 15, 16

Closed-loop gain and phase vs Frequency 17, 18 V P

Output noise voltage vs Frequency 19, 20

Power dissipation vs Output power 21, 22

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

TYPICAL CHARACTERISTICS

Table of Graphs

FIGURE

vs Supply voltage 5 vs Load resistance 6 vs Frequency 7, 8, 11, 12 vs Output power 9, 10, 13, 14

SUPPLY RIPPLE REJECTION RATIO SUPPLY CURRENT

vs vs

FREQUENCY SUPPLY VOLTAGE

Figure 2. Figure 3.

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VDD − Supply Voltage − V

20 10

3 43.5 4.5

SHUTDOWN = V

RF = 10 kΩ

5.52.5

− Supply Current − Aµ

VDD − Supply Voltage − V

600

400

200

2.5 3.53 4 5.5

1000

4.5 5

− Output Power − mW

800

THD+N 1% f = 1 kHz

RL = 32 Ω

RL = 8 Ω

RL − Load Resistance − Ω

300

200

100

16 3224 40 64

800

48 56

− Output Power − mW

400

THD+N = 1% f = 1 kHz

VDD = 5 V

500

600

VDD = 3.3 V

700

f − Frequency − Hz

THD+N −Total Harmonic Distortion + Noise − %

AV = −2 V/V

VDD = 3.3 V PO = 250 mW RL = 8 Ω

20 1k 10k

0.01

0.1

20k100

AV = −20 V/V

AV = −10 V/V

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

SUPPLY CURRENT OUTPUT POWER

vs vs

SUPPLY VOLTAGE SUPPLY VOLTAGE

Figure 4. Figure 5.

OUTPUT POWER TOTAL HARMONIC DISTORTION + NOISE

vs vs

LOAD RESISTANCE FREQUENCY

Figure 6. Figure 7.

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f − Frequency − Hz

THD+N −Total Harmonic Distortion + Noise − %

PO = 125 mW

VDD = 3.3 V RL = 8 Ω AV = −2 V/V

20 1k 10k

0.01

0.1

20k100

PO = 50 mW

PO = 250 mW

PO − Output Power − W

THD+N −Total Harmonic Distortion + Noise − %

0 0.15 0.4

0.01

0.1

0.2 0.25 0.3 0.35

VDD = 3.3 V f = 1 kHz AV = −2 V/V

0.05 0.1

RL = 8 Ω

PO − Output Power − W

THD+N −Total Harmonic Distortion + Noise − %

f = 20 kHz

VDD = 3.3 V RL = 8 Ω CB = 1 µF AV = −2 V/V

0.01 0.1 1

0.01

0.1

f = 1 kHz

f = 10 kHz

f = 20 Hz

f − Frequency − Hz

THD+N −Total Harmonic Distortion + Noise − %

AV =− 2 V/V

VDD = 5 V PO = 700 mW RL = 8 Ω

20 1k 10k

0.01

0.1

20k100

AV = −20 V/V

AV = −10 V/V

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISE

vs vs

FREQUENCY OUTPUT POWER

Figure 8. Figure 9.

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISE

vs vs

OUTPUT POWER FREQUENCY

Figure 10. Figure 11.

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f − Frequency − Hz

THD+N −Total Harmonic Distortion + Noise − %

PO = 700 mW

VDD = 5 V RL = 8 Ω AV = −2 V/V

20 1k 10k

0.01

0.1

20k100

PO = 50 mW

PO = 350 mW

PO − Output Power − W

0.1 0.2 10.4 0.5 0.7 0.8

THD+N −Total Harmonic Distortion + Noise − %

RL = 8 Ω

VDD = 5 V f = 1 kHz AV = −2 V/V

0.01

0.1

0.3 0.6 0.9

−20

−30

f − Frequency − kHz

−10

180°

−180°

Phase

60°

−60°

Open-Loop Gain − dB

Phase

50 40

140° 100°

20°

−20°

−100°

−140°

VDD = 3.3 V RL = Open

Gain

PO − Output Power − W

THD+N −Total Harmonic Distortion + Noise − %

f = 20 Hz

VDD = 5 V RL = 8 Ω CB = 1 µF AV = −2 V/V

0.01 0.1 1

0.01

0.1

f = 1 kHz

f = 10 kHz

f = 20 kHz

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISE

vs vs

FREQUENCY OUTPUT POWER

Figure 12. Figure 13.

TOTAL HARMONIC DISTORTION + NOISE OPEN-LOOP GAIN AND PHASE

vs vs

OUTPUT POWER FREQUENCY

Figure 14. Figure 15.

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−0.5

−1

−1.5

−2

f − Frequency − Hz

−0.25

−0.75

−1.25

−1.75

0.5

Closed-Loop Gain − dB

0.25

0.75

130°

120°

140°

Phase

150°

160°

VDD = 3.3 V RL = 8 Ω PO = 250 mW

170°

180°

Gain

Phase

−20

−30 1

f − Frequency − kHz

−10

Gain

Phase

Open-Loop Gain − dB

50 40

VDD = 5 V RL = Open

180°

−180°

60°

−60°

Phase

140° 100°

20°

−20°

−100°

−140°

−0.5

−1

−1.5

−2

f − Frequency − Hz

−0.25

−0.75

−1.25

−1.75

0.5

Closed-Loop Gain − dB

0.25

0.75

130°

120°

140°

Phase

150°

160°

VDD = 5 V RL = 8 Ω PO = 700 mW

170°

180°

Gain

Phase

− Output Noise Voltage − VµV

f − Frequency − Hz

20 1k 10k

100

20k100

VO BTL

VDD = 3.3 V BW = 22 Hz to 22 kHz RL = 8 Ω or 32 Ω AV = −1 V/V

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

OPEN-LOOP GAIN AND PHASE CLOSED-LOOP GAIN AND PHASE

vs vs

FREQUENCY FREQUENCY

Figure 16. Figure 17.

CLOSED-LOOP GAIN AND PHASE OUTPUT NOISE VOLTAGE

vs vs

FREQUENCY FREQUENCY

Figure 18. Figure 19.

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− Output Noise Voltage − VµV

f − Frequency − Hz

20 1k 10k

100

20k100

VDD = 5 V BW = 22 Hz to 22 kHz RL = 8 Ω or 32 Ω AV = −1 V/V

VO BTL

PD − Output Power − mW

6000

150

100

350

− Power Dissipation − mW

200

250

300

RL = 8 Ω

200 400

RL = 32 Ω

VDD = 3.3 V

PD − Output Power − mW

400 6000 1000

400

300

100

800

− Power Dissipation − mW

500

700

600

200

RL = 32 Ω

200 800

VDD = 5 V

RL = 8 Ω

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

OUTPUT NOISE VOLTAGE POWER DISSIPATION

vs vs

FREQUENCY OUTPUT POWER

Figure 20. Figure 21.

POWER DISSIPATION

OUTPUT POWER

Figure 22.

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Power 

(rms)



O(PP)

2 2



2x V

O(PP)

–V

O(PP)



2 RLC

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

APPLICATION INFORMATION

BRIDGE-TIED LOAD

Figure 23 shows a linear audio power amplifier (APA) in a BTL configuration. The TPA741 BTL amplifier consists of two linear amplifiers driving both ends of the load. There are several potential benefits to this differential drive configuration, but initially consider power to the load. The differential drive to the speaker means that as one side is slewing up, the other side is slewing down, and vice versa. This, in effect, doubles the voltage swing on the load as compared to a ground referenced load. Plugging 2 × V squared, yields 4× the output power from the same supply rail and load impedance (see Equation 1 ).

into the power equation, where voltage is

O(PP)

(1)

Figure 23. Bridge-Tied Load Configuration

In a typical portable handheld equipment sound channel operating at 3.3 V, bridging raises the power into an 8-Ω speaker from a singled-ended (SE, ground reference) limit of 62.5 mW to 250 mW. In sound power that is a 6-dB improvement, which is loudness that can be heard. In addition to increased power, there are frequency response concerns. Consider the single-supply, SE configuration shown in Figure 24 . A coupling capacitor is required to block the dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 µF to 1000 µF), so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting low-frequency performance of the system. This frequency-limiting effect is due to the high-pass filter network created with the speaker impedance and the coupling capacitance and is calculated with Equation 2 .

For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency performance is then limited only by the input network and speaker response. Cost and PCB space are also minimized by eliminating the bulky coupling capacitor.

(2)

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O(PP)

–3 dB

L(RMS)

DD(RMS)

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

APPLICATION INFORMATION (continued)

Figure 24. Single-Ended Configuration and Frequency Response

Increasing power to the load does carry a penalty of increased internal power dissipation. The increased dissipation is understandable considering that the BTL configuration produces 4× the output power of a SE configuration. Internal dissipation versus output power is discussed further in the thermal considerations section.

BTL AMPLIFIER EFFICIENCY

Linear amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across the output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc voltage drop that varies inversely to output power. The second component is due to the sine-wave nature of the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from V The internal voltage drop multiplied by the RMS value of the supply current, IDDrms, determines the internal power dissipation of the amplifier.

An easy-to-use equation to calculate efficiency starts out being equal to the ratio of power from the power supply to the power delivered to the load. To accurately calculate the RMS values of power in the load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 25 ).

Figure 25. Voltage and Current Waveforms for BTL Amplifiers

Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are different between SE and BTL configurations. In an SE application, the current waveform is a half-wave rectified shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different. Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform. The following equations are the basis for calculating amplifier efficiency.

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DD(RMS)



 R

SUP

 VDDI

DD(RMS)



VDD2V

 R

Efficiency 

SUP

where



L(RMS)



L(RMS)





Efficiency of a BTL configuration 

 V





2 PLR



12

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

APPLICATION INFORMATION (continued)

Table 1 employs Equation 4 to calculate efficiencies for three different output power levels. The efficiency of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased, resulting in a nearly flat internal power dissipation over the normal operating range. The internal dissipation at full output power is less than in the half power range. Calculating the efficiency for a specific system is the key to proper power supply design.

(3)

(4)

OUTPUT POWER EFFICIENCY PEAK VOLTAGE

(1) High-peak voltage values cause the THD to increase.

A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in the efficiency equation to utmost advantage when possible. In Equation 4 , V that as V

goes down, efficiency goes up.

Table 1. Efficiency vs Output Power in 3.3-V, 8-Ω, BTL Systems

(W) (%) (V)

0.125 33.6 1.41 0.26

0.25 47.6 2.00 0.29

0.375 58.3 2.45

(1)

is in the denominator. This indicates

INTERNAL

DISSIPATION

(W)

0.28

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Audio

Input

Bias

Control

700 mW

VO+

24BYPASS

IN–

VDD/2

2.2

SHUTDOWN

VO– 8

GND

From System Control

3 IN+

10 kΩ

50 kΩ

–

Audio Input–

Bias

Control

700 mW

VO+

24BYPASS

IN–

VDD/2

2.2

SHUTDOWN

VO– 8

GND

From System Control

3 IN+

10 kΩ

50 kΩ

–

10 kΩ

Audio Input+

50 kΩ

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

APPLICATION SCHEMATICS

Figure 26 is a schematic diagram of a typical handheld audio application circuit, configured for a gain of –10 V/V.

Figure 26. TPA741 Application Circuit

Figure 27 is a schematic diagram of a typical handheld audio application circuit, configured for a gain of –10 V/V with a differential input.

It is important to note that using the additional R to shift slightly, which could influence the THD+N performance of the amplifier. Although an additional external operational amplifier could be used to buffer BYPASS from RF, tests in the laboratory have shown that the THD+N performance is only minimally affected by operating in the fully differential mode as shown in Figure 27 . The following sections discuss the selection of the components used in Figure 26 and Figure 27 .

Figure 27. TPA741 Application Circuit With Differential Input

resistor connected between IN+ and BYPASS will cause V

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COMPONENT SELECTION

BTL gain   2





Effective impedance 

RFR

RF R

(7)



2 RFC

−3 dB

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

Gain-Setting Resistors, R

The gain for each audio input of the TPA741 is set by resistors R

and R

and RIaccording to Equation 5 for BTL mode.

BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the voltage swing across the load. Given that the TPA741 is an MOS amplifier, the input impedance is high; consequently, input leakage currents are not generally a concern, although noise in the circuit increases as the value of R

increases. In addition, a certain range of R

values is required for proper start-up operation of the

amplifier. Taken together, it is recommended that the effective impedance seen by the inverting node of the amplifier be set between 5 kΩ and 20 kΩ. The effective impedance is calculated in Equation 6 .

As an example, consider an input resistance of 10 kΩ and a feedback resistor of 50 kΩ. The BTL gain of the amplifier would be -10 V/V and the effective impedance at the inverting terminal would be 8.3 kΩ, which is well within the recommended range.

For high-performance applications, metal film resistors are recommended because they tend to have lower noise levels than carbon resistors. For values of R formed from R

and the inherent input capacitance of the MOS input structure. For this reason, a small

compensation capacitor of approximately 5 pF should be placed in parallel with R

above 50 kΩ, the amplifier tends to become unstable due to a pole

when R

is greater than 50

kΩ. This, in effect, creates a low-pass filter network with the cutoff frequency defined in Equation 7 .

(5)

(6)

For example, if R

Input Capacitor, C

In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the proper dc level for optimum operation. In this case, C determined in Equation 8 .

is 100 kΩ and C

is 5 pF, then fcis 318 kHz, which is well outside of the audio range.

and R

form a high-pass filter with the corner frequency

(7)

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

2 RIC

−3 dB



2 RIf



CB 250 k

Ω







RF R



TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

The value of CIis important to consider as it directly affects the bass (low-frequency) performance of the circuit. Consider the example where R Equation 8 is reconfigured as Equation 9 .

In this example, C

is 0.4 µF, so one would likely choose a value in the range of 0.47 µF to 1 µF. A further

consideration for this capacitor is the leakage path from the input source through the input network (R the feedback resistor (R

) to the load. This leakage current creates a dc offset voltage at the input to the amplifier

that reduces useful headroom, especially in high-gain applications. For this reason, a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications as the dc level there is held at V than the source dc level. It is important to confirm the capacitor polarity in the application.

is 10 kΩ and the specification calls for a flat bass response down to 40 Hz.

/2, which is likely higher

, CI) and

(8)

(9)

Power Supply Decoupling, C

The TPA741 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved by using two capacitors of different types that target different types of noise on the power supply leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor, typically 0.1 µF, placed as close as possible to the device V

lead, works best. For filtering lower

frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater, placed near the audio power amplifier is recommended.

Midrail Bypass Capacitor, C

The midrail bypass capacitor, CB, is the most critical capacitor and serves several important functions. During start-up or recovery from shutdown mode, C

determines the rate at which the amplifier starts up. The second

function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and THD + N. The capacitor is fed from a 250-kΩ source inside the amplifier. To keep the start-up pop as low as possible, the relationship shown in Equation 10 should be maintained. This ensures that the input capacitor is fully charged before the bypass capacitor is fully charged and the amplifier starts up.

As an example, consider a circuit where C

is 2.2 µF, CIis 0.47 µF, R

is 50 kΩ, and RIis 10 kΩ. Inserting these

values into the Equation 10 we get:

18.2 ≤ 35.5 which satisfies the rule. Bypass capacitor, CB, values of 0.1-µF to 2.2-µF ceramic or tantalum low-ESRcapacitors

are recommended for the best THD and noise performance.

(10)

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PdB 10Log

ref

 10Log

700 mW

1 W

 –1.5 dB

 10

PdB10

x P

ref

TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

USING LOW-ESR CAPACITORS

Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal) capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance, the more the real capacitor behaves like an ideal capacitor.

5-V Versus 3.3-V OPERATION

The TPA741 operates over a supply range of 2.5 V to 5.5 V. This data sheet provides full specifications for 5-V and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no special considerations for 3.3-V versus 5-V operation with respect to supply bypassing, gain setting, or stability. The most important consideration is that of output power. Each amplifier in TPA741 can produce a maximum voltage swing of V opposed to V

O(PP)

power into an 8-Ω load before distortion becomes significant. Operation from 3.3-V supplies, as can be shown from the efficiency formula in Equation 4 , consumes

approximately two-thirds the supply power of operation from 5-V supplies for a given output-power level.

HEADROOM AND THERMAL CONSIDERATIONS

Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions. A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion as compared with the average power output. From the TPA741 data sheet, one can see that when the TPA741 is operating from a 5-V supply into a 8-Ω speaker that 700-mW peaks are available. Converting watts to dB:

- 1 V. This means, for 3.3-V operation, clipping starts to occur when V

O(PP)

= 4 V for 5-V operation. The reduced voltage swing subsequently reduces maximum output

= 2.3 V, as

Subtracting the headroom restriction to obtain the average listening level without distortion yields:

1.5 dB – 15 dB = –16.5 (15-dB headroom)

1.5 dB – 12 dB = –13.5 (12-dB headroom)

1.5 dB – 9 dB = –10.5 (9-dB headroom)

1.5 dB – 6 dB = –7.5 (6-dB headroom)

1.5 dB – 3 dB = –4.5 (3-dB headroom)

Converting dB back into watts:

= 22 mW (15-dB headroom) = 44 mW (12-dB headroom) = 88 mW (9-dB headroom) = 175 mW (6-dB headroom) = 350 mW (3- dB headroom)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the amplifier system. Comparing the absolute worst case, which is 700 mW of continuous power output with 0 dB of headroom, against 12-dB and 15-dB applications drastically affects maximum ambient temperature ratings for the system. Using the power dissipation curves for a 5-V, 8-Ω system, the internal dissipation in the TPA741 and maximum ambient temperatures is shown in Table 2 .

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TPA741

SLOS316C – JUNE 2000 – REVISED JUNE 2004

Table 2. TPA741 Power Rating, 5-V, 8-Ω, BTL

D PACKAGE DGN PACKAGE

PEAK OUTPUT POWER

POWER DISSIPATION

(mW) (mW)

700 700 mW 675 34°C 110°C 700 350 mW (3 dB) 595 47°C 115°C 700 176 mW (6 dB) 475 68°C 122°C 700 88 mW (9 dB) 350 89°C 125°C 700 44 mW (12 dB) 225 111°C 125°C

AVERAGE

OUTPUT POWER

Table 2 shows that the TPA741 can be used to its full 700-mW rating without any heat sinking in still air up to 110°C and 34°C for the DGN package (MSOP) and D package (SOIC), respectively.

(SOIC) (MSOP)

MAXIMUM AMBIENT MAXIMUM AMBIENT

TEMPERATURE TEMPERATURE

(0° CFM) (0° CFM)

PACKAGE OPTION ADDENDUM

www.ti.com

7-May-2007

PACKAGING INFORMATION

Orderable Device Status

(1)

Package

Type

Package Drawing

Pins Package

Qty

Eco Plan

TPA741D ACTIVE SOIC D 8 75 Green (RoHS &

no Sb/Br)

TPA741DG4 ACTIVE SOIC D 8 75 Green (RoHS &

no Sb/Br)

TPA741DGN ACTIVE MSOP-

Power

DGN 8 80 Green (RoHS &

no Sb/Br)

PAD

TPA741DGNG4 ACTIVE MSOP-

Power

DGN 8 80 Green (RoHS &

no Sb/Br)

PAD

TPA741DGNR ACTIVE MSOP-

Power

DGN 8 2500 Green (RoHS &

no Sb/Br)

PAD

TPA741DGNRG4 ACTIVE MSOP-

Power

DGN 8 2500 Green (RoHS &

no Sb/Br)

PAD

TPA741DR ACTIVE SOIC D 8 2500 Green (RoHS &

no Sb/Br)

TPA741DRG4 ACTIVE SOIC D 8 2500 Green (RoHS &

no Sb/Br)

(1)

The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in

a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device.

(2)

Lead/Ball Finish MSL Peak Temp

CU NIPDAU Level-1-260C-UNLIM

(3)

(2)

Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check

http://www.ti.com/productcontent for the latest availability information and additional product content details.

TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements

for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)

(3)

MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder

temperature.

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Addendum-Page 1

PACKAGE MATERIALS INFORMATION

www.ti.com

TAPE AND REEL INFORMATION

11-Mar-2008

*All dimensions are nominal

Device Package

TPA741DGNR MSOP-

Power

TPA741DR SOIC D 8 2500 330.0 12.4 6.4 5.2 2.1 8.0 12.0 Q1

Type

PAD

Package Drawing

DGN 8 2500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1

Pins SPQ Reel

Diameter

(mm)

Reel

Width

W1 (mm)

A0 (mm) B0 (mm) K0 (mm) P1

(mm)W(mm)

Pin1

Quadrant

Pack Materials-Page 1

PACKAGE MATERIALS INFORMATION

www.ti.com

11-Mar-2008

*All dimensions are nominal

Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)

TPA741DGNR MSOP-PowerPAD DGN 8 2500 358.0 335.0 35.0

TPA741DR SOIC D 8 2500 346.0 346.0 29.0

Pack Materials-Page 2

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TEXAS INSTRUMENTS TPA741 Technical data

Specifications and Main Features

Frequently Asked Questions

User Manual

FEATURES DESCRIPTION

ABSOLUTE MAXIMUM RATINGS

DISSIPATION RATING TABLE

RECOMMENDED OPERATING CONDITIONS

ELECTRICAL CHARACTERISTICS

OPERATING CONDITIONS

ELECTRICAL CHARACTERISTICS

OPERATING CHARACTERISTICS

PARAMETER MEASUREMENT INFORMATION

TYPICAL CHARACTERISTICS

APPLICATION INFORMATION

BRIDGE-TIED LOAD

BTL AMPLIFIER EFFICIENCY

APPLICATION SCHEMATICS

COMPONENT SELECTION

USING LOW-ESR CAPACITORS

5-V Versus 3.3-V OPERATION

HEADROOM AND THERMAL CONSIDERATIONS