TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
1
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
D
Fully Specified for 3.3-V and 5-V Operation
D
Wide Power Supply Compatibility
2.5 V – 5.5 V
D
Output Power for RL = 8 Ω
– 700 mW at VDD = 5 V, BTL
– 250 mW at V
DD
= 3.3 V, BTL
D
Integrated Depop Circuitry
D
Thermal and Short-Circuit Protection
D
Surface-Mount Packaging
– SOIC
– PowerP AD MSOP
description
The TP A721 is a bridge-tied load (BTL) audio power amplifier developed especially for low-voltage applications
where internal speakers are required. Operating with a 3.3-V supply, the TPA721 can deliver 250-mW of
continuous power into a BTL 8-Ω load at less than 0.6% THD+N throughout voice band frequencies. Although
this device is characterized out to 20 kHz, its operation was optimized for narrower band applications such as
wireless communications. The BTL configuration eliminates the need for external coupling capacitors on the
output in most applications, which is particularly important for small battery-powered equipment. This device
features a shutdown mode for power-sensitive applications with a supply current of 7 µA during shutdown. The
TP A721 is available in an 8-pin SOIC surface-mount package and the surface-mount PowerP AD MSOP, which
reduces board space by 50% and height by 40%.
Audio
Input
Bias
Control
V
DD
700 mW
6
5
7
VO+
V
DD
1
24BYPASS
IN –
VDD/2
C
I
R
I
C
S
C
B
R
F
SHUTDOWN
VO–8
GND
From System Control
3 IN+
–
+
–
+
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
Copyright 2000, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
1
2
3
4
8
7
6
5
SHUTDOWN
BYPASS
IN+
IN–
V
O
–
GND
V
DD
VO+
D OR DGN PACKAGE
(TOP VIEW)
PowerPAD is a trademark of Texas Instruments.
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
2
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
AVAILABLE OPTIONS
PACKAGED DEVICES
T
A
SMALL OUTLINE
†
(D)
MSOP
‡
(DGN)
MSOP
Symbolization
–40°C to 85°C TPA721D TPA721DGN ABC
†
In the D package, the maximum output power is thermally limited to 350 mW; 700 mW peaks
can be driven, as long as the RMS value is less than 350 mW.
‡
The D and DGN packages are available taped and reeled. T o order a taped and reeled part, add
the suffix R to the part number (e.g., TP A301DR).
Terminal Functions
TERMINAL
NAME NO.
I/O
DESCRIPTION
BYPASS 2 I
BYPASS is the tap to the voltage divider for internal mid-supply bias. This terminal should be connected
to a 0.1-µF to 2.2-µF capacitor when used as an audio amplifier.
GND 7 GND is the ground connection.
IN– 4 I IN– is the inverting input. IN– is typically used as the audio input terminal.
IN+ 3 I IN+ is the noninverting input. IN+ is typically tied to the BYPASS terminal.
SHUTDOWN 1 I SHUTDOWN places the entire device in shutdown mode when held high (IDD < 7 µA).
V
DD
6 VDD is the supply voltage terminal.
VO+ 5 O VO+ is the positive BTL output.
VO– 8 O VO– is the negative BTL output.
absolute maximum ratings over operating free-air temperature range (unless otherwise noted)
§
Supply voltage, VDD 6 V. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Input voltage, VI –0.3 V to VDD +0.3 V. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Continuous total power dissipation internally limited (see Dissipation Rating Table). . . . . . . . . . . . . . . . . . . . .
Operating free-air temperature range, T
A
–40°C to 85°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Operating junction temperature range, TJ –40°C to 150°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Storage temperature range, T
stg
–65°C to 150°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds 260°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
§
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
TA ≤ 25°C DERATING FACTOR TA = 70°C TA = 85°C
D 725 mW 5.8 mW/°C 464 mW 377 mW
DGN 2.14 W
¶
17.1 mW/°C 1.37 W 1.11 W
¶
Please see the Texas Instruments document,
PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled
T exas Instruments Recommended
Board for PowerPAD
on page 33 of the before mentioned document.
recommended operating conditions
MIN MAX UNIT
Supply voltage, V
DD
2.5
5.5
V
Operating free-air temperature, T
A
–40
85
°C
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
3
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
electrical characteristics at specified free-air temperature, VDD = 3.3 V , TA = 25°C (unless otherwise
noted)
PARAMETER TEST CONDITIONS
MIN TYP MAX UNIT
V
OO
Output offset voltage (measured differentially)
See Note 1
20
mV
PSRR
Power supply rejection ratio
VDD = 3.2 V to 3.4 V
85
dB
I
DD
Supply current
BTL mode
1.25
2.5
mA
I
DD(SD)
Supply current, shutdown mode (see Figure 4)
7
50
µA
NOTE 1: At 3 V < VDD < 5 V the dc output voltage is approximately VDD/2.
operating characteristics, VDD = 3.3 V, T
A
= 25°C, R
L
= 8 Ω
PARAMETER TEST CONDITIONS
MIN TYP MAX UNIT
P
O
БББББББББ
Output power, see Note 2
THD = 0.5%,
See Figure 9
250
mW
THD + N
БББББББББ
Total harmonic distortion plus noise
PO = 250 mW,
f = 200 Hz to 4 kHz, See Figure 7
0.55%
B
OM
БББББББББ
Maximum output power bandwidth
Gain = 2,
THD = 2%, See Figure 7
20
kHz
B
1
БББББББББ
Unity-gain bandwidth
Open Loop,
See Figure 15
1.4
MHz
k
SVR
БББББББББ
Supply ripple rejection ratio
f = 1 kHz,
CB = 1 µF, See Figure 2
79
dB
V
n
БББББББББ
Noise output voltage
Gain = 1,
CB = 0.1 µF, See Figure 19
17
µV(rms)
NOTE 2: Output power is measured at the output terminals of the device at f = 1 kHz.
electrical characteristics at specified free-air temperature, VDD = 5 V , TA = 25°C (unless otherwise
noted)
PARAMETER TEST CONDITIONS
MIN TYP MAX UNIT
V
OO
Output offset voltage (measured differentially)
20
mV
PSRR
Power supply rejection ratio
VDD = 4.9 V to 5.1 V
78
dB
I
DD
Supply current
1.25
2.5
mA
I
DD(SD)
Supply current, shutdown mode (see Figure 4)
50
100
µA
operating characteristics, VDD = 5 V, T
A
= 25°C, RL = 8 Ω
PARAMETER TEST CONDITIONS
MIN TYP MAX UNIT
P
O
БББББББББ
Output power
THD = 0.5%,
See Figure 13
700
†
mW
THD + N
БББББББББ
Total harmonic distortion plus noise
PO = 250 mW,
f = 200 Hz to 4 kHz, See Figure 11
0.5%
B
OM
БББББББББ
Maximum output power bandwidth
Gain = 2,
THD = 2%, See Figure 11
20
kHz
B
1
БББББББББ
Unity-gain bandwidth
Open Loop,
See Figure 16
1.4
MHz
k
SVR
БББББББББ
Supply ripple rejection ratio
f = 1 kHz,
CB = 1 µF, See Figure 2
80
dB
V
n
БББББББББ
Noise output voltage
Gain = 1,
CB = 0.1 µF, See Figure 20
17
µV(rms)
†
The DGN package, properly mounted, can conduct 700 mW RMS power continuously. The D package can only conduct 350 mW RMS power
continuously wtih peaks to 700 mW.
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
4
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
PARAMETER MEASUREMENT INFORMATION
Audio
Input
Bias
Control
V
DD
6
5
7
VO+
V
DD
1
24BYPASS
IN –
VDD/2
C
I
R
I
C
S
C
B
R
F
SHUTDOWN
VO–8
RL = 8
Ω
GND
3 IN+
–
+
–
+
Figure 1. BTL Mode Test Circuit
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
k
SVR
Supply ripple rejection ratio vs Frequency 2
I
DD
Supply current vs Supply voltage 3, 4
p
p
vs Supply voltage 5
POOutput power
vs Load resistance 6
p
vs Frequency 7, 8, 11, 12
THD+N
Total harmonic distortion plus noise
vs Output power 9, 10, 13, 14
Open loop gain and phase vs Frequency 15, 16
Closed loop gain and phase vs Frequency 17, 18
V
n
Output noise voltage vs Frequency 19, 20
P
D
Power dissipation vs Output power 21, 22
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
5
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 2
–50
–60
–80
–100
20 100 1k
–30
–20
f – Frequency – Hz
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
0
10k 20k
–10
–40
–70
–90
VDD = 5 V
VDD = 3.3 V
RL = 8 Ω
CB = 1 µF
BTL
k
SVR
–Supply Ripple Rejection Ratio – dB
Figure 3
VDD – Supply Voltage – V
SUPPLY CURRENT
vs
SUPPLY VOLTAGE
1.8
0.8
0.6
1
34
5.5
5
I
DD
– Supply Current – mA
BTL
2.5 3.5 4.5
1.6
1.2
1.4
VDD – Supply Voltage – V
SUPPLY CURRENT
vs
SUPPLY VOLTAGE
20
10
0
343.5 4.5
60
5
30
SHUTDOWN = High
40
50
5.52.5
I
DD
– Supply Current – Aµ
70
80
90
Figure 4
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
6
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
VDD – Supply Voltage – V
OUTPUT POWER
vs
SUPPLY VOLTAGE
600
400
200
0
2.5 3.53 4 5.5
1000
P
4.5 5
O
– Output Power – mW
800
THD+N 1%
f = 1 kHz
BTL
RL = 32 Ω
RL = 8 Ω
Figure 5
RL – Load Resistance – Ω
OUTPUT POWER
vs
LOAD RESISTANCE
300
200
100
0
16 3224 40 64
800
8
P
48 56
O
– Output Power – mW
400
THD+N = 1%
f = 1 kHz
BTL
VDD = 5 V
500
600
VDD = 3.3 V
700
Figure 6
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
7
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 7
f – Frequency – Hz
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
AV = –2 V/V
VDD = 3.3 V
PO = 250 mW
RL = 8 Ω
BTL
20 1k 10k
1
0.01
10
0.1
20k100
AV = –20 V/V
AV = –10 V/V
Figure 8
f – Frequency – Hz
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
PO = 125 mW
VDD = 3.3 V
RL = 8 Ω
AV = –2 V/V
BTL
20 1k 10k
1
0.01
10
0.1
20k100
PO = 50 mW
PO = 250 mW
Figure 9
PO – Output Power – W
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
0 0.15 0.4
1
0.01
10
0.1
0.2 0.25 0.3 0.35
VDD = 3.3 V
f = 1 kHz
AV = –2 V/V
BTL
0.05 0.1
RL = 8 Ω
Figure 10
PO – Output Power – W
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
f = 20 kHz
VDD = 3.3 V
RL = 8 Ω
CB = 1 µF
AV = –2 V/V
BTL
0.01 0.1 1
1
0.01
10
0.1
f = 1 kHz
f = 10 kHz
f = 20 Hz
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
8
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 11
f – Frequency – Hz
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
AV =– 2 V/V
VDD = 5 V
PO = 700 mW
RL = 8 Ω
BTL
20 1k 10k
1
0.01
10
0.1
20k100
AV = –20 V/V
AV = –10 V/V
Figure 12
f – Frequency – Hz
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
PO = 700 mW
VDD = 5 V
RL = 8 Ω
AV = –2 V/V
BTL
20 1k 10k
1
0.01
10
0.1
20k100
PO = 50 mW
PO = 350 mW
Figure 13
PO – Output Power – W
0.1 0.2 10.4 0.5 0.7 0.8
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
RL = 8 Ω
VDD = 5 V
f = 1 kHz
AV = –2 V/V
BTL
1
0.01
10
0.1
0.3 0.6 0.9
Figure 14
PO – Output Power – W
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
f = 20 Hz
VDD = 5 V
RL = 8 Ω
CB = 1 µF
AV = –2 V/V
BTL
0.01 0.1 1
1
0.01
10
0.1
f = 1 kHz
f = 10 kHz
f = 20 kHz
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
9
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
10
0
–20
–30
20
30
f – Frequency – kHz
80
–10
180°
–180°
Phase
60°
–60°
OPEN-LOOP GAIN AND PHASE
vs
FREQUENCY
Open-Loop Gain – dB
Phase
1
10
1
10
2
10
3
10
4
50
40
60
70
140°
100°
20°
–20°
–100°
–140°
VDD = 3.3 V
RL = Open
BTL
Gain
Figure 15
10
0
–20
–30
1
20
30
f – Frequency – kHz
80
–10
Gain
Phase
OPEN-LOOP GAIN AND PHASE
vs
FREQUENCY
Open-Loop Gain – dB
10
1
10
2
10
3
10
4
50
40
60
70
VDD = 5 V
RL = Open
BTL
180°
–180°
60°
–60°
Phase
140°
100°
20°
–20°
–100°
–140°
Figure 16
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
10
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE
vs
FREQUENCY
–0.5
–1
–1.5
–2
f – Frequency – Hz
–0.25
–0.75
–1.25
–1.75
0
0.5
Closed-Loop Gain – dB
0.25
0.75
130°
120°
140°
Phase
150°
160°
VDD = 3.3 V
RL = 8 Ω
PO = 250 mW
BTL
1
170°
180°
Gain
Phase
10
1
10
2
10
3
10
4
10
5
10
6
Figure 17
CLOSED-LOOP GAIN AND PHASE
vs
FREQUENCY
–0.5
–1
–1.5
–2
f – Frequency – Hz
–0.25
–0.75
–1.25
–1.75
0
0.5
Closed-Loop Gain – dB
0.25
0.75
130°
120°
140°
Phase
150°
160°
VDD = 5 V
RL = 8 Ω
PO = 700 mW
BTL
1
170°
180°
Gain
Phase
10
1
10
2
10
3
10
4
10
5
10
6
Figure 18
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
11
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 19
– Output Noise Voltage – VµV
n
f – Frequency – Hz
OUTPUT NOISE VOLTAGE
vs
FREQUENCY
20 1k 10k
10
1
100
20k100
VO BTL
VDD = 3.3 V
BW = 22 Hz to 22 kHz
RL = 8 Ω or 32 Ω
AV = –1 V/V
V
o+
Figure 20
– Output Noise Voltage – VµV
n
f – Frequency – Hz
OUTPUT NOISE VOLTAGE
vs
FREQUENCY
20 1k 10k
10
1
100
20k100
VDD = 5 V
BW = 22 Hz to 22 kHz
RL = 8 Ω or 32 Ω
AV = –1 V/V
VO BTL
V
o+
Figure 21
PD – Output Power – mW
POWER DISSIPATION
vs
OUTPUT POWER
6000
150
100
50
0
350
P
D
– Power Dissipation – mW
200
250
300
RL = 8 Ω
200 400
RL = 32 Ω
BTL Mode
VDD = 3.3 V
Figure 22
PD – Output Power – mW
POWER DISSIPATION
vs
OUTPUT POWER
400 6000 1000
400
300
100
0
800
P
D
– Power Dissipation – mW
500
700
600
200
RL = 32 Ω
200 800
BTL Mode
VDD = 5 V
RL = 8 Ω
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
12
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
bridged-tied load
Figure 23 shows a linear audio power amplifier (AP A) in a BTL configuration. The TPA721 BTL amplifier consists
of two linear amplifiers driving both ends of the load. There are several potential benefits to this differential drive
configuration but initially consider power to the load. The differential drive to the speaker means that as one side
is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage swing on the
load as compared to a ground referenced load. Plugging 2 × V
O(PP)
into the power equation, where voltage is
squared, yields 4× the output power from the same supply rail and load impedance (see equation 1).
Power
+
V
(rms)
2
R
L
(1)
V
(rms)
+
V
O(PP)
22
Ǹ
R
L
2x V
O(PP)
V
O(PP)
–V
O(PP)
V
DD
V
DD
Figure 23. Bridge-Tied Load Configuration
In a typical portable handheld equipment sound channel operating at 3.3 V, bridging raises the power into an
8-Ω speaker from a singled-ended (SE, ground reference) limit of 62.5 mW to 250 mW. In sound power that is
a 6-dB improvement — which is loudness that can be heard. In addition to increased power, there are frequency
response concerns. Consider the single-supply SE configuration shown in Figure 24. A coupling capacitor is
required to block the dc offset voltage from reaching the load. These capacitors can be quite large
(approximately 33 µF to 1000 µF) so they tend to be expensive, heavy, occupy valuable PCB area, and have
the additional drawback of limiting low-frequency performance of the system. This frequency limiting effect is
due to the high pass filter network created with the speaker impedance and the coupling capacitance and is
calculated with equation 2.
f
(corner)
+
1
2pRLC
C
(2)
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
13
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
bridged-tied load (continued)
For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
R
L
C
C
V
O(PP)
V
O(PP)
V
DD
–3 dB
f
c
Figure 24. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4× the output power of a SE
configuration. Internal dissipation versus output power is discussed further in the
thermal considerations
section.
BTL amplifier efficiency
Linear amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across the
output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc
voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the
output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from V
DD
.
The internal voltage drop multiplied by the RMS value of the supply current, IDDrms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out being equal to the ratio of power from the power supply
to the power delivered to the load. To accurately calculate the RMS values of power in the load and in the
amplifier, the current and voltage waveform shapes must first be understood (see Figure 25).
V
(LRMS)
V
O
I
DD
I
DD(RMS)
Figure 25. Voltage and Current Waveforms for BTL Amplifiers
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
14
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
BTL amplifier efficiency (continued)
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
IDDrms
+
2V
P
p
R
L
P
SUP
+
VDDIDDrms
+
V
DD2VP
p
R
L
Efficiency
+
P
L
P
SUP
Efficiency of a BTL Configuration
+
p
V
P
2V
+
p
ǒ
PLR
L
2
Ǔ
1ń2
2V
(3)
Where:
(4)
PL+
VLrms
2
R
L
+
V
p
2
2R
L
VLrms
+
V
P
2
Ǹ
T able 1 employs equation 4 to calculate efficiencies for three dif ferent output power levels. The efficiency of the
amplifier is quite low for lower power levels and rises sharply as power to the load is increased, resulting in a
nearly flat internal power dissipation over the normal operating range. The internal dissipation at full output
power is less than in the half power range. Calculating the efficiency for a specific system is the key to proper
power supply design.
Table 1. Efficiency vs Output Power in 3.3-V 8-Ω BTL Systems
OUTPUT POWER
(W)
EFFICIENCY
(%)
PEAK-to-PEAK
VOLTAGE
(V)
INTERNAL
DISSIPATION
(W)
0.125 33.6 1.41 0.26
0.25 47.6 2.00 0.29
0.375 58.3 2.45
†
0.28
†
High-peak voltage values cause the THD to increase.
A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. In equation 4, VDD is in the denominator. This indicates
that as VDD goes down, efficiency goes up.
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
15
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
application schematic
Figure 26 is a schematic diagram of a typical handheld audio application circuit, configured for a gain of
–10 V/V.
Audio
Input
Bias
Control
V
DD
700 mW
6
5
7
VO+
V
DD
1
24BYPASS
IN –
VDD/2
C
I
C
S
1 µF
C
B
2.2 µF
SHUTDOWN
VO–8
GND
From System Control
3 IN+
R
I
10 kΩ
R
F
50 kΩ
–
+
–
+
Figure 26. TPA721 Application Circuit
The following sections discuss the selection of the components used in Figure 26.
component selection
gain setting resistors, RF and R
I
The gain for each audio input of the TP A721 is set by resistors RF and RI according to equation 5 for BTL mode.
(5)
BTL Gain
+*
2
ǒ
R
F
R
I
Ǔ
BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the
voltage swing across the load. Given that the TPA721 is a MOS amplifier, the input impedance is very high;
consequently input leakage currents are not generally a concern, although noise in the circuit increases as the
value of RF increases. In addition, a certain range of RF values is required for proper startup operation of the
amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the
amplifier be set between 5 kΩ and 20 kΩ. The effective impedance is calculated in equation 6.
(6)
Effective Impedance
+
RFR
I
RF)
R
I
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
16
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
gain setting resistors, RF and RI (continued)
As an example consider an input resistance of 10 kΩ and a feedback resistor of 50 kΩ. The BTL gain of the
amplifier would be –10 V/V and the effective impedance at the inverting terminal would be 8.3 kΩ, which is well
within the recommended range.
For high performance applications, metal film resistors are recommended because they tend to have lower
noise levels than carbon resistors. For values of RF above 50 kΩ, the amplifier tends to become unstable due
to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capacitor of approximately 5 pF should be placed in parallel with R
F
when RF is greater than
50 kΩ. This, in effect, creates a low pass filter network with the cutoff frequency defined in equation 7.
(7)
f
co(lowpass)
+
1
2pRFC
F
–3 dB
f
c
For example, if RF is 100 kΩ and CF is 5 pF, then fco is 318 kHz, which is well outside of the audio range.
input capacitor, C
I
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency
determined in equation 8.
(8)
f
co(highpass)
+
1
2pRIC
I
–3 dB
f
c
The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where RI is 10 kΩ and the specification calls for a flat bass response down to 40 Hz.
Equation 8 is reconfigured as equation 9.
(9)
CI+
1
2pRIf
co
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
17
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
input capacitor, CI (continued)
In this example, CI is 0.40 µF, so one would likely choose a value in the range of 0.47 µF to 1 µF. A further
consideration for this capacitor is the leakage path from the input source through the input network (RI, CI) and
the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier
that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at V
DD
/2, which is likely higher
than the source dc level. It is important to confirm the capacitor polarity in the application.
power supply decoupling, C
S
The TP A721 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents
oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved
by using two capacitors of different types that target different types of noise on the power supply leads. For
higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR)
ceramic capacitor, typically 0.1 µF placed as close as possible to the device V
DD
lead, works best. For filtering
lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near the audio
power amplifier is recommended.
midrail bypass capacitor, C
B
The midrail bypass capacitor, CB, is the most critical capacitor and serves several important functions. During
startup or recovery from shutdown mode, CB determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD + N. The capacitor is fed from a 250-kΩ source inside the amplifier. To keep the start-up pop as low as
possible, the relationship shown in equation 10 should be maintained. This insures the input capacitor is fully
charged before the bypass capacitor is fully charged and the amplifier starts up.
(10)
10
ǒ
CB
250 kΩ
Ǔ
v
1
ǒ
RF)
R
I
Ǔ
C
I
As an example, consider a circuit where CB is 2.2 µF, CI is 0.47 µF, RF is 50 kΩ, and RI is 10 kΩ. Inserting these
values into the equation 10 we get:
18.2v35.5
which satisfies the rule. Bypass capacitor, CB, values of 0.1 µF to 2.2 µF ceramic or tantalum low-ESR
capacitors are recommended for the best THD and noise performance.
using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance, the more the real capacitor behaves like an ideal capacitor.
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
18
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
5-V versus 3.3-V operation
The TP A721 operates over a supply range of 2.5 V to 5.5 V. This data sheet provides full specifications for 5-V
and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no
special considerations for 3.3-V versus 5-V operation with respect to supply bypassing, gain setting, or stability .
The most important consideration is that of output power. Each amplifier in TPA721 can produce a maximum
voltage swing of V
DD
– 1 V. This means, for 3.3-V operation, clipping starts to occur when V
O(PP)
= 2.3 V as
opposed to V
O(PP)
= 4 V at 5 V . The reduced voltage swing subsequently reduces maximum output power into
an 8-Ω load before distortion becomes significant.
Operation from 3.3-V supplies, as can be shown from the efficiency formula in equation 4, consumes
approximately two-thirds the supply power than operation from 5-V supplies for a given output-power level.
headroom and thermal considerations
Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions.
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TP A721 data sheet, one can see that when the TP A721
is operating from a 5-V supply into a 8-Ω speaker that 700 mW peaks are available. Converting watts to dB:
PdB+
10LogPW+
10Log 700 mW+–1.5 dB
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
–1.5 dB – 15 dB = –16.5 (15 dB headroom)
–1.5 dB – 12 dB = –13.5 (12 dB headroom)
–1.5 dB – 9 dB = –10.5 (9 dB headroom)
–1.5 dB – 6 dB = –7.5 (6 dB headroom)
–1.5 dB – 3 dB = –4.5 (3 dB headroom)
Converting dB back into watts:
PW+
10
PdBń10
+
22 mW (15 dB headroom)
+
44 mW (12 dB headroom)
+
88 mW (9 dB headroom)
+
175 mW (6 dB headroom)
+
350 mW (3 dB headroom)
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
19
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
headroom and thermal considerations (continued)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 700 mW of continuous power output with 0 dB
of headroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings
for the system. Using the power dissipation curves for a 5-V , 8-Ω system, the internal dissipation in the TP A721
and maximum ambient temperatures is shown in Table 2.
Table 2. TPA721 Power Rating, 5-V, 8-Ω, BTL
PEAK OUTPUT
POWER
D PACKAGE
(SOIC)
DGN PACKAGE
(MSOP)
POWER
(mW)
AVERAGE OUTPUT
POWER
DISSIPATION
(mW)
MAXIMUM AMBIENT
TEMPERATURE
(0 CFM)
MAXIMUM AMBIENT
TEMPERATURE
(0 CFM)
700 700 mW 675 34°C 110°C
700 350 mW (3 dB) 595 47°C 115°C
700 176 mW (6 dB) 475 68°C 122°C
700 88 mW (9 dB) 350 89°C 125°C
700 44 mW (12 dB) 225 111°C 125°C
Table 2 shows that the TPA721 can be used to its full 700-mW rating without any heat sinking in still air up to
110°C and 34°C for the DGN package (MSOP) and D pacakge (SOIC) respectively.
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
20
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
MECHANICAL DATA
D (R-PDSO-G**) PLASTIC SMALL-OUTLINE PACKAGE
14 PINS SHOWN
4040047/D 10/96
0.228 (5,80)
0.244 (6,20)
0.069 (1,75) MAX
0.010 (0,25)
0.004 (0,10)
1
14
0.014 (0,35)
0.020 (0,51)
A
0.157 (4,00)
0.150 (3,81)
7
8
0.044 (1,12)
0.016 (0,40)
Seating Plane
0.010 (0,25)
PINS **
0.008 (0,20) NOM
A MIN
A MAX
DIM
Gage Plane
0.189
(4,80)
(5,00)
0.197
8
(8,55)
(8,75)
0.337
14
0.344
(9,80)
16
0.394
(10,00)
0.386
0.004 (0,10)
M
0.010 (0,25)
0.050 (1,27)
0°–8°
NOTES: A. All linear dimensions are in inches (millimeters).
B. This drawing is subject to change without notice.
C. Body dimensions do not include mold flash or protrusion, not to exceed 0.006 (0,15).
D. Falls within JEDEC MS-012
TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B – NOVEMBER 1998 – REVISED MARCH 2000
21
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
MECHANICAL DATA
DGN (S-PDSO-G8) PowerPAD PLASTIC SMALL-OUTLINE PACKAGE
0,69
0,41
0,25
Thermal Pad
(See Note D)
0,15 NOM
Gage Plane
4073271/A 04/98
4,98
0,25
5
3,05
4,78
2,95
8
4
3,05
2,95
1
0,38
0,15
0,05
1,07 MAX
Seating Plane
0,10
0,65
M
0,25
0°–6°
NOTES: A. All linear dimensions are in millimeters.
B. This drawing is subject to change without notice.
C. Body dimensions include mold flash or protrusions.
D. The package thermal performance may be enhanced by attaching an external heat sink to the thermal pad.
This pad is electrically and thermally connected to the backside of the die and possibly selected leads.
E. Falls within JEDEC MO-187
PowerPAD is a trademark of Texas Instruments.
IMPORTANT NOTICE
T exas Instruments and its subsidiaries (TI) reserve the right to make changes to their products or to discontinue
any product or service without notice, and advise customers to obtain the latest version of relevant information
to verify, before placing orders, that information being relied on is current and complete. All products are sold
subject to the terms and conditions of sale supplied at the time of order acknowledgment, including those
pertaining to warranty, patent infringement, and limitation of liability.
TI warrants performance of its semiconductor products to the specifications applicable at the time of sale in
accordance with TI’s standard warranty. Testing and other quality control techniques are utilized to the extent
TI deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily
performed, except those mandated by government requirements.
Customers are responsible for their applications using TI components.
In order to minimize risks associated with the customer’s applications, adequate design and operating
safeguards must be provided by the customer to minimize inherent or procedural hazards.
TI assumes no liability for applications assistance or customer product design. TI does not warrant or represent
that any license, either express or implied, is granted under any patent right, copyright, mask work right, or other
intellectual property right of TI covering or relating to any combination, machine, or process in which such
semiconductor products or services might be or are used. TI’s publication of information regarding any third
party’s products or services does not constitute TI’s approval, warranty or endorsement thereof.
Copyright 2000, Texas Instruments Incorporated
Loading...