TEXAS INSTRUMENTS TPA2006D1 Technical data

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PWM H-

IN-

Bridge

O+

O-

Internal

Oscillator

ToBattery

GND

Bias

Circuitry

Differential

Input

TPA2006D1

SHUTDOWN

IN+

IN−

O−

GND

O+

8-PINQFN(DRB)PACKAGE

(TOP VIEW)

NC − Nointernalconnection

IN+

1.45-W MONO FILTER-FREE CLASS-D AUDIO POWER AMPLIFIER WITH 1.8-V COMPATIBLE INPUT THRESHOLDS

FEATURES APPLICATIONS

• Maximum Battery Life and Minimum Heat – Efficiency With an 8- Ω Speaker:

• 88% at 400 mW

• 80% at 100 mW

– 2.8-mA Quiescent Current – 0.5- µ A Shutdown Current

• Shutdown Pin has 1.8-V Compatible Thresholds

• Only Three External Components

• Optimized PWM Output Stage Eliminates

LC Output Filter

• Internally Generated 250-kHz Switching

Frequency Eliminates Capacitor and Resistor

• Improved PSRR (–75 dB) and Wide

Supply Voltage (2.5 V to 5.5 V) Eliminates Need for a Voltage Regulator

• Fully Differential Design Reduces RF

Rectification and Eliminates Bypass Capacitor

• Improved CMRR Eliminates Two Input

Coupling Capacitors

• Space Saving 3 mm x 3 mm QFN Package (DRB)

TPA2006D1

SLOS498 – SEPTEMBER 2006

• Ideal for Wireless or Cellular Handsets and

PDAs

DESCRIPTION

The TPA2006D1 is a 1.45-W high efficiency filter-free class-D audio power amplifier in a 3 mm × 3 mm QFN package that requires only three external components. The SHUTDOWN pin is fully compatible with 1.8-V logic GPIO, such as are used on low power cellular chipsets.

Features like 88% efficiency, –75-dB PSRR, improved RF-rectification immunity, and very small total PCB footprint make the TPA2006D1 ideal for cellular handsets. A fast start-up time of 1 ms with minimal pop makes the TPA2006D1 ideal for PDA applications.

In cellular handsets, the earpiece, speaker phone, and melody ringer can each be driven by the TPA2006D1. The TPA2006D1 allows independent gain while summing signals from separate sources, and has a low 36 µ V noise floor, A-weighted.

PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters.

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

APPLICATION CIRCUIT

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TPA2006D1

SLOS498 – SEPTEMBER 2006

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates.

ORDERING INFORMATION

PACKAGE

–40 ° C to 85 ° C 8-pin QFN (DRB) TPA2006D1DRB BTQ

(1) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI

Web site at www.ti.com .

ABSOLUTE MAXIMUM RATINGS

over operating free-air temperature range unless otherwise noted

Supply voltage

Input voltage –0.3 V to V

Continuous total power dissipation See Dissipation Rating Table

Operating free-air temperature –40 ° C to 85 ° C

Operating junction temperature –40 ° C to 125 ° C

Storage temperature –65 ° C to 150 ° C

stg

(1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings

only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

(1)

PART NUMBER SYMBOL

TPA2006D1

In active mode –0.3 V to 6 V In SHUTDOWN mode –0.3 V to 7 V

+ 0.3 V

RECOMMENDED OPERATING CONDITIONS

MIN NOM MAX UNIT

Supply voltage 2.5 5.5 V

High-level input voltage SHUTDOWN 1.3 V

Low-level input voltage SHUTDOWN 0 0.35 V

Input resistor Gain ≤ 20 V/V (26 dB) 15 k Ω

Common mode input voltage range V

Operating free-air temperature –40 85 ° C

= 2.5 V, 5.5 V, CMRR ≤ –49 dB 0.5 VDD–0.8 V

PACKAGE DISSIPATION RATINGS

PACKAGE DERATING FACTOR

(1)

DRB 21.8 mW/ ° C 2.7 W 1.7 W 1.4 W

(1) Derating factor measure with High K board.

TA≤ 25 ° C TA= 70 ° C TA= 85 ° C

POWER RATING POWER RATING POWER RATING

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V V

285 kW

300 kW

315 kW

TPA2006D1

SLOS498 – SEPTEMBER 2006

ELECTRICAL CHARACTERISTICS

TA= 25 ° C (unless otherwise noted)

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

|V PSRR Power supply rejection ratio V CMRR Common mode rejection ratio –68 –49 dB |IIH| High-level input current V

|IIL| Low-level input current V

(Q)

(SD)

DS(on)

(sw)

Output offset voltage

| VI= 0 V, AV= 2 V/V, V

(measured differentially)

= 2.5 V to 5.5 V –75 –55 dB

= 2.5 V to 5.5 V, VIC= VDD/2 to 0.5 V,

VIC= VDD/2 to V

= 5.5 V, VI= 5.8 V 100 µ A

= 5.5 V, VI= –0.3 V 5 µ A

= 5.5 V, no load 3.4 4.9

Quiescent current V

Shutdown current V

Static drain-source on-state resistance

Output impedance in SHUTDOWN V Switching frequency V

= 3.6 V, no load 2.8 mA

= 2.5 V, no load 2.2 3.2

DD ( SHUTDOWN)

= 2.5 V 770

= 3.6 V 590 m Ω

= 5.5 V 500

DD ( SHUTDOWN)

= 2.5 V to 5.5 V 200 250 300 kHz

= 2.5 V to 5.5 V 25 mV

–0.8 V

= 0.35 V, V

= 2.5 V to 5.5 V 0.5 2 µ A

= 0.35 V >1 k Ω

Gain V

Resistance from shutdown to GND 300 k Ω

OPERATING CHARACTERISTICS

TA= 25 ° C, Gain = 2 V/V, RL= 8 Ω (unless otherwise noted)

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

THD + N = 10%, f = 1 kHz, RL= 8 Ω V

THD+N V

SVR

SNR Signal-to-noise ratio V

CMRR Common mode rejection ratio V Z

Output power

THD + N = 1%, f = 1 kHz, RL= 8 Ω V

= 5 V, PO= 1 W, RL= 8 Ω , f = 1 kHz 0.19% Total harmonic distortion plus noise

Supply ripple rejection ratio V

Output voltage noise µ V

= 3.6 V, PO= 0.5 W, RL= 8 Ω , f = 1 kHz 0.19%

= 2.5 V, PO= 200 mW, RL= 8 Ω , f = 1 kHz 0.20%

= 3.6 V, Inputs ac-grounded

with Ci= 2 µ F

= 5 V, PO= 1 W, RL= 8 Ω , A-weighted 97 dB

= 3.6 V, f = 20 Hz to 20 kHz,

Inputs ac-grounded with Ci= 2 µ F

= 3.6 V, VIC= 1 V

Input impedance 142 150 158 k Ω Start-up time from shutdown V

= 3.6 V 1 ms

= 2.5 V to 5.5 V

= 5 V 1.45

= 3.6 V 0.73 W

= 2.5 V 0.33

= 5 V 1.19

= 3.6 V 0.59 W

= 2.5 V 0.26

f = 217 Hz,

= 200 –67 dB

(RIPPLE)

No weighting 48 A weighting 36

f = 217 Hz –63 dB

RMS

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150kW

300kW

150kW

TPA2006D1

SLOS498 – SEPTEMBER 2006

Terminal Functions

TERMINAL

NAME DRB

IN– 4 I Negative differential input IN+ 3 I Positive differential input V

O+

GND 7 O High-current ground V

O-

SHUTDOWN 1 I Shutdown terminal (active low logic) NC 2 - No Connect, not connected internal to the device. May be left unconnected Thermal Pad O Should be soldered to a grounded thermal pad on PCB for best thermal performance

6 I Power supply 5 O Positive BTL output

8 O Negative BTL output

FUNCTIONAL BLOCK DIAGRAM

I/O DESCRIPTION

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TPA2006D1

IN+

IN-

OUT+

OUT-

GND

Measurement

Output

1 Fm

Load

30-kHz

Low-Pass

Filter

Measurement

Input

SLOS498 – SEPTEMBER 2006

TYPICAL CHARACTERISTICS

TABLE OF GRAPHS

FIGURE

Efficiency vs Output power 1

(Q)

(SD)

THD+N Total harmonic distortion plus noise vs Frequency 10, 11, 12

CMRR Common-mode rejection ratio

Power dissipation vs Output power 2

Supply current vs Output power 3 Quiescent current vs Supply voltage 4 Shutdown current vs Shutdown voltage 5

Output power

vs Supply voltage 8 vs Load resistance 6, 7 vs Output power 9

vs Common-mode input voltage 13

Supply ripple rejection ratio vs Frequency 14, 15

SVR

GSM power supply rejection

Supply ripple rejection ratio vs Common-mode input voltage 18

SVR

vs Time 16 vs Frequency 17

vs Frequency 19 vs Common-mode input voltage 20

TPA2006D1

TEST SET-UP FOR GRAPHS

A. CIis shorted for any common-mode input voltage measurement. B. A 33- µ H inductor is placed in series with the load resistor to emulate a small speaker for efficiency measurements. C. The 30-kHz low-pass filter is required even if the analyzer has an internal low-pass filter. An RC low-pass filter

(100 Ω , 47 nF) is used on each output for the data sheet graphs.

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0 0.2 0.4 0.6 0.8 1 1.2

=2.5V,

R =8 ,33 HW m

Class-AB, V =5V,

R =8

P -OutputPower-W

Efficiency-%

=5V,

R =8 ,33 HW m

100

150

200

250

300

0 0.2 0.4 0.6 0.8 1 1.2

-OutputPower-W

V =2.5V,

R =8 ,33 H

W m

V =3.6V,

R =8 ,33 H

W m

V =5V,

R =8 ,33 H

W m

SupplyCurrent-mA

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0 0.2 0.4 0.6 0.8 1 1.2

PowerDissipation-W

PO-OutputPower-W

Class-AB, V =5V,R =8

DD L

Class-AB, V =3.6V,

R =8

V =3.6V,

R =8 ,33 H

W m

V =5V,

R =8 ,33 H

W m

0.5

1.5

0 0.1 0.2 0.3 0.4 0.5

Shutdown Voltage − V

− Shutdown Current −

(SD)

Aµ

VDD = 5 V

VDD = 3.6 V

VDD = 2.5 V

0.2

0.4

0.6

0.8

1.2

1.4

1.6

8 12 16 20 24 28 32

VDD=5V

V =3.6V

V =2.5V

R -LoadResistance-LW

P -OutputPower-W

f=1kHz THD+N=10% Gain=2V/V

2.2

2.4

2.6

2.8

3.2

3.4

3.6

3.8

2.5 3 3.5 4 4.5 5 5.5

(Q)

− QuiescentCurrent −

V − V

− SupplyVoltage

NoLoad

R =8 ,33 H

W m

2.5 3 3.5 4 4.5 5

V -SupplyVDDoltage-V

P -OutputPower-W

R =8 f=1kHz

Gain=2V/V

THD+N=1%

THD+N=10%

0.2

0.4

0.6

0.8

1.2

1.4

1.6

0.1

0.001 0.01 1k 10k

PowerOutput − W

THD+N − TotalHarmonicDistortion+Noise − %

0.1k

R =8 f=1kHz

2.5V

3.6V

5V

0.2

0.4

0.6

0.8

1.2

1.4

8 12 16 20 28

R -LoadResistance-LW

P -OutputPower-W

3224

f=1kHz THD+N=1% Gain=2V/V

V =2.5V

V =3.6V

VDD=5V

TPA2006D1

SLOS498 – SEPTEMBER 2006

EFFICIENCY POWER DISSIPATION SUPPLY CURRENT

vs vs vs

OUTPUT POWER OUTPUT POWER OUTPUT POWER

Figure 1. Figure 2. Figure 3.

QUIESCENT CURRENT SUPPLY CURRENT OUTPUT POWER

vs vs vs

SUPPLY VOLTAGE SHUTDOWN VOLTAGE LOAD RESISTANCE

Figure 4. Figure 5. Figure 6.

TOTAL HARMONIC DISTORTION +

OUTPUT POWER OUTPUT POWER NOISE

vs vs vs

LOAD RESISTANCE SUPPLY VOLTAGE OUTPUT POWER

Figure 7. Figure 8. Figure 9.

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0.01

0.1

20 100 10k 20k

f − Frequency − Hz

THD+N − TotalHarmonicDistortion+Noise − %

P =1W

P =0.25W

P =0.5W

V =5V

R =8LW

1k

0.001

0.1

20 100 10k 20k

f − Frequency − Hz

THD+N − TotalHarmonicDistortion+Noise − %

0.01

1k

V =3.6V

R =8LW

P =0.25W

P =0.5W

P =0.125W

0.001

0.1

20 100 10k 20k

f − Frequency − Hz

THD+N − TotalHarmonicDistortion+Noise − %

0.01

1k

V =2.5V

R =8LW

P =0.2W

P =0.075W

P =0.015W

−90

−80

−70

−60

−50

−40

−30

20 100 1k 10k

VDD=5V

VDD=2.5V

f − Frequency − Hz

SopplyRippleRejectionRatio − dB

Inputsfloating R =8LW

20k

VDD=3.6V

−90

−80

−70

−60

−50

−40

−30

20 100 1k 20k

f − Frequency − Hz

SupplyRippleRejectionRatio − dB

VDD=2.5V

VDD=3.6V

VDD=5V

Inputsac-grounded C

=2 Fm

=8 W

Gain=2V/V

10k

0.1

0 0.5 1 1.5 2 2.5

f = 1 kHz PO = 200 mW

− Common Mode Input Voltage − V

THD+N − Total Harmonic Distortion + Noise − %

3 3.5 4 4.5 5

VDD = 2.5 V

VDD = 5 V

VDD = 3.6 V

C1 − High

3.6 V

C1 − Amp 512 mV

C1 − Duty 12%

t − Time − 2 ms/div

200 mV/div

OUT

20 mV/div

−150

−100

−50

0 400 800 1200 1600 2000

−150

−100

−50

f − Frequency − Hz

− Output Voltage − dBVV O

− Supply Voltage − dBVV DD

VDD Shown in Figure 22 CI = 2 µF, Inputs ac-grounded Gain = 2V/V

TPA2006D1

SLOS498 – SEPTEMBER 2006

TOTAL HARMONIC DISTORTION + TOTAL HARMONIC DISTORTION + TOTAL HARMONIC DISTORTION +

NOISE NOISE NOISE

vs vs vs

FREQUENCY FREQUENCY FREQUENCY

Figure 10. Figure 11. Figure 12.

TOTAL HARMONIC DISTORTION +

NOISE SUPPLY RIPPLE REJECTION RATIO SUPPLY RIPPLE REJECTION RATIO

vs vs vs

COMMON MODE INPUT VOLTAGE FREQUENCY FREQUENCY

Figure 13. Figure 14. Figure 15.

GSM POWER SUPPLY REJECTION GSM POWER SUPPLY REJECTION

vs vs

TIME FREQUENCY

Figure 16. Figure 17.

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−100

−90

−80

−70

−60

−50

−40

−30

−20

−10

0 1 2 3 4 5

VIC − Common Mode Input Voltage − V

CMRR − Common Mode Rejection Ratio − dB

VDD = 5 V, Gain = 2

VDD = 2.5 V

VDD = 3.6 V

−75

−70

−65

−60

−55

−50

20 100 1 k 20 k

VDD = 3.6 V

f − Frequency − Hz

CMRR − Common Mode Rejection Ratio − dB

= 200 mV

RL = 8 Ω Gain = 2 V/V

10 k

−80

−70

−60

−50

−40

−30

−20

−10

0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5

DC Common Mode Voltage − V

Sopply Ripple Rejection Ratio − dB

VDD = 2. 5 V

VDD = 3.6 V

VDD = 5 V

TPA2006D1

SLOS498 – SEPTEMBER 2006

SUPPLY RIPPLE REJECTION RATIO COMMON-MODE REJECTION RATIO COMMON-MODE REJECTION RATIO

vs vs vs

DC COMMON MODE VOLTAGE FREQUENCY COMMON-MODE INPUT VOLTAGE

Figure 18. Figure 19. Figure 20.

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TPA2006D1

SLOS498 – SEPTEMBER 2006

APPLICATION INFORMATION

FULLY DIFFERENTIAL AMPLIFIER

The TPA2006D1 is a fully differential amplifier with differential inputs and outputs. The fully differential amplifier consists of a differential amplifier and a common-mode amplifier. The differential amplifier ensures that the amplifier outputs a differential voltage on the output that is equal to the differential input times the gain. The common-mode feedback ensures that the common-mode voltage at the output is biased around V regardless of the common-mode voltage at the input. The fully differential TPA2006D1 can still be used with a single-ended input; however, the TPA2006D1 should be used with differential inputs when in a noisy environment, like a wireless handset, to ensure maximum noise rejection.

Advantages of Fully Differential Amplifiers

• Input-coupling capacitors not required: – The fully differential amplifier allows the inputs to be biased at voltage other than mid-supply. For example,

if a codec has a mid-supply lower than the mid-supply of the TPA2006D1, the common-mode feedback circuit will adjust, and the TPA2006D1 outputs will still be biased at mid-supply of the TPA2006D1. The inputs of the TPA2006D1 can be biased from 0.5 V to V range, input-coupling capacitors are required.

• Mid-supply bypass capacitor, C

(BYPASS)

, not required:

– The fully differential amplifier does not require a bypass capacitor. This is because any shift in the

midsupply affects both positive and negative channels equally and cancels at the differential output.

• Better RF-immunity: – GSM handsets save power by turning on and shutting off the RF transmitter at a rate of 217 Hz. The

transmitted signal is picked-up on input and output traces. The fully differential amplifier cancels the signal much better than the typical audio amplifier.

– 0.8 V. If the inputs are biased outside of that

COMPONENT SELECTION

Figure 21 shows the TPA2006D1 typical schematic with differential inputs and Figure 22 shows the TPA2006D1

with differential inputs and input capacitors, and Figure 23 shows the TPA2006D1 with single-ended inputs. Differential inputs should be used whenever possible because the single-ended inputs are much more susceptible to noise.

Table 1. Typical Component Values

REF DES VALUE EIA SIZE MANUFACTURER PART NUMBER

S (1)

(1) CIis only needed for single-ended input or if V

(with RI= 150 k Ω ) gives a high-pass corner frequency of 321 Hz.

150 k Ω ( ± 0.5%) 0402 Panasonic ERJ2RHD154V

1 µ F (+22%, -80%) 0402 Murata GRP155F50J105Z

3.3 nF ( ± 10%) 0201 Murata GRP033B10J332K is not between 0.5 V and VDD– 0.8 V. CI= 3.3 nF

ICM

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Gain +

2 x 150 kW

V V

2p RIC

2p RIf

TPA2006D1

SLOS498 – SEPTEMBER 2006

Input Resistors (R

The input resistors (R

Resistor matching is very important in fully differential amplifiers. The balance of the output on the reference voltage depends on matched ratios of the resistors. CMRR, PSRR, and cancellation of the second harmonic distortion diminish if resistor mismatch occurs. Therefore, it is recommended to use 1% tolerance resistors or better to keep the performance optimized. Matching is more important than overall tolerance. Resistor arrays with 1% matching can be used with a tolerance greater than 1%.

Place the input resistors very close to the TPA2006D1 to limit noise injection on the high-impedance nodes. For optimal performance the gain should be set to 2 V/V or lower. Lower gain allows the TPA2006D1 to operate

at its best, and keeps a high voltage at the input making the inputs less susceptible to noise.

Decoupling Capacitor (C

The TPA2006D1 is a high-performance class-D audio amplifier that requires adequate power supply decoupling to ensure the efficiency is high and total harmonic distortion (THD) is low. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor, typically 1 µ F, placed as close as possible to the device V the TPA2006D1 is very important for the efficiency of the class-D amplifier, because any resistance or inductance in the trace between the device and the capacitor can cause a loss in efficiency. For filtering lower-frequency noise signals, a 10 µ F or greater capacitor placed near the audio power amplifier would also help, but it is not required in most applications because of the high PSRR of this device.

)

) set the gain of the amplifier according to Equation 1 .

)

lead works best. Placing this decoupling capacitor close to

(1)

Input Capacitors (C

The TPA2006D1 does not require input coupling capacitors if the design uses a differential source that is biased from 0.5 V to V common-mode input range, if needing to use the input as a high pass filter (shown in Figure 22 ), or if using a single-ended source (shown in Figure 23 ), input coupling capacitors are required.

The input capacitors and input resistors form a high-pass filter with the corner frequency, fc, determined in

Equation 2 .

The value of the input capacitor is important to consider as it directly affects the bass (low frequency) performance of the circuit. Speakers in wireless phones cannot usually respond well to low frequencies, so the corner frequency can be set to block low frequencies in this application.

Equation 3 is reconfigured to solve for the input coupling capacitance.

If the corner frequency is within the audio band, the capacitors should have a tolerance of ± 10% or better, because any mismatch in capacitance causes an impedance mismatch at the corner frequency and below.

For a flat low-frequency response, use large input coupling capacitors (1 µ F). However, in a GSM phone the ground signal is fluctuating at 217 Hz, but the signal from the codec does not have the same 217-Hz fluctuation. The difference between the two signals is amplified, sent to the speaker, and heard as a 217-Hz hum.

)

– 0.8 V (shown in Figure 21 ). If the input signal is not biased within the recommended

(2)

(3)

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IN-

IN+

PWM H-

Bridge

O+

O-

Internal

Oscillator

ToBattery

GND

Bias

Circuitry

Differential

Input

TPA2006D1

Filter-FreeClassD

SHUTDOWN

IN-

IN+

PWM H-

Bridge

O+

O-

Internal

Oscillator

ToBattery

GND

Bias

Circuitry

Differential

Input

TPA2006D1

Filter-FreeClassD

SHUTDOWN

TPA2006D1

Filter-FreeClassD

SHUTDOWN

SLOS498 – SEPTEMBER 2006

Figure 21. Typical TPA2006D1 Application Schematic With Differential Input for a Wireless Phone

TPA2006D1

Figure 22. TPA2006D1 Application Schematic With Differential Input and Input Capacitors

Figure 23. TPA2006D1 Application Schematic With Single-Ended Input

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Gain 1 +

2 x 150 kW

V V

Gain 2 +

2 x 150 kW

V V

Filter-FreeClassD

SHUTDOWN

TPA2006D1

SLOS498 – SEPTEMBER 2006

SUMMING INPUT SIGNALS WITH THE TPA2006D1

Most wireless phones or PDAs need to sum signals at the audio power amplifier or just have two signal sources that need separate gain. The TPA2006D1 makes it easy to sum signals or use separate signal sources with different gains. Many phones now use the same speaker for the earpiece and ringer, where the wireless phone would require a much lower gain for the phone earpiece than for the ringer. PDAs and phones that have stereo headphones require summing of the right and left channels to output the stereo signal to the mono speaker.

Summing Two Differential Input Signals

Two extra resistors are needed for summing differential signals (a total of 5 components). The gain for each input source can be set independently (see Equation 4 and Equation 5 , and Figure 24 ).

If summing left and right inputs with a gain of 1 V/V, use R

= R

= 300 k Ω .

(4)

(5)

Figure 24. Application Schematic With TPA2006D1 Summing Two Differential Inputs

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Gain 1 +

2 x 150 kW

V V

Gain 2 +

2 x 150 kW

V V

2p RI2f

2p 150kW 20Hz

C >53nF

_ +

IN-

IN+

PWM H-

Bridge

O+

O-

Internal

Oscillator

To Battery

GND

Bias

Circuitry

Differential

Input 1

Filter-Free Class D

SHUTDOWN

Single-Ended

Input 2

TPA2006D1

SLOS498 – SEPTEMBER 2006

Summing a Differential Input Signal and a Single-Ended Input Signal

Figure 25 shows how to sum a differential input signal and a single-ended input signal. Ground noise can couple

in through IN+ with this method. It is better to use differential inputs. The corner frequency of the single-ended input is set by C driven by a low-impedance source even if the input is not in use

If summing a ring tone and a phone signal, the phone signal should use a differential input signal while the ring tone might be limited to a single-ended signal.

The high pass corner frequency of the single-ended input is set by CI2. If the desired corner frequency is less than 20 Hz:

, shown in Equation 8 . To assure that each input is balanced, the single-ended input must be

(6)

(7)

(8)

(9)

(10)

Figure 25. Application Schematic With TPA2006D1 Summing Differential Input and Single-Ended Input

Signals

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Gain 1 +

2 x 150 kW

V V

Gain 2 +

2 x 150 kW

V V

2p RI1f

2p RI2f

CP+ CI1) C

RI1 R

RI1) R

_ +

IN-

IN+

PWM H-

Bridge

O+

O-

Internal

Oscillator

To Battery

GND

Bias

Circuitry

Filter-Free Class D

SHUTDOWN

Single-Ended

Input 2

Single-Ended

Input 1

TPA2006D1

SLOS498 – SEPTEMBER 2006

Summing Two Single-Ended Input Signals

Four resistors and three capacitors are needed for summing single-ended input signals. The gain and corner frequencies (f and Figure 26 ). Resistor, RP, and capacitor, CP, are needed on the IN+ terminal to match the impedance on the IN- terminal. The single-ended inputs must be driven by low impedance sources even if one of the inputs is not outputting an ac signal.

and fc2) for each input source can be set independently (see Equation 11 through Equation 14 ,

(11)

(12)

(13)

(14) (15)

Component Location

Place all the external components very close to the TPA2006D1. The input resistors need to be very close to the TPA2006D1 input pins so noise does not couple on the high impedance nodes between the input resistors and the input amplifier of the TPA2006D1. Placing the decoupling capacitor, CS, close to the TPA2006D1 is important for the efficiency of the class-D amplifier. Any resistance or inductance in the trace between the device and the capacitor can cause a loss in efficiency.

Figure 26. Application Schematic With TPA2006D1 Summing Two Single-Ended Inputs

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(16)

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Derating Factor

0.0218

=45.9 C/W

A J JA Dmax

Max= T Max P =125 45.9(0.2)=115.8 C- q -

0 V

-5 V

+5 V

Current

OUT+

Differential Voltage

Across Load

OUT-

TPA2006D1

SLOS498 – SEPTEMBER 2006

EFFICIENCY AND THERMAL INFORMATION

The maximum ambient temperature depends on the heat-sinking ability of the PCB system. The derating factor for the DRB package is shown in the dissipation rating table. Converting this to θJA:

Given θ dissipation of 0.2 W (Po=1.45 W, 8- Ω load, 5-V supply, from Figure 2 ), the maximum ambient temperature can be calculated with the following equation.

Equation 18 shows that the calculated maximum ambient temperature is 115.8 ° C at maximum power dissipation

with a 5-V supply and 8- Ω a load, see Figure 2 . The TPA2006D1 is designed with thermal protection that turns the device off when the junction temperature surpasses 150 ° C to prevent damage to the IC.

ELIMINATING THE OUTPUT FILTER WITH THE TPA2006D1

This section focuses on why the user can eliminate the output filter with the TPA2006D1.

Effect on Audio

The class-D amplifier outputs a pulse-width modulated (PWM) square wave, which is the sum of the switching waveform and the amplified input audio signal. The human ear acts as a band-pass filter such that only the frequencies between approximately 20 Hz and 20 kHz are passed. The switching frequency components are much greater than 20 kHz, so the only signal heard is the amplified input audio signal.

of 45.9 ° C/W, the maximum allowable junction temperature of 125 ° C, and the maximum internal

(17)

(18)

Traditional Class-D Modulation Scheme

The traditional class-D modulation scheme, which is used in the TPA005Dxx family, has a differential output where each output is 180 degrees out of phase and changes from ground to the supply voltage, V the differential pre-filtered output varies between positive and negative V

, where filtered 50% duty cycle yields

. Therefore,

0 volts across the load. The traditional class-D modulation scheme with voltage and current waveforms is shown in Figure 27 . Note that even at an average of 0 volts across the load (50% duty cycle), the current to the load is high causing a high loss and thus causing a high supply current.

Figure 27. Traditional Class-D Modulation Scheme's Output Voltage and Current Waveforms Into an

Inductive Load With no Input

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0 V

-5 V

+5 V

Current

OUT+

OUT-

Voltage

Across

Load

0 V

-5 V

+5 V

Current

OUT+

OUT-

Voltage

Across

Load

Output = 0 V

Output > 0 V

TPA2006D1

SLOS498 – SEPTEMBER 2006

TPA2006D1 Modulation Scheme

The TPA2006D1 uses a modulation scheme that still has each output switching from 0 to the supply voltage. However, OUT+ and OUT- are now in phase with each other with no input. The duty cycle of OUT+ is greater than 50% and OUT- is less than 50% for positive voltages. The duty cycle of OUT+ is less than 50% and OUTis greater than 50% for negative voltages. The voltage across the load sits at 0 volts throughout most of the switching period greatly reducing the switching current, which reduces any I2R losses in the load.

Figure 28. The TPA2006D1 Output Voltage and Current Waveforms Into an Inductive Load

Efficiency: Why You Must Use a Filter With the Traditional Class-D Modulation Scheme

The main reason that the traditional class-D amplifier needs an output filter is that the switching waveform results in maximum current flow. This causes more loss in the load, which causes lower efficiency. The ripple current is large for the traditional modulation scheme because the ripple current is proportional to voltage multiplied by the time at that voltage. The differential voltage swing is 2 × V half the period for the traditional modulation scheme. An ideal LC filter is needed to store the ripple current from each half cycle for the next half cycle, while any resistance causes power dissipation. The speaker is both resistive and reactive, whereas an LC filter is almost purely reactive.

The TPA2006D1 modulation scheme has very little loss in the load without a filter because the pulses are very short and the change in voltage is V making the ripple current larger. Ripple current could be filtered with an LC filter for increased efficiency, but for

instead of 2 × V

. As the output power increases, the pulses widen

and the time at each voltage is

most applications the filter is not needed. An LC filter with a cutoff frequency less than the class-D switching frequency allows the switching current to flow

through the filter instead of the load. The filter has less resistance than the speaker that results in less power dissipated, which increases efficiency.

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SPKR

+ P

SUP–PSUP THEORETICAL

(at max output power)

SPKR

SUP

OUT

–

SUP THEORETICAL

OUT

(at max output power)

SPKR

+ P

OUT

MEASURED

THEORETICAL

(at max output power)

hTHEORETICAL +

RL) 2r

DS(on)

(at max output power)

TPA2006D1

SLOS498 – SEPTEMBER 2006

Effects of Applying a Square Wave Into a Speaker

If the amplitude of a square wave is high enough and the frequency of the square wave is within the bandwidth of the speaker, a square wave could cause the voice coil to jump out of the air gap and/or scar the voice coil. A 250-kHz switching frequency, however, is not significant because the speaker cone movement is proportional to

1/f

for frequencies beyond the audio band. Therefore, the amount of cone movement at the switching frequency is very small. However, damage could occur to the speaker if the voice coil is not designed to handle the additional power. To size the speaker for added power, the ripple current dissipated in the load needs to be calculated by subtracting the theoretical supplied power, P maximum output power, P efficiency, η

MEASURED

, minus the theoretical efficiency, η

. The switching power dissipated in the speaker is the inverse of the measured

OUT

SUP THEORETICAL

THEORETICAL

The maximum efficiency of the TPA2006D1 with a 3.6 V supply and an 8- Ω load is 86% from Equation 22 . Using equation Equation 21 with the efficiency at maximum power (84%), we see that there is an additional 17 mW dissipated in the speaker. The added power dissipated in the speaker is not an issue as long as it is taken into account when choosing the speaker.

, from the actual supply power, P

SUP

, at

(19)

(20)

(21)

(22)

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1nF

Ferrite

ChipBead

O-

Ferrite

ChipBead

1nF

O+

0.1 Fm

33 Hm

O-

O+

0.1 Fm

0.47 Fm

TPA2006D1

SLOS498 – SEPTEMBER 2006

When to Use an Output Filter

Design the TPA2006D1 without an output filter if the traces from amplifier to speaker are short. The TPA2006D1 passed FCC and CE radiated emissions with no shielding with speaker trace wires 100 mm long or less. Wireless handsets and PDAs are great applications for class-D without a filter.

A ferrite bead filter can often be used if the design is failing radiated emissions without an LC filter, and the frequency sensitive circuit is greater than 1 MHz. This is good for circuits that just have to pass FCC and CE because FCC and CE only test radiated emissions greater than 30 MHz. If choosing a ferrite bead, choose one with high impedance at high frequencies, but very low impedance at low frequencies.

Use an LC output filter if there are low frequency (< 1 MHz) EMI sensitive circuits and/or there are long leads from amplifier to speaker.

Figure 29 and Figure 30 show typical ferrite bead and LC output filters.

Figure 29. Typical Ferrite Chip Bead Filter (Chip bead example: NEC/Tokin: N2012ZPS121)

Figure 30. Typical LC Output Filter, Cutoff Frequency of 27 kHz

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PACKAGE OPTION ADDENDUM

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2-Oct-2006

PACKAGING INFORMATION

Orderable Device Status

(1)

Package

Type

Package Drawing

Pins Package

Qty

Eco Plan

TPA2006D1DRBR ACTIVE SON DRB 8 3000 Green (RoHS &

no Sb/Br)

TPA2006D1DRBRG4 ACTIVE SON DRB 8 3000 Green (RoHS &

no Sb/Br)

TPA2006D1DRBT ACTIVE SON DRB 8 250 Green (RoHS &

no Sb/Br)

TPA2006D1DRBTG4 ACTIVE SON DRB 8 250 Green (RoHS &

no Sb/Br)

(1)

The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in

a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device.

(2)

Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check

http://www.ti.com/productcontent for the latest availability information and additional product content details.

TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements

for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)

(2)

Lead/Ball Finish MSL Peak Temp

CU NIPDAU Level-2-260C-1 YEAR

(3)

MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder

temperature.

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In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

Addendum-Page 1

PACKAGE MATERIALS INFORMATION

www.ti.com

TAPE AND REEL INFORMATION

11-Mar-2008

*All dimensions are nominal

Device Package

TPA2006D1DRBR SON DRB 8 3000 330.0 12.4 3.3 3.3 1.1 8.0 12.0 Q2

TPA2006D1DRBT SON DRB 8 250 180.0 12.4 3.3 3.3 1.1 8.0 12.0 Q2

Type

Package Drawing

Pins SPQ Reel

Diameter

(mm)

Reel

Width

W1 (mm)

A0 (mm) B0 (mm) K0 (mm) P1

(mm)W(mm)

Pin1

Quadrant

Pack Materials-Page 1

PACKAGE MATERIALS INFORMATION

www.ti.com

11-Mar-2008

*All dimensions are nominal

Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)

TPA2006D1DRBR SON DRB 8 3000 346.0 346.0 29.0 TPA2006D1DRBT SON DRB 8 250 190.5 212.7 31.8

Pack Materials-Page 2

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TEXAS INSTRUMENTS TPA2006D1 Technical data

Specifications and Main Features

Frequently Asked Questions

User Manual

FEATURES APPLICATIONS

DESCRIPTION

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

APPLICATION CIRCUIT

ORDERING INFORMATION

ABSOLUTE MAXIMUM RATINGS

RECOMMENDED OPERATING CONDITIONS

PACKAGE DISSIPATION RATINGS

ELECTRICAL CHARACTERISTICS

OPERATING CHARACTERISTICS

FUNCTIONAL BLOCK DIAGRAM

TYPICAL CHARACTERISTICS

TABLE OF GRAPHS

TEST SET-UP FOR GRAPHS

APPLICATION INFORMATION

FULLY DIFFERENTIAL AMPLIFIER

Advantages of Fully Differential Amplifiers

COMPONENT SELECTION

SUMMING INPUT SIGNALS WITH THE TPA2006D1

Summing Two Differential Input Signals

Summing a Differential Input Signal and a Single-Ended Input Signal

Summing Two Single-Ended Input Signals

Component Location

EFFICIENCY AND THERMAL INFORMATION

ELIMINATING THE OUTPUT FILTER WITH THE TPA2006D1

Effect on Audio

Traditional Class-D Modulation Scheme

TPA2006D1 Modulation Scheme

Efficiency: Why You Must Use a Filter With the Traditional Class-D Modulation Scheme

Effects of Applying a Square Wave Into a Speaker

When to Use an Output Filter