TEXAS INSTRUMENTS TPA112 Technical data

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1 2 3 4

8 7 6 5

VO1 IN1– IN1+

GND

VO2 IN2– IN2+

D OR DGN PACKAGE

(TOP VIEW)

Short-Circuit

Protection

Over-Temperature

Protection

IN1− IN1+

IN2− IN2+

VO1

VO2

DD/2

LIN−

LIN+

RIN−

RIN+

To Headphone Jack (See TPA152)

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

150-mW STEREO AUDIO POWER AMPLIFIER

FEATURES DESCRIPTION

• 150-mW Stereo Output

• Wide Range of Supply Voltages

– Fully Specified for 3.3-V and 5-V Operation – Operational From 2.5 V to 5.5 V

• Thermal and Short-Circuit Protection input channel and does not require external compen-

• Surface-Mount Packaging

– PowerPAD™ MSOP – SOIC

• Standard Operational Amplifier Pinout

The TPA112 is a stereo audio power amplifier packaged in an 8-pin PowerPAD™ MSOP package capable of delivering 150 mW of continuous RMS power per channel into 8-Ω loads. Amplifier gain is externally configured by means of two resistors per

sation for settings of 1 to 10. THD+N when driving an 8-Ω load from 5 V is 0.1% at

1 kHz, and less than 2% across the audio band of 20 Hz to 20 kHz. For 32-Ω loads, the THD+N is reduced to less than 0.06% at 1 kHz, and is less than 1% across the audio band of 20 Hz to 20 kHz. For 10-kΩ loads, the THD+N performance is 0.01% at 1 kHz, and less than 0.02% across the audio band of 20 Hz to 20 kHz.

TPA112

FUNCTIONAL BLOCK DIAGRAM

PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters.

PowerPAD is a trademark of Texas Instruments.

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

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TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates.

AVAILABLE OPTIONS

PACKAGED DEVICES

SMALL OUTLINE

(D) (DGN)

–40°C to 85°C TPA112D TPA112DGN TI AAD

(1) The D and DGN packages are available in left-ended tape and reel only (e.g., TPA112DR,

TPA112DGNR).

TERMINAL

NAME NO.

I/O DESCRIPTION

GND 4 I GND is the ground connection. IN1- 2 I IN1- is the inverting input for channel 1. IN1+ 3 I IN1+ is the noninverting input for channel 1. IN2- 6 I IN2- is the inverting input for channel 2. IN2+ 5 I IN2+ is the noninverting input for channel 2. V

8 I V

is the supply voltage terminal.

VO1 1 O VO1 is the audio output for channel 1. VO2 7 O VO2 is the audio output for channel 2.

(1)

Terminal Functions

(1)

MSOP

SYMBOLIZATION

ABSOLUTE MAXIMUM RATINGS

over operating free-air temperature range (unless otherwise noted)

V V I

T T

(1) Stresses beyond those listed under "absolute maximum ratings” may cause permanent damage to the device. These are stress ratings

Supply voltage 6 V

Differential input voltage –0.3 V to V

Input current ±2.5 µA Output current ±250 mA Continuous total power dissipation Internally llimited Operating junction temperature range –40°C to 150°C

Storage temperature range –65°C to 150°C

stg

Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds 260°C

only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

(1)

UNIT

DISSIPATION RATING TABLE

PACKAGE

D 725 mW 5.8 mW/°C 464 mW 377 mW

DGN 2.14 W

(1) See the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report

(SLMA002), for more information on the PowerPAD package. The thermal data was measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended Board for PowerPAD, of that document.

TA≤ 25°C DERATING FACTOR TA= 70°C TA= 85°C

POWER RATING ABOVE TA= 25°C POWER RATING POWER RATING

(1)

17.1 mW/°C 1.37 W 1.11 W

+ 0.3 V

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SLOS212E – AUGUST 1998 – REVISED JUNE 2004

RECOMMENDED OPERATING CONDITIONS

MIN MAX UNIT

V T

Supply voltage 2.5 5.5 V

Operating free-air temperature –40 85 °C

DC ELECTRICAL CHARACTERISTICS

at TA= 25°C, V

PSRR Power supply rejection ratio V I

DD(q)

Output offset voltage 10 mV

Supply current 1.5 3 mA Input impedance > 1 MΩ

= 3.3 V

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

= 3.2 V to 3.4 V 83 dB

AC OPERATING CHARACTERISTICS

= 3.3 V, TA= 25°C, RL= 8 Ω

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

Output power (each channel) THD ≤ 0.1% 70 THD+N Total harmonic distortion + noise PO= 70 mW, 20 Hz–20 kHz 2% B

Maximum output power BW G = 10, THD < 5% > 20 kHz

Phase margin Open loop 58° S

VRR

Supply ripple rejection f = 1 kHz 68 dB

Channel/channel output separation f = 1 kHz 86 dB SNR Signal-to-noise ratio PO= 100 mW 100 dB V

Noise output voltage 9.5 µV(rms)

(1) Measured at 1 kHz

(1)

TPA112

DC ELECTRICAL CHARACTERISTICS

at TA= 25°C, V

V PSRR Power supply rejection ratio V I Z

Output offset voltage 10 mV

Supply current 1.5 3 mA

DD(q)

Input impedance > 1 MΩ

= 5 V

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

= 4.9 V to 5.1 V 76 dB

AC OPERATING CHARACTERISTICS

= 5 V, TA= 25°C, RL= 8 Ω

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

Output power (each channel) THD ≤ 0.1% 70

THD+N Total harmonic distortion + noise PO= 150 mW, 20 Hz–20 kHz 2% B

Maximum output power BW G = 10, THD < 5% > 20 kHz

Phase margin Open loop 56°

Supply ripple rejection f = 1 kHz 68 dB

VRR

Channel/channel output separation f = 1 kHz 86 dB SNR Signal-to-noise ratio PO= 150 mW 100 dB V

Noise output voltage 9.5 µV(rms)

(1) Measured at 1 kHz

(1)

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TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

AC OPERATING CHARACTERISTICS

= 3.3 V, TA= 25°C, RL= 32 Ω

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

Output power (each channel) THD ≤ 0.1% 40 THD+N Total harmonic distortion + noise PO= 30 mW, 20 Hz–20 kHz 0.5% B

Maximum output power BW G = 10, THD < 2% > 20 kHz

Phase margin Open loop 58° S

VRR

Supply ripple rejection f = 1 kHz 68 dB

Channel/channel output separation f = 1 kHz 86 dB SNR Signal-to-noise ratio PO= 100 mW 100 dB V

Noise output voltage 9.5 µV(rms)

(1) Measured at 1 kHz

AC OPERATING CHARACTERISTICS

= 5 V, TA= 25°C, RL= 32 Ω

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

P THD+N Total harmonic distortion + noise PO= 60 mW, 20 Hz–20 kHz 0.4% B

SNR Signal-to-noise ratio PO= 150 mW 100 dB V

(1) Measured at 1 kHz

Output power (each channel) THD ≤ 0.1% 40

Maximum output power BW G = 10, THD < 2% > 20 kHz

Phase margin Open loop 56° Supply ripple rejection f = 1 kHz 68 dB

VRR

Channel/channel output separation f = 1 kHz 86 dB

Noise output voltage 9.5 µV(rms)

(1)

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0.1

0.01

0.001

20 100 1k 10k 20k

AV = 10

AV = 5

THD+N −Total Harmonic Distortion + Noise − %

f − Frequency − Hz

AV = 1

VDD = 3.3 V PO = 30 mW CB = 1 µ F RL = 32 Ω

0.1

0.01

0.001

20 100 1k 10k 20k

VDD = 3.3 V AV = 1 V/V RL = 32 Ω CB = 1 µ F

PO = 10 mW

THD+N −Total Harmonic Distortion + Noise − %

f − Frequency − Hz

PO = 15 mW

PO = 30 mW

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

TYPICAL CHARACTERISTICS

Table of Graphs

THD+N Total harmonic distortion plus noise vs Frequency

vs Output power 3, 6, 9, 12, 15, 18 PSSR Power supply rejection ratio vs Frequency 19, 20 V

SNR Signal-to-noise ratio vs Voltage gain 35

Output noise voltage vs Frequency 21, 22

Crosstalk vs Frequency 23-26, 37, 38 Mute attenuation vs Frequency 27, 28 Open-loop gain vs Frequency 29, 30 Phase margin vs Frequency 29, 30 Phase vs Frequency 39-44 Output power vs Load resistance 31, 32 Supply current vs Supply voltage 33

Closed-loop gain vs Frequency 39-44 Power dissipation/amplifier vs Output power 45, 46

1, 2, 4, 5, 7, 8, 10, 11,

13, 14, 16, 17, 34, 36

TPA112

FIGURE

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISE

vs vs

FREQUENCY FREQUENCY

Figure 1. Figure 2.

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10 kHz

0.1

0.01

THD+N −Total Harmonic Distortion + Noise − %

PO − Output Power − mW

1 10 50

VDD = 3.3 V RL = 32 Ω AV = 1 V/V CB = 1 µF

20 kHz

1 kHz

20 Hz

0.1

0.01

0.001

20 100 1k 10k 20k

VDD = 5 V PO = 60 mW RL = 32 Ω CB = 1 µF

AV = 10 mW

THD+N −Total Harmonic Distortion + Noise − %

f − Frequency − Hz

AV = 5 mW

AV = 1 mW

0.1

0.01

0.001

20 100 1k 10k 20k

VDD = 5 V RL = 32 Ω AV = 1 V/V CB = 1 µF

PO = 15 mW

THD+N −Total Harmonic Distortion + Noise − %

f − Frequency − Hz

PO = 30 mW

PO = 60 mW

THD+N −Total Harmonic Distortion + Noise − %

20 kHz

0.1

0.01

PO − Output Power − W

VDD = 5 V AV = 1 V/V RL = 32 Ω CB = 1 µF

10 kHz

1 kHz

20 Hz

0.002 0.01 0.1 0.2

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISE

vs vs

OUTPUT POWER FREQUENCY

Figure 3. Figure 4.

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISE

vs vs

FREQUENCY OUTPUT POWER

Figure 5. Figure 6.

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0.1

0.01

0.001

20 100 1k 10k 20k

VDD = 3.3 V RL = 10 kΩ PO = 100 µF CB = 1 µF

THD+N −Total Harmonic Distortion + Noise − %

f − Frequency − Hz

AV = 5 mW

AV = 2 mW

0.1

0.01

0.001

20 100 1k 10k 20k

VDD = 3.3 V RL = 10 kΩ AV = 1 V/V CB = 1 µF

THD+N −Total Harmonic Distortion + Noise − %

f − Frequency − Hz

PO = 45 µW

PO = 130 µW

PO = 90 µW

5 10 100 200

THD+N −Total Harmonic Distortion + Noise − %

20 Hz

0.01

0.001

PO − Output Power − µW

10 kHz

1 kHz

20 Hz

0.1

VDD = 3.3 V RL = 10 k Ω AV = 1 V/V CB = 1 µF

0.1

0.01

0.001

20 100 1k 10k 20k

VDD = 5 V RL = 10 kΩ PO = 300 µW CB = 1 µF

THD+N −Total Harmonic Distortion + Noise − %

f − Frequency − Hz

AV = 1

AV = 2

AV = 5

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISE

vs vs

FREQUENCY FREQUENCY

Figure 7. Figure 8.

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISE

vs vs

OUTPUT POWER FREQUENCY

Figure 9. Figure 10.

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0.1

0.01

0.001

20 100 1k 10k 20k

VDD = 5 V RL = 10 kΩ AV = 1 V/V CB = 1 µF

THD+N −Total Harmonic Distortion + Noise − %

f − Frequency − Hz

PO = 300 µW

PO = 200 µW

PO = 100 µW

0.1

0.01

0.001

5 10 100 500

VDD = 5 V RL = 10 kΩ AV = 1 V/V CB = 1 µ F

THD+N −Total Harmonic Distortion + Noise − %

20 Hz

1 kHz

20 kHz

10 kHz

PO − Output Power − µW

THD+N − Total Harmonic Distortion Plus Noise − %

f − Frequency − Hz

0.1

0.01

0.001 100 1k 10k 20k

AV = 1

AV = 2

AV = 5

= 3.3 V PO = 75 mW RL = 8 Ω CB = 1 µF

0.1

0.01

0.001

20 100 1k 10k 20k

VDD = 3.3 V RL = 8 Ω AV = 1 V/V

THD+N −Total Harmonic Distortion + Noise − %

f − Frequency − Hz

PO = 75 mW

PO = 15 mW

PO = 30 mW

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISE

vs vs

FREQUENCY OUTPUT POWER

Figure 11. Figure 12.

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISE

vs vs

FREQUENCY FREQUENCY

Figure 13. Figure 14.

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THD+N − Total Harmonic Distortion Plus Noise − %

f − Frequency − Hz

0.1

0.01

0.001 100 1k 10k 20k

AV = 1

AV = 2

AV = 5

= 5 V PO = 100 mW RL = 8 Ω CB = 1 µF

20 kHz

0.1

0.01

THD+N −Total Harmonic Distortion + Noise − %

PO − Output Power − W

10m 0.1 0.3

VDD = 3.3 V RL = 8 Ω AV = 1 V/V

10 kHz

1 kHz

20 Hz

0.1

0.01

0.001

20 100 1k 10k 20k

THD+N −Total Harmonic Distortion + Noise − %

f − Frequency − Hz

VDD = 5 V RL = 8 kΩ AV = 1 V/V

PO = 30 mW

PO = 60 mW

PO = 10 mW

20 kHz

0.1

0.01

THD+N −Total Harmonic Distortion + Noise − %

PO − Output Power − W

10m 0.1 1

1 kHz

20 Hz

10 kHz

VDD = 5 V RL = 8 Ω AV = 1 V/V

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISE

vs vs

OUTPUT POWER FREQUENCY

Figure 15. Figure 16.

TOTAL HARMONIC DISTORTION + NOISE TOTAL HARMONIC DISTORTION + NOISE

vs vs

FREQUENCY OUTPUT POWER

Figure 17. Figure 18.

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20 100 20k

f − Frequency − Hz

−50

−70

−90

−60

−80

−100

VDD = 3.3 V RL = 8 Ω to 10 kΩ

−40

−10

−30

−20

PSRR − Power Supply Rejection Ratio − dB

10k

CB = 0.1 µF

CB = 1 µF

CB = 2 µF

Bypass = 1.65 V

20 100 20k

f − Frequency − Hz

−50

−70

−90

−60

−80

−100

VDD = 5 V RL = 8 Ω to 10 kΩ

−40

−10

−30

−20

PSRR − Power Supply Rejection Ratio − dB

10k

CB = 0.1 µF

CB = 1 µF

CB = 2 µF

Bypass = 2.5 V

f − Frequency − Hz

20 100 1k 10k 20k

VDD = 3.3 V BW = 10 Hz to 22 kHz AV = 1 V/V RL = 8 Ω to 10 kΩ

− Output Noise Voltage − VµV

f − Frequency − Hz

20 100 1k 10k 20k

= 5 V

BW = 10 Hz to 22 kHz RL = 8 Ω to 10 kΩ AV = 1 V/V

− Output Noise Voltage − VµV

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

POWER SUPPLY REJECTION RATIO POWER SUPPLY REJECTION RATIO

vs vs

FREQUENCY FREQUENCY

Figure 19. Figure 20.

OUTPUT NOISE VOLTAGE OUTPUT NOISE VOLTAGE

vs vs

FREQUENCY FREQUENCY

Figure 21. Figure 22.

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20 100 20k

f − Frequency − Hz

−85

−95

−105

−90

−100

−110

PO = 25 mW VDD = 3.3 V RL = 32 Ω CB = 1 µF AV = 1 V/V

−80

−65

−75

−60

−70

Crosstalk − dB

10k

IN2 TO OUT1

IN1 TO OUT2

20 100 20k

f − Frequency − Hz

−75

−85

−95

−80

−90

−100

PO = 100 mW VDD = 3.3 V RL = 8 Ω CB = 1 µF AV = 1 V/V

−70

−55

−65

−50

−60

Crosstalk − dB

10k

IN2 TO OUT1

IN1 TO OUT2

20 100 10k

f − Frequency − Hz

−90

−100

−110

−95

−105

−85

−65

−80

−60

−75

−65

20k

= 5 V

PO = 25 mW CB = 1 µF RL = 32 Ω AV = 1 V/V

Crosstalk − dB

IN2 TO OUT1

IN1 TO OUT2

20 100 10k

f − Frequency − Hz

−80

−90

−100

−85

−95

−75

−55

−70

−50

−65

−60

20k

= 5 V

PO = 100 mW CB = 1 µF RL = 8 Ω AV = 1 V/V

Crosstalk − dB

IN2 TO OUT1

IN1 TO OUT2

CROSSTALK CROSSTALK

vs vs

FREQUENCY FREQUENCY

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

Figure 23. Figure 24.

CROSSTALK CROSSTALK

vs vs

FREQUENCY FREQUENCY

Figure 25. Figure 26.

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20 100 20k

f − Frequency − Hz

−50

−70

−90

−60

−80

−100

VDD = 3.3 V RL = 32 Ω CB = 1 µF

−40

−10

−30

−20

Mute Attenuation − dB

10k

20 100 10k

f − Frequency − Hz

−60

−80

−100

−70

−90

−50

−10

−40

−30

−20

20k

= 5 V

CB = 1 µF RL = 32 Ω

Mute Attenuation − dB

−20

−30°

0°

VDD = 3.3 V TA = 25°C No Load

100

30°

60°

90°

120°

150°

φ − Phase Margin

1k 100k

f − Frequency − Hz

10k 1M 10M100

Open-Loop Gain − dB

Phase

Gain

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

MUTE ATTENUATION MUTE ATTENUATION

vs vs

FREQUENCY FREQUENCY

Figure 27. Figure 28.

OPEN-LOOP GAIN AND PHASE MARGIN

FREQUENCY

Figure 29.

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f − Frequency − Hz

−20

100 1k 10k 10M1M100k

Open-Loop Gain − dB

100

VDD = 5 V TA = 25°C No Load

−30°

0°

30°

60°

90°

120°

150°

Phase

Gain

φ − Phase Margin

. .

RL − Load Resistance − Ω

100

16 32

24 40 64

120

8 48 56

THD+N = 1 % VDD = 3.3 V AV = 1 V/V

− Output Power − mW

250

100

16 32

200

150

24 40 64

300

8 48 56

THD+N = 1 % VDD = 5 V AV = 1 V/V

− Output Power − mW

RL − Load Resistance − Ω

OPEN-LOOP GAIN AND PHASE MARGIN

FREQUENCY

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

Figure 30.

OUTPUT POWER OUTPUT POWER

vs vs

LOAD RESISTANCE LOAD RESISTANCE

Figure 31. Figure 32.

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THD+N − Total Harmonic Distortion Plus Noise − %

f − Frequency − Hz

0.1

0.01

0.001 100 1k 10k 20k

= 1 V AV = 1 V/V RL = 10 kΩ CB = 1 µF

VDD − Supply Voltage − V

0.6

0.2

3 4

0.8

0.4

3.5 4.5

1.4

2.5 5 5.5

1.2

− Supply Current − mA

THD+N − Total Harmonic Distortion Plus Noise − %

f − Frequency − Hz

0.1

0.01

0.001 100 1k 10k 20k

= 5 V AV = 1 V/V RL = 10 kΩ CB = 1 µF

SNR − Signal−to−Ratio − dB

AV − Voltage Gain − V/V

104

100

5 7 9 10

102

862 43

= 1 V

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

SUPPLY CURRENT TOTAL HARMONIC DISTORTION + NOISE

vs vs

SUPPLY VOLTAGE FREQUENCY

Figure 33. Figure 34.

SIGNAL-TO-NOISE RATIO TOTAL HARMONIC DISTORTION + NOISE

vs vs

VOLTAGE GAIN FREQUENCY

Figure 35. Figure 36.

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Crosstalk − dB

f − Frequency − Hz

−60

−100

−150 100 1k 10k 20k

−70

−80

−90

−110

−120

−130

−140

VDD = 3.3 V VO = 1 V RL = 10 kΩ CB = 1 µF

IN2 to OUT1

IN1 to OUT2

Crosstalk − dB

f − Frequency − Hz

−60

−100

−150 100 1k 10k 20k

−70

−80

−90

−110

−120

−130

−140

VDD = 5 V VO = 1 V RL = 10 kΩ CB = 1 µF

IN1 to OUT2

IN2 to OUT1

Closed−Loop Gain − dB

f − Frequency − Hz

−10 100 1k 10k 1M

30 20

100k

200° 180°

160°

140°

120° 100°

80°

Gain

Phase

VDD = 3.3 V RI = 20 kΩ RF = 20 kΩ RL = 32 Ω CI = 1 µF AV = −1 V/V

CROSSTALK CROSSTALK

vs vs

FREQUENCY FREQUENCY

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

Figure 37. Figure 38.

CLOSED-LOOP GAIN AND PHASE

FREQUENCY

Figure 39.

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Closed−Loop Gain − dB

f − Frequency − Hz

−10 100 1k 10k 1M

30 20

100k

Phase

200° 180°

160°

140°

120° 100°

80°

Gain

Phase

VDD = 5 V RI = 20 kΩ RF = 20 kΩ RL = 32 Ω CI = 1 µF AV = −1 V/V

Closed−Loop Gain − dB

f − Frequency − Hz

−20 100 1k 10k 1M

40 20

100k

Phase

200° 180°

160°

140°

120° 100°

80° 60°

Gain

Phase

VDD = 3.3 V RI = 20 kΩ RF = 20 kΩ RL = 8 Ω CI = 1 µF AV = −1 V/V

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

CLOSED-LOOP GAIN AND PHASE

FREQUENCY

CLOSED-LOOP GAIN AND PHASE

FREQUENCY

Figure 40.

Figure 41.

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Closed−Loop Gain − dB

f − Frequency − Hz

−10 100 1k 10k 1M

30 20

100k

Phase

200° 180°

160°

140°

120° 100°

80°

Gain

Phase

VDD = 3.3 V RI = 20 kΩ RF = 20 kΩ RL = 10 kΩ CI = 1 µF AV = −1 V/V

Closed−Loop Gain − dB

f − Frequency − Hz

−20 100 1k 10k 1M

100k

Phase

200° 180°

160°

140°

120° 100°

80° 60°

40°

Gain

Phase

VDD = 5 V RI = 20 kΩ RF = 20 kΩ RL = 8 Ω CI = 1 µF AV = −1 V/V

CLOSED-LOOP GAIN AND PHASE

FREQUENCY

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

CLOSED-LOOP GAIN AND PHASE

FREQUENCY

Figure 42.

Figure 43.

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Closed−Loop Gain − dB

f − Frequency − Hz

−10 100 1k 10k 1M

30 20

100k

Phase

200° 180°

160°

140°

120° 100°

80°

Gain

Phase

VDD = 5 V RI = 20 kΩ RF = 20 kΩ RL = 10 kΩ CI = 1 µF AV = −1 V/V

Amplifier Power − mW

Load Power − mW

80 120 180 200

14010020 6040

160

VDD = 3.3 V

8 Ω

16 Ω

64 Ω

32 Ω

Amplifier Power − mW

Load Power − mW

180

100

80 120 180 200

120

14010020 6040

160

140

160

VDD = 5 V

8 Ω

16 Ω

64 Ω

32 Ω

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

CLOSED-LOOP GAIN AND PHASE

FREQUENCY

POWER DISSIPATION/AMPLIFIER POWER DISSIPATION/AMPLIFIER

vs vs

OUTPUT POWER OUTPUT POWER

Figure 45. Figure 46.

Figure 44.

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Gain  





Effective Impedance 

RFR

RF R

co(lowpass)



2 R

FCF

co(highpass)



2 R



2 R

co(highpass)

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

APPLICATION INFORMATION

GAIN SETTING RESISTORS, R

The gain for the TPA112 is set by resistors R

and R

and RIaccording to Equation 1 .

Given that the TPA112 is an MOS amplifier, the input impedance is high. Consequently, input leakage currents are not generally a concern, although noise in the circuit increases as the value of R certain range of R

values is required for proper start-up operation of the amplifier. Taken together, it is

increases. In addition, a

recommended that the effective impedance seen by the inverting node of the amplifier be set between 5 kΩ and 20 kΩ. The effective impedance is calculated in Equation 2 .

As an example, consider an input resistance of 20 kΩ and a feedback resistor of 20 kΩ. The gain of the amplifier would be -1 and the effective impedance at the inverting terminal would be 10 kΩ, which is within the recommended range.

For high-performance applications, metal film resistors are recommended because they tend to have lower noise levels than carbon resistors. For values of R formed from R

and the inherent input capacitance of the MOS input structure. For this reason, a small

above 50 kΩ, the amplifier tends to become unstable due to a pole

compensation capacitor of approximately 5 pF should be placed in parallel with RF. In effect, this creates a low-pass filter network with the cutoff frequency defined in Equation 3 .

For example, if R

is 100 kΩ and C

is 5 pF then f

co(lowpass)

is 318 kHz, which is well outside the audio range.

(1)

(2)

(3)

INPUT CAPACITOR, C

In the typical application, input capacitor CIis required to allow the amplifier to bias the input signal to the proper dc level for optimum operation. In this case, C

and R

form a high-pass filter with the corner frequency

determined in Equation 4 .

The value of CIis important to consider, as it directly affects the bass (low-frequency) performance of the circuit. Consider the example where R

is 20 kΩ and the specification calls for a flat bass response down to 20 Hz.

Equation 4 is reconfigured as Equation 5 .

In this example, C

is 0.4 µF, so one would likely choose a value in the range of 0.47 µF to 1 µF. A further

consideration for this capacitor is the leakage path from the input source through the input network (R the feedback resistor (R

) to the load. This leakage current creates a dc offset voltage at the input to the amplifier

that reduces useful headroom, especially in high-gain applications (> 10). For this reason a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications, as the dc level there is held at V

/2, which is likely higher

that the source dc level. It is important to confirm the capacitor polarity in the application.

(4)

(5)

, CI) and

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Midrail

BYPASS

Midrail

BYPASS

TLV2460

a) Midrail Voltage Generator Using a Simple

Resistor-Divider

b) Buffered Midrail Voltage Generator to Provide Low Output Impedance

TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

APPLICATION INFORMATION (continued) POWER SUPPLY DECOUPLING, C

The TPA112 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure that the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved by using two capacitors of different types that target different types of noise on the power supply leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor; typically, 0.1 µF, placed as close as possible to the device V lower frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near the power amplifier is recommended.

MIDRAIL VOLTAGE

The TPA112 is a single-supply amplifier; so, it must be properly biased to accommodate audio signals. Normally, the amplifier is biased at V biasing the amplifier at a point other than V applications where the circuitry driving the TPA112 has a different midrail voltage, it might make sense to use the same midrail voltage for the TPA112, and possibly eliminate the use of the dc-blocking capacitors.

The two concerns with the midrail voltage source are the amount of noise present and its output impedance. Any noise present on the midrail voltage source that is not present on the audio input signal will be input to the amplifier, and passed to the output (and increased by the gain of the circuit). Common-mode noise is cancelled out by the differential configuration of the circuit.

The output impedance of the circuit used to generate the midrail voltage needs to be low enough so as not to be influenced by the audio signal path. A common method of generating the midrail voltage is to form a voltage divider from the supply to ground, with a bypass capacitor from the common node to ground. This capacitor improves the PSRR of the circuit. However, this circuit has a limited range of output impedances; so, to achieve low output impedances, the voltage generated by the voltage divider is fed into a unity-gain amplifier to lower the output impedance of the circuit.

/2, but it can actually be biased at any voltage between V

lead, works best. For filtering

and ground. However,

/2 reduces the amplifier's maximum output swing. In some

If a voltage step is applied to a speaker, it causes a noise pop. To reduce popping, the midrail voltage should rise at a subsonic rate. That is, a rate less than the rise time of a 20-Hz waveform. If the voltage rises faster than that, there is the possibility of a pop from the speaker.

Pop can also be heard in the speaker if the midrail voltage rises faster than the charge of either the input coupling capacitor or the output coupling capacitor. If midrail rises first, the charging of the input and output capacitors is heard in the speaker. To keep this noise as low as possible, the relationship shown in Equation 6 should be maintained.

Figure 47. Midrail Voltage Generator

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

 R

SOURCE







CIR





RLC



 R

SOURCE







CIR



6.25  50

(out high)



2 R

LCC

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

APPLICATION INFORMATION (continued)

Where C

BYPASS

voltage divider (the parallel combination of the two resistors). For example, if the voltage divider is constructed using two 20-kΩ resistors, then R

is the value of the bypass capacitor, and R

SOURCE

is 10 kΩ.

SOURCE

is the equivalent source impedance of the

TPA112

(6)

MIDRAIL BYPASS CAPACITOR, C

The midrail bypass capacitor C

serves several important functions. During start-up, C

determines the rate at

which the amplifier starts up. This helps to push the start-up pop noise into the subaudible range (so slow it can not be heard). The second function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This noise is from the midrail generation circuit internal to the amplifier. The capacitor is fed from the resistor divider with equivalent resistance of R

. To keep the start-up pop as low as possible, the

SOURCE

relationship shown in Equation 7 should be maintained.

As an example, consider a circuit where C

is 1 µF, R

SOURCE

= 160 kΩ, C

is 1 µF, and RIis 20 kΩ. Inserting

these values into the Equation 8 results in:

which satisfies the rule. Recommended values for bypass capacitor C

are 0.1 µF to 1 µF, ceramic or tantalum

low-ESR, for the best THD and noise performance.

OUTPUT COUPLING CAPACITOR, C

In the typical single-supply, single-ended (SE) configuration, an output coupling capacitor (C

) is required to

block the dc bias at the output of the amplifier, thus preventing dc currents in the load. As with the input coupling capacitor, the output coupling capacitor and impedance of the load form a high-pass filter governed by Equation 9 .

The main disadvantage, from a performance standpoint, is that the typically small load impedances drive the low-frequency corner higher. Large values of C example where a C

of 68 µF is chosen and loads vary from 32 Ω to 47 kΩ. Table 1 summarizes the frequency

are required to pass low frequencies into the load. Consider the

response characteristics of each configuration.

(7)

(8)

(9)

Table 1. Common Load Impedances vs Low Frequency

Output Characteristics in SE Mode

32 Ω 68 µF 73 Hz 10,000 Ω 68 µF 0.23 Hz 47,000 Ω 68 µF 0.05 Hz

LOWEST FREQUENCY

As Table 1 indicates, headphone response is adequate and drive into line level inputs (a home stereo for example) is good.

The output coupling capacitor required in single-supply, SE mode also places additional constraints on the selection of other components in the amplifier circuit. With the rules described earlier still valid, add the following relationship:

• Output Pulldown Resistor, R – Placing a 100-Ω resistor, R

+ R

, from the output side of the coupling capacitor to ground ensures the coupling

capacitor, CC, is charged before a plug is inserted into the jack. Without this resistor, the coupling capacitor would charge rapidly upon insertion of a plug, leading to an audible pop in the headphones.

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TPA112

SLOS212E – AUGUST 1998 – REVISED JUNE 2004

– Placing a 20-kΩ resistor, R

, from the output of the IC to ground ensures that the coupling capacitor fully

discharges at power down. If the supply is rapidly cycled without this capacitor, a small pop may be audible in 10-kΩ loads.

• Using Low-ESR Capacitors

– Low-ESR capacitors are recommended throughout this application. A real capacitor can be modeled simply

as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance, the more the real capacitor behaves like an ideal capacitor.

5-V VERSUS 3.3-V OPERATION

The TPA112 is designed for operation over a supply range of 2.5 V to 5.5 V. This data sheet provides full specifications for 5-V and 3.3-V operation because these are considered to be the two most common standard voltages. There are no special considerations for 3.3-V versus 5-V operation as far as supply bypassing, gain setting, or stability. The most important consideration is that of output power. Each amplifier in the TPA112 can produce a maximum voltage swing of V V

= 2.3 V, as opposed to V

O(PP)

maximum output power into the load before distortion begins to become significant.

– 1 V. This means, for 3.3-V operation, clipping starts to occur when

= 4 V for 5-V operation. The reduced voltage swing subsequently reduces

PACKAGE OPTION ADDENDUM

www.ti.com

7-May-2007

PACKAGING INFORMATION

Orderable Device Status

(1)

Package

Type

Package

Drawing

Pins Package

Qty

Eco Plan

TPA112D ACTIVE SOIC D 8 75 Green (RoHS &

no Sb/Br)

TPA112DG4 ACTIVE SOIC D 8 75 Green (RoHS &

no Sb/Br)

TPA112DGN ACTIVE MSOP-

Power

DGN 8 80 Green (RoHS &

no Sb/Br)

PAD

TPA112DGNG4 ACTIVE MSOP-

Power

DGN 8 80 Green (RoHS &

no Sb/Br)

PAD

TPA112DGNR ACTIVE MSOP-

Power

DGN 8 2500 Green (RoHS &

no Sb/Br)

PAD

TPA112DGNRG4 ACTIVE MSOP-

Power

DGN 8 2500 Green (RoHS &

no Sb/Br)

PAD

TPA112DR ACTIVE SOIC D 8 2500 Green (RoHS &

no Sb/Br)

TPA112DRG4 ACTIVE SOIC D 8 2500 Green (RoHS &

no Sb/Br)

TPA112EVM OBSOLETE 0 TBD Call TI Call TI

(1)

The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in

a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device.

(2)

Lead/Ball Finish MSL Peak Temp

CU NIPDAU Level-1-260C-UNLIM

(3)

(2)

Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check

http://www.ti.com/productcontent for the latest availability information and additional product content details.

TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements

for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)

(3)

MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder

temperature.

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In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

Addendum-Page 1

PACKAGE MATERIALS INFORMATION

www.ti.com

TAPE AND REEL INFORMATION

11-Mar-2008

*All dimensions are nominal

Device Package

TPA112DGNR MSOP-

Power

TPA112DR SOIC D 8 2500 330.0 12.4 6.4 5.2 2.1 8.0 12.0 Q1

Type

PAD

Package

Drawing

Pins SPQ Reel

Diameter

(mm)

DGN 8 2500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1

Reel

Width

W1 (mm)

A0 (mm) B0 (mm) K0 (mm) P1

(mm)W(mm)

Pin1

Quadrant

Pack Materials-Page 1

PACKAGE MATERIALS INFORMATION

www.ti.com

11-Mar-2008

*All dimensions are nominal

Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)

TPA112DGNR MSOP-PowerPAD DGN 8 2500 358.0 335.0 35.0

TPA112DR SOIC D 8 2500 346.0 346.0 29.0

Pack Materials-Page 2

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TEXAS INSTRUMENTS TPA112 Technical data

Specifications and Main Features

Frequently Asked Questions

User Manual

FEATURES DESCRIPTION

FUNCTIONAL BLOCK DIAGRAM

ABSOLUTE MAXIMUM RATINGS

DISSIPATION RATING TABLE

RECOMMENDED OPERATING CONDITIONS

DC ELECTRICAL CHARACTERISTICS

AC OPERATING CHARACTERISTICS

DC ELECTRICAL CHARACTERISTICS

AC OPERATING CHARACTERISTICS

AC OPERATING CHARACTERISTICS

AC OPERATING CHARACTERISTICS

TYPICAL CHARACTERISTICS

Table of Graphs

APPLICATION INFORMATION

MIDRAIL VOLTAGE

5-V VERSUS 3.3-V OPERATION