This application note describes a low power , (output power of 4.1W) general purpose
adapter which is able to handle a wide range input voltages (88V
(Order Code STEVAL-ISA011V1) is based on the Viper12A monolithic device that has the
power switch as well as the basic control function needed to implement a current mode
flyback converter.
In order to improve regulation, the feedback loop is designed to have enough bandwidth so
the converter can react on time to load changes. As is shown in the
Load Regulation Tests on page 20
, the board is able to handle high load step changes with
very low variations in the output voltage.
The flyback converter is designed to work in Discontinuous Conduction Mode (DCM) in all
operating conditions (i.e. Minimum Input Voltage, Maximum Load), because it provides
better dynamic performance.
1.1 Primary Side
1.1.1 Step 1, Input Capacitor Selection
Section 2.3: Dynamic
The first design step is to calculate the input capacitor value (C2a + C2b see
Where,
ΔT = the time between the two conduction cycles of the input bridge diodes, and
f
= line frequency.
line
using
The calculated value of C
and C2b, see
This means that C
STEVAL-ISA011V Demo Board Schematic on page 30
= 20µF. This value was selected because the tolerance for an
IN
IN
Equation 1
electrolytic capacitor is usually around 20%.
1.1.2 Step 2, Transformer Selection
The next step is selecting a transformer with a Primary Inductance (LP) that allows the
system to work at the boundary between Continuous Conduction Mode (CC M) and
Discontinuous Conduction Mode (DCM). The worst case is minimum input voltage and full
load. This value is expressed as:
The conduction losses in the main switch depend on the VIPer12A I
resistance, and are expressed as:
Equation 9
Where,
P
VIPer12A
r
ds(on)
= VIPer12A conduction losses, and
= VIPer12A ON resistance.
1.2 Secondary Side
In order to select the output rectifier (secondary) diode D11, the designer needs to know the
maximum reverse voltage that the diode has to sustain, as well as the average and root
mean square of the current flowing through it (see
page 30
Equation 10
). V
is calculated as follows:
R(max)
P
VIPer12Ards on()
V
Rmax()VOUT
PRMS(max)
2
I
⋅=
PRMS max()
STEVAL-ISA011V1 Schematic on
V
OUT
------- ---------- -
V
⋅+=
R
DC m ax()
V
and ON
Where,
V
V
V
V
= maximum reverse vo ltage,
R(max)
= output voltage,
OUT
= reflected voltage, and
R
DC(max)
= selected maximum input voltage.
A commonly used selection method is to choose a diode with a 40% to 50% safety margin
from the value given by the V
calculation when a Schottky diode is used, or a safety
R(max)
margin of 20% to 30% if a standard “fast” diode is used. The safety margin prevents diode
breakdown from oscillation caused by circuit parasitic elements (e.g. transformer secondary
inductance leakage or parasitic diode capacitance) when the MOSFET is turned ON.
If the calculated V
the D
value is about 34V. This makes the STPS340U (with 40V breakdown voltage) an
11
is 23V and a Schottky diode is used (adding a 50% safety margin),
The output capacitor selection (C11, see
depends on the output voltage ripple specification (ΔV
STEVAL-ISA011V1 Schematic on page 30
= 300mV), and the ripple current
O
)
rate of the capacitor itself. The output voltage ripple is mainly due to the Equivalent Series
Resistor (ESR), so we have to select a capacitor with an ESR lower than the maximum
allowed ESR value:
Equation 17
ΔV
ESR
MAX
---------------=
I
PKS
O
Where,
ESR
ΔV
I
PKS
= maximum allowed ESR rating,
MAX
= output voltage ripple, and
O
= peak current at secondary winding.
The AC component of the current flowing through the output diode is also that of the current
flowing through the capacitor. The C11 capacitor current rate has to be higher than the
calculated current, which is expressed as:
Equation 18
I
CAPRMS
2
I
DRMS
2
I
–=
O
Where,
I
CAPRMS
I
DRMS
I
O
= capacitor current root mean square,
= diode current root mean square, and
= output current.
The MBZ Type 1500 μF 10V by RUBYCON capacitor was selected for this application.
All of the calculations for the transformer design are complete. They include:
●Primar y Inducta nce,
●Turns Ratio, and
●Winding Current Values (RMS, Average, and Peak).
Notes:
1.In order to prevent transformer saturation during the start-up phase, the current limit of
the VIPer12A (I
= 480mA, see datasheet for details) must be considered as the peak
LIM
current.
2. For thermal limits (power dissipated in the magnetic core), the peak current (calculated
in
Equation 6: on page 7
) must be used.
3. The RMS value of the current flowing through the windings is used first fo r calculating
the power dissipated in the windings, then fo r winding size selection.
The transformer (reference number SRW16ES_E44H013) was designed and manufactured
by TDK using aforementioned the data.
1.4 Feedback Loop
The transfer function 'control-to-output' for a flyback converter operating in DCM is given by
the following formula:
●The flyback zero value (for two poles, one zero compensation network) is expressed
= equivalent series resistor output.
OUT
STEVAL-ISA011V1 Schematic on page 30
),
as:
Equation 22
OUT
1
ESR
⋅
OUT
z
fly
------------------------------------------- -=
C
Where,
z
= flyback zer o.
fly
One pole is located at zero frequency in order to maximize the precision of the
regulation. Compensation zero was used in order to compensate the p
it has to be located between one-half and double the p
frequency. The last pole of the
fly
compensation network is used to compensate flyback zero due to the ESR.
Note: Using these resistance and capacitance values as guidelines will provide the
user with a stable loop as well as the required converter bandwidth.
The tests performed with the STEVAL-ISA011V1 demo board are used to evaluate the
converter behavior in terms of:
●efficiency,
●safe operating area of the devices,
●line regulation, and
●load regu lation.
2.1 Start-up Tests
The diagnostic board will handle a wide range of AC input voltage (88VAC to 265VAC), and
its maximum output power is 4.1W with one output of 4.5V. Its maximum output current is
900mA (see
For flyback conv erters, the most critical conditions for the main switch in terms of Maximum
Drain Current and of Maximum Drain Voltage (when no abnorm al event occurs), are those
that exist during the start-up phase. The maximum values for drain voltage and current are
measured in both full load and no load conditions (the two extreme points in terms of load),
and for minimum, maximum, and nominal input voltages (see
Table 1
).
Table
).
All the measured values are within the rated maximum values of the VIPer12A so they are
not critical for device operation.
These tests verify the board’s device and component temperatures.
temperatures (the most stress measured, in terms of power dissipation) for the board’s main
components.
Note: The tests were performed at 25°C (ambient temperature), in Full Load conditions.
Table 3.Component Critical Temperature Measurements
These tests monitor and verify the stability and quality of the system response to load
changes, in terms of speed and overshoot.
Figure 11
load changes, for the minimum (88V
and 230V
, and
Figure 12 on page 22
).
AC
AC
During these tests, load changes from a minimum of 180mA to a maximum of 900mA are
applied to the circuit as squarewaves, with 3ms periods and a duty cycle of 50%.
●The output voltage (Ch3) has a variation of some tenths of a mV (about 40mV), with
some mV overshoots. These results indicate very good dynamic behavior on the part of
the system.
●The VIPer12A feedback pin voltage (Ch1) in
input voltage is 265V
, the load is 180mV (its minimum value) while the output and
AC
feedback pin voltages show some oscillation. This oscill ation is not relat ed to a low
phase margin of the Loop Gain, but is related to the VIPer12A Burst mode operation.
Note: Even with the oscillation, the output voltages are still regulated well.
Figure 9
show the waveforms as they occur during the circuit
Pre-compliant tests with European Normative EN55022 for electromagnetic interference
(EMI) were performed.
illustrate that the conducted EMI induced by the converter to the main are below the
normative limits.
Figure 21
and
Figure 22
, and
Figure 23
and
Figure 24 on page 29
Figure 21
Note:
to 30MHz frequency range.
Figure 21. 115V
Figure 22. 115V
through
Line Voltage
AC
Line Neutra l
AC
Figure 24
show the Input current spectrum to be inside the 150kHz
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