Philips Power Semiconductor Service Manual

S.M.P.S. Power Semiconductor Applications

Philips Semiconductors

CHAPTER 2

Switched Mode Power Supplies

2.1 Using Power Semiconductors in Switched Mode Topologies (including transistor selection guides)

2.2 Output Rectification

2.3 Design Examples

2.4 Magnetics Design

2.5 Resonant Power Supplies

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Using Power Semiconductors in Switched Mode Topologies

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2.1.1 An Introduction to Switched Mode Power Supply Topologies

Formanyyearstheworldof power supply designhas seen a gradual movement away from the use of linear power suppliestothemorepracticalswitchedmodepower supply (S.M.P.S.). The linear power supply contains a mains transformer and a dissipative seriesregulator. Thismeans the supply has extremely large and heavy 50/60 Hz transformers, and also very poor power conversion efficiencies,both seriousdrawbacks. Typical efficienciesof 30% are standard for a linear. This compares with efficiencies of between 70 and 80%, currently available using S.M.P.S. designs.

Furthermore,by employing high switching frequencies, the sizes of the power transformer and associated filtering components in the S.M.P.S. are dramatically reduced in comparison to the linear. For example, an S.M.P.S. operating at 20kHz produces a 4 times reduction in component size, and this increases to about 8 times at 100kHz and above. This means an S.M.P.S. design can producevery compactand lightweightsupplies.This isnow an essential requirement for the majority of electronic systems.The supply must slot into an ever shrinking space left for it by electronic system designers.

Outline

At the heart of theconverter is the highfrequency inverter section, where the input supply is chopped at very high frequencies(20to200kHzusingpresenttechnologies) then filtered and smoothed to produce dc outputs. The circuit configuration which determines how the power is

transferred is called the TOPOLOGY of the S.M.P.S.,and is an extremely important part of the design process. The topology consists of an arrangement of transformer, inductors, capacitors and power semiconductors (bipolar or MOSFET power transistors andpower rectifiers).

Presently, there is a very wide choice of topologies available, each one having its own particular advantages and disadvantages, making it suitable for specific power supply applications. Basic operation, advantages, drawbacks and most common areas of use for the most commontopologies arediscussed inthe followingsections. A selection guide to the Philips range of power semiconductors (including bipolars, MOSFETs and rectifiers) suitable for usein S.M.P.S. applications isgiven at the end of each section.

(1) Basic switched mode supply circuit.

An S.M.P.S. can be a fairly complicated circuit, as can be seen from the block diagram shown in Fig. 1. (This configuration assumes a 50/60Hz mains input supply is used.) The ac supply is first rectified, and then filtered by the input reservoir capacitor to produce a rough dc input supply. This level can fluctuate widely due to variations in the mains. In addition the capacitance on the input has to be fairly large to hold up the supply in case of a severe droop in the mains. (The S.M.P.S. can also be configured tooperate fromany suitable dcinput, in thiscase thesupply is called a dc to dc converter.)

ac input

supply

Input rectification

and filtering

duty cycle

High

Frequency

switch

mosfet or

control

bipolar

control circuitry

Power

Transformer

PWM

OSC

Output rectification

and filtering

Vref

Fig. 1. Basic switched mode power supply block diagram.

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Theunregulated dc isfed directly to the central block of the supply, the high frequency power switching section. Fast switchingpowersemiconductordevices suchas MOSFETs and Bipolars are driven on and off, and switch the input voltage across the primary of the power transformer. The drive pulses are normally fixed frequency (20 to 200kHz) and variable duty cycle. Hence, a voltage pulse train of suitable magnitude and duty ratio appears on the transformer secondaries. This voltage pulse train is appropriately rectified, and then smoothed by the output filter, which is either a capacitor or capacitor / inductor arrangement, depending upon the topology used. This transferofpowerhas tobe carriedout withthelowest losses possible, to maintain efficiency. Thus, optimum design of thepassive andmagnetic components, andselection ofthe correct power semiconductors is critical.

Regulation of the output to provide a stabilised dc supply is carried out by the control / feedback block. Generally, mostS.M.P.S. systemsoperateon a fixed frequency pulse width modulation basis, where the duration of the on time ofthe drive to the power switch is varied on acyclebycycle basis. This compensates for changes in the input supply and output load. The output voltage is compared to an accurate reference supply, and theerror voltageproduced by the comparator is used by dedicated control logic to terminatethedrivepulsetothe mainpower switch/switches atthe correct instance. Correctly designed, this will provide a very stable dc outputsupply.

It is essential that delays in the control loop are kept to a minimum,otherwisestabilityproblemswouldoccur. Hence, veryhigh speed components must be selected forthe loop. In transformer-coupled supplies, in order to keep the isolation barrier intact, some type of electronic isolation is required in the feedback. Thisis usually achievedby using asmall pulse transformer or anopto-isolator,hence adding to the component count.

In most applications, the S.M.P.S. topology contains a power transformer. This provides isolation, voltagescaling through the turns ratio, and the ability to provide multiple outputs. However, there are non-isolated topologies (without transformers) such as the buck and the boost converters, where the power processing is achieved by inductive energy transfer alone. All of the more complex arrangements are based on thesenon-isolated types.

(2) Non-Isolated converters.

The majority of the topologies used in today’s converters are all derived from the following three non-isolated versions called the buck, the boost and the buck-boost. These are the simplest configurations possible, and have the lowest component count, requiring only one inductor, capacitor, transistor and diode to generate their single output.If isolation between theinputand output is required, a transformer must be includedbefore the converter.

(a) The Buck converter.

The forward converter family which includes thepush-pull and bridge types, are all based on the buck converter, shown in Fig. 2. Its operation is straightforward. When switch TR1 is turned on, the input voltage is applied to inductor L1 and power is delivered to the output. Inductor current also builds up according to Faraday’s law shown below:-

V =L

When the switch is turned off, the voltage across the inductor reverses and freewheel diode D1 becomes forwardbiased.Thisallows theenergystoredin theinductor tobe deliveredto theoutput. Thiscontinuous currentis then smoothedby outputcapacitor Co. Typical buck waveforms are also shown in Fig. 2.

toff

T = ton + toff

Vin

Applied voltage

Inductor current

Inductor voltage

TR1

current

A 0

L 0

Iin

ton toff

TR1

CONTROL

CIRCUIT

ton

Vin

Vin - Vo

Fig. 2 Buck Regulator (step-down).

The LC filter has an averaging effect on the applied pulsating input, producinga smoothdc output voltageand current, with very small ripple components superimposed. The average voltage/sec across the inductor over a complete switching cycle must equal zero in the steady state. (The same applies toall ofthe regulatorsthat willbe discussed.)

Continuous mode

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Neglecting circuit losses, the average voltage at the input side of the inductor is VinD, while Vois the output side voltage. Thus, in the steady state, for the average voltage acrossthe inductor to bezero, the basic dcequation ofthe buck is simply:-

= D

D is the transistor switch duty cycle, defined as the conduction time divided by one switching period, usually expressed in the form shownbelow:-

; where T =t

D =

on+toff

Thus,thebuck isa stepdowntype, wherethe outputvoltage isalwayslower thantheinput.(Since Dnever reachesone.) Output voltage regulation is provided by varying the duty cycle of the switch. The LC arrangement provides very effective filtering of the inductor current. Hence, the buck and its derivatives all have very low output ripple characteristics. The buck is normally always operated in continuous mode ( inductor current never falls to zero) where peak currents are lower, and the smoothing capacitor requirements are smaller. There are no major control problems with the continuous mode buck.

(b) The Boost Converter.

Operation of another fundamental regulator, the boost, shown in Fig. 3 is more complex than the buck. When the switch is on,diode D1 is reversebiased, and Vinis applied across inductor, L1. Current builds up in the inductor to a peak value, either from zero current in a discontinuous mode, or an initial value in the continuousmode. When the switch turns off, the voltage across L1 reverses, causing thevoltage at the diode to rise above the input voltage.The diodethen conductsthe energy stored inthe inductor, plus energy direct from the supply to the smoothing capacitor and load. Hence, Vois always greater than Vin, making this a stepup converter. For continuous mode operation, the boost dc equation is obtained by a similar process as for the buck, and is givenbelow:-

1− D

Again, the output only depends upon the input and duty cycle.Thus, by controlling the duty cycle, outputregulation is achieved.

From the boost waveforms shown in Fig. 3, it is clear that thecurrent supplied to the output smoothingcapacitor from the converter is the diode current, which will always be discontinuous. This means that the output capacitor must be large, with a low equivalent series resistance (e.s.r) to

produce a relatively acceptable output ripple. This is in contrast to the buck output capacitor requirements describedearlier.Ontheother hand,the boostinput current is the continuous inductor current, and this provides low input ripple characteristics. The boost is very popular for capacitive load applications such as photo-flashers and batterychargers.Furthermore, thecontinuous inputcurrent makestheboost apopularchoice asapre-regulator,placed before the main converter. The main functions being to regulate the input supply, and to greatly improve the line powerfactor. This requirementhas become very important in recent years, in a concertedeffort to improve the power factor of the mains supplies.

TR1

CONTINUOUS MODE

Vin

TR1 voltage

Inductor current

Diode current

TR1

current

CONTROL

CIRCUIT

ton toff

Fig. 3 Boost Regulator (step-up).

If the boost is used in discontinuous mode, the peak transistor and diode currentswill behigher, andthe output capacitor will need to be doubled in size to achieve the sameoutput rippleas in continuousmode. Furthermore, in discontinuous operation, the output voltage also becomes dependent on the load, resulting in poorerload regulation.

Unfortunately, there are major control and regulation problems with the boost when operated in continuous mode. The pseudo LC filter effectively causes a complex second order characteristic in the small signal (control) response. In the discontinuous mode, the energy in the inductorat the startof each cycle is zero. Thisremoves the inductancefrom the small signalresponse, leaving only the output capacitance effect. This produces a much simpler response, which is far easierto compensate and control.

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Thevery popular flyback converter(see section 5(a)) is not actually derived solely from the boost. The flyback only delivers stored inductor energy during the switch off-time. The boost, however, also delivers energy from the input. The flyback is actually based on a combined topology of the previous two, called the buck-boost or non isolated flyback regulator. This topology isshown in Fig. 4.

Vin

TR1

CONTROL

CIRCUIT

Step up / down Polarity inversion

Fig. 4 Buck-Boost (Flyback) Regulator.

When the switch ison, thediode isreverse biased andthe inputis connectedacrossthe inductor,whichstores energy as previously explained. At turn-off, the inductor voltage reverses and the stored energy is then passed to the capacitor and load through the forward biased rectifier diode.

-Vo

The waveforms are similar to the boost except that the transistorswitch now has to support thesum of Vinand Vo across it. Clearly, both the input and output currents must be discontinuous. There is also a polarity inversion, the output voltage generated is negative with respect to the input. Close inspection reveals that the continuous mode dc transfer function is asshown below:-

1− D

Observation shows that the value of the switch duty ratio, D canbe selected such that the output voltage can either be higher or lower than the input voltage. This gives the converter the flexibility to either step up or step down the supply.

Thisregulator also suffers from the same continuous mode control problems as the boost, and discontinuous modeis usually favoured.

Since both input and output currents are pulsating, low ripple levels are very difficult to achieve using the buck-boost. Very large outputfilter capacitorsare needed, typically up to 8 timesthat of a buck regulator. The transistor switch also needs to be ableto conduct the highpeakcurrent,aswellas supportingthehigher summed voltage.Theflyback regulator(buck-boost) topologyplaces the most stress on the transistor. The rectifier diode also hasto carry high peak currentsandso the r.m.s conduction losses will be higher thanthose of the buck.

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(3) Transformers in S.M.P.S. converters.

The non-isolated versions have very limited use, such as dc-dcregulators onlycapable of producing asingle output. Theoutput range is also limited by the input and dutycycle. The addition of a transformer removes most of these constraints and provides a converter with the following advantages:-

1) Input to output isolation is provided. This is normally alwaysnecessaryfor 220/ 110V mainsapplications, where a degree of safety isprovided for the outputs.

2) The transformer turns ratio can be selected to provide outputs widely different from the input; non-isolated versions are limited to a range of approximately 5 times. By selecting the correct turns ratio, the duty cycle of the converter can also be optimised and the peak currents flowing minimised. The polarity of each output is also selectable, dependent upon the polarity of the secondary w.r.t the primary.

3) Multiple outputs are very easily obtained, simply by adding more secondary windings tothe transformer. There are some disadvantages withtransformers, suchas theiradditional size,weight and powerloss. Thegeneration of voltage spikes due to leakageinductance may alsobe a problem.

Theisolatedconverters tobecovered aresplitintotwo main categories, called asymmetrical and symmetrical converters, depending upon how the transformer is operated.

asymmetrical converters

forward

symmetrical converters

2Bs

Fig. 5 Comparative core usage of asymmetrical and

symmetrical converters.

converter

symmetrical

converters

available flux swing

flyback converter

Inasymmetrical convertersthemagnetic operatingpoint of the transformer is always in one quadrant i.e the flux and the magnetic field never changes sign. Thecore has to be reseteachcycle toavoid saturation, meaning that only half of the usable flux is ever exploited. This can be seen in Fig. 5, which shows the operating mode of eachconverter. The flyback and forward converter are both asymmetrical types.Thediagramalso indicatesthat theflyback converter is operated at a lower permeability (B/H) and lower inductance than the others. This is because the flyback transformeractuallystoresalloftheenergy beforedumping into the load, hence an air gap is required to store this energyand avoidcoresaturation. Theair gap hastheeffect of reducing the overall permeability of the core. All of the other converters have true transformer action and ideally store no energy, hence, noair gap is needed.

Inthesymmetrical converterswhichalways requirean even number of transistor switches, the full available flux swing in both quadrants of the B / H loop is used, thus utilising the core much more effectively. Symmetrical converters cantherefore producemore power thantheir asymmetrical cousins. The 3 major symmetrical topologies used in practiceare thepush-pull, the half-bridgeand thefull bridge types.

Table 1 outlines the typical maximum output power available from each topology using present day technologies:-

Converter Topology Typical max output power

Flyback 200W

Forward 300W

Two transistor forward / 400W

flyback

Push-pull 500W Half-Bridge 1000W Full-Bridge >1000W

Table 1. Converter output power range.

Manyother topologies exist, butthe types outlined in Table 1 are by far the most commonly used in present S.M.P.S. designs. Each is now looked at in more detail, with a selection guide for the most suitable Philips power semiconductors included.

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(4) Selection of the power semiconductors.

The Power Transistor. The two mostcommon power semiconductorsused in the

S.M.P.S.arethe Bipolartransistorand thepowerMOSFET. The Bipolar transistor is normally limited to use at frequencies up to 30kHz, due to switching loss. However, it has very low on-state losses and is a relatively cheap device, making it the most suitable for lower frequency applications.The MOSFETis selected forhigher frequency operation because of its very fast switching speeds, resulting in low (frequency dependent) switching losses. The driving of the MOSFET is also far simpler and less expensive than thatrequired for theBipolar. However, the on-state losses of the MOSFET are far higher than the Bipolar, and they are also usually more expensive. The selection of which particular device to use is normally a compromise between the cost, and the performance required.

(i) Voltage limiting value:After deciding uponwhether to use a Bipolar orMOSFET,

the next step in deciding upon a suitable type is by the correct selection of the transistor voltage. For transformer coupled topologies, the maximum voltage developed across the device is normally at turn-off.This willbe either half,fullordoublethemagnitude ofthe inputsupply voltage, dependent upon the topology used. There may also be a significant voltage spike due to transformer leakage inductance that must be included. The transistor must safely withstand these worst case values withoutbreaking down. Hence, for a bipolar device, a suitably high V must be selected, and for a MOSFET, a suitably high V

.Atpresent 1750V is themaximum blocking voltage

BR(DSS)

available for power Bipolars,and a maximum of 1000V for power MOSFETs.

The selection guides assume that a rectified220V or 110V mainsinput is used. The maximum dc link voltages that will be produced for these conditions are 385V and 190V respectively.Thesevalues arethe inputvoltage levelsused to select the correct devicevoltage rating.

(ii) Current limiting value:The Bipolar device has a very low voltage drop across it

during conduction, which is relatively constant within the rated current range. Hence, for maximum utilisation of a bipolar transistor, it should be run close to its I This gives a good compromise between cost, drive requirements and switching. The maximum current for a particularthroughputpower iscalculated for each topology

Csat

ces(max)

value.

using simple equations. These equations are listed in the appropriatesections,andthe levelsobtainedused toselect a suitable Bipolar device.

The MOSFET device operatesdifferently from the bipolar in that the voltage developed across it (hence, transistor dissipation) is dependent upon the current flowingand the device "on-resistance" which is variable with temperature. Hence, the optimum MOSFET for a given converter can onlybechosen onthe basisthatthedevicemust notexceed a certain percentageof throughput (output) power.(In this selection a 5% loss in the MOSFET was assumed). A set of equations used to estimate the correct MOSFET R

DS(on)

valuefor a particular power level has been derived foreach topology. These equations are included in Appendix A at the end of the paper. The value of RDS(on) obtained was then used to select a suitable MOSFET device for each requirement.

NOTE! This method assumes negligible switching losses in the MOSFET. However for frequencies above 50kHz, switching losses become increasingly significant.

Rectifiers Two types of output rectifier are specified from the Philips

range. For very low output voltages below 10V it is necessarytohaveanextremely lowrectifierforwardvoltage drop,VF,inorder tokeep converterefficiency high.Schottky typesare specifiedhere,since theyhave verylowVFvalues (typically 0.5V). The Schottky also has negligible switching losses and can be used at very high frequencies. Unfortunately,theverylowVFoftheSchottkyis lostathigher reverseblocking voltages (typicallyabove100V ) and other diode types become more suitable. This means that the Schottky is normally reserved for useon outputsup to 20V or so.

Note. A suitable guideline in selecting the correct rectifier reversevoltageis toensurethedevice willblock 4to6times theoutputvoltageitisusedtoprovide(depends ontopology and whether rugged devices arebeing used).

For higher voltage outputs the most suitable rectifier is the fastrecovery epitaxial diode (FRED). This device has been optimised for use in high frequency rectification. Its characteristics include low VF(approx. 1V) with very fast and efficient switching characteristics. The FRED has reverse voltage blocking capabilities up to 800V. They are therefore suitable for use in outputs from 10 to 200V.

Therectifier devices specified in each selection guide were chosenashaving thecorrectvoltage limitingvalue andhigh enoughcurrent handling capability for theparticular output power specified. (A single outputis assumed).

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(5) Standard isolated topologies. (a) The Flyback converter.

Operation Of all the isolated converters, by far the simplest is the

single-ended flyback converter shown in Fig. 6.The use of a single transistor switch means that the transformer can only be driven unipolar (asymmetrical). This results in a largecore size. The flyback, which is anisolated version of the buck-boost, does not in truth contain a transformer but a coupled inductor arrangement. When the transistor is turned on, current builds up in the primary and energy is storedin the core, this energy is thenreleased to the output circuitthroughthe secondary when the switch is turned off. (A normal transformer such as the types used in the buck derived topologies couples the energy directly during transistor on-time, ideally storing noenergy).

TR1

Ip = Vin.ton/Lp

(discontinuous)

n:1

Isec = Idiode

leakage

inductance

spike

Vin

Discontinuous

Vin

Primary

current

Switch voltage

sec

Vce

Vds

Vin + Vo n1

ton toff

Fig. 6 Flyback converter circuit and waveforms.

The polarity of the windings is such that the output diode blocks during the transistor on time. When the transistor turns off, the secondary voltage reverses, maintaining a constant flux in the core and forcing secondary current to flow through the diode to the output load. The magnitude

of the peak secondary current is the peak primary current reached at transistor turn-off reflected through the turns ratio, thus maintaining a constant Ampere-turn balance.

The fact that all of the output power of the flyback has to be stored in thecore as 1/2LI2energy means thatthe core size and cost will be much greater than in the other topologies, where only the core excitation (magnetisation) energy, which is normallysmall, isstored. This, in addition to the initial poor unipolar core utilisation, means that the transformer bulk is one of the major drawbacks of the flyback converter.

Inorder toobtain sufficientlyhigh stored energy,theflyback primary inductance has to be significantly lower than required for a true transformer, since high peak currents areneeded. This is normally achieved bygappingthe core. Thegap reduces the inductance,andmost of thehighpeak energy is then stored in thegap, thusavoiding transformer saturation.

When the transistor turns off, the output voltage is back reflectedthroughthetransformertotheprimaryand inmany cases this can be nearly as high as the supply voltage. There is also a voltage spike at turn-off due to the stored energy in the transformer leakage inductance.This means that the transistor must be capable of blocking approximately twice the supply voltage plus the leakage spike. Hence, for a 220V ac application where the dc link canbe upto 385V, thetransistor voltage limitingvaluemust lie between 800 and 1000V.

Using a 1000V Bipolar transistor such as the BUT11A or BUW13Aallows a switching frequencyof 30kHz to be used at output powers up to 200Watts.

MOSFETs with 800V and 1000V limiting values can also beused,suchas theBUK456-800Awhichcansupply100W

atswitching frequencies anywhereupto 300kHz.Although the MOSFET can be switched much faster and has lower switching losses , it does suffer from significant on-state

losses, especially in the higher voltage devices when compared to the bipolars. Anoutline of suitable transistors and output rectifiers for different input and power levels using the flyback is given in Table 2.

Onewayofremovingthe transformerleakage voltagespike is to add a clamp winding as shown in Fig. 8. This allows the leakage energy to be returned to the input instead of stressing the transistor. The diode is always placed at the high voltage end so that the clamp winding capacitance does not interfere with the transistor turn-oncurrent spike, whichwould happen if the diode was connected toground. This clamp is optional and depends on the designer’s particular requirements.

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Advantages. The action of the flyback means that the secondary

inductance is in series with the output diode when current is delivered to the load; i.e driven from a current source. This means that no filter inductor is needed in the output circuit. Hence, each output requires only one diode and output filter capacitor. This means the flyback is the ideal choiceforgeneratinglow cost,multipleoutput supplies.The crossregulationobtained usingmultiple outputsis alsovery good (load changes on one output have little effect on the others) because of the absence ofthe output choke, which degrades this dynamic performance.

Theflybackisalso ideallysuited for generatinghigh voltage outputs.If a buck type LC filter was used to generate a high voltage, a very large inductancevalue wouldbe neededto reduce the ripple current levels sufficiently to achieve the continuous mode operationrequired. This restriction does not apply to theflyback, since itdoes notrequire anoutput inductance for successful operation.

Disadvantages. From the flyback waveforms in Fig. 6 it is clear that the

output capacitor is only supplied during the transistor off time. This means that the capacitor has to smooth a pulsatingoutput current which has higher peak values than the continuous output current that would be producedin a forward converter, for example. In order to achieve low outputripple, very large output capacitorsare needed,with very low equivalent series resistance (e.s.r). It can be shown that at the same frequency, an LC filter is approximately 8 times more effective at ripple reduction than a capacitor alone. Hence, flybacks have inherently much higher output ripples than other topologies. This, togetherwiththe higherpeak currents, largecapacitors and transformers, limits the flyback to lower output power applications in the 20 to 200W range. (It should be noted that at higher voltages, the required output voltage ripple magnitudes are not normally as stringent, and this means that the e.s.r requirement andhence capacitor sizewill not be as large as expected.)

Two transistor flyback. One possible solution to the 1000V transistor requirement

is the two transistor flyback version shown in Fig. 7. Both transistorsare switched simultaneously,andall waveforms are exactly the same, except that the voltageacross each transistor never exceeds the input voltage. The clamp winding is now redundant, sincethe two clamp diodes act to return leakage energy to the input. Two 400 or 500V devices can now be selected, which will have faster switching andlower conduction losses. The output power and switching frequencies can thus be significantly increased. The drawbacks of the two transistor version are the extra cost and more complex isolated base drive needed for the top floating transistor.

Vin

isolated

base drive

TR2

TR1

n : 1

Fig. 7 Two transistor Flyback.

Continuous Vs Discontinuous operation. As with the buck-boost, the flyback can operate in both

continuous and discontinuous modes. The waveforms in Fig. 6 show discontinuous mode operation. In discontinuous mode, thesecondary current fallsto zero in each switching period, and all of the energy is removed from the transformer. In continuous mode there is current flowing in the coupled inductor at all times, resulting in trapezoidal current waveforms. Themain plus of continuous mode is that thepeakcurrents flowing are only halfthat ofthe discontinuous for the same output power, hence, lower output ripple is possible. However, the core size is about 2 to 4 times larger in continuous mode to achieve the increased inductance needed to reduce the peakcurrents to achieve continuity.

A further disadvantage of continuous mode is that the closed loop is far more difficult to control than the discontinuousmode flyback.(Continuous mode contains a right hand plane zeroin itsopen loop frequency response, the discontinuous flyback does not. See Ref[2] for further explanation.) This means that much more time and effort is required for continuous mode to design the much more complicatedcompensationcomponentsneeded toachieve stability.

There is negligible turn-on dissipation in the transistor in discontinuous mode, whereas this dissipationcan befairly high in continuous mode, especially when the additional effects of the output diodereverse recovery current, which only occurs in the continuous case, is included. This normally means that a snubber must be added to protect the transistor against switch-on stresses.

Oneadvantage ofthe continuous mode isthatits open loop gain is independentof the output load i.e Voonly depends uponD and Vinas shown in the dc gainequation at the end of the section. Continuous mode has excellent open loop loadregulation, i.e varying the output load will not affect Vo. Discontinuous mode, on the other-hand, does have a dependency on theoutput, expressed as RLin the dc gain equation. Hence, discontinuous mode has a much poorer

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openloop loadregulation,i.e changing the outputwill affect Vo. This problem disappears, however, when the control loop is closed, and the load regulation problem is usually completely overcome.

Theuse ofcurrentmode control with discontinuous flyback (where both the primary current and output voltage are sensed and combined to control the duty cycle) produces

a much improved overall loop regulation, requiring less closed loop gain.

Although the discontinuous mode has the major disadvantageofveryhighpeakcurrentsand alarge output capacitor requirement, it ismuch easierto implement, and is by far the more common of the two methods used in present day designs.

Output power 50W 100W 200W

Line voltage, Vin 110V ac 220V ac 110V ac 220V ac 110V ac 220V ac

Transistor requirements

Max current 2.25A 1.2A 4A 2.5A 8A 4.4A Max voltage 400V 800V 400V 800V 400V 800V

Bipolar transistors.

TO-220 BUT11 BUX85 BUT12 BUT11A --- BUT12A

Isolated SOT-186 BUT11F BUX85F BUT12F BUT11AF --- BUT12AF

SOT-93 --- --- --- --- BUW13 ---

Isolated SOT-199 --- --- --- --- BUW13F ---

Power MOSFET

TO-220 BUK454-400B BUK454-800A BUK455-400B BUK456-800A --- ---

Isolated SOT-186 BUK444-400B BUK444-800A BUK445-400B BUK446-800A --- ---

SOT-93 --- --- --- --- BUK437-400B BUK438-800A

Output Rectifiers

O/P voltage

5V PBYR1635 PBYR2535CT ---

10V PBYR10100 PBYR20100CT PBYR30100PT

BYW29E-100/150/200 BYV79E-100/150/200 BYV42E-100/150/200

BYV72E-100/150/200

20V PBYR10100 PBYR10100 PBYR20100CT

BYW29E-100/150/200 BYW29E-100/150/200 BYV32E-100/150/200

50V BYV29-300 BYV29-300 BYV29-300

100V BYV29-500 BYV29-500 BYV29-500

Table 2. Recommended Power Semiconductors for single-ended flyback.

Note! The above values are for discontinuous mode. In continuous mode the peak transistor currents are approximately halved and the output power available is thus increased.

Converter efficiency, η = 80%; Max duty cycle, D

Max transistor voltage, V

Maxtransistorcurrent,IC; ID= 2

dc voltage gain:- (a) continuous (b) Discontinuous

Vin

= n

1− D

or V

= 2V

+ leakage spike

in(max)

η D

max

out

maxVmin

= 0.45

Vin

= D

RLT

2 L

√

Applications:- Lowest cost, multiple output supplies in the 20 to 200W range. E.g. mains input T.V. supplies, small

computer supplies, E.H.T. supplies.

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(b) The Forward converter.

Operation. The forward converter is also a single switch isolated

topology, and is shown inFig. 8. Thisis based onthe buck converter described earlier, with the addition of a transformer and another diode in the output circuit. The characteristic LC output filter isclearly present.

In contrast tothe flyback, the forwardconverter has a true transformer action, where energyis transferred directly to the output through the inductor during the transistor on-time. It can be seen that the polarity of the secondary winding is opposite to that of the flyback, hence allowing direct current flow through blocking diode D1. During the on-time, the current flowing causesenergy to bebuilt up in the output inductor L1. When the transistor turns off, the secondary voltage reverses, D1 goes from conducting to blocking mode and the freewheel diode D2 thenbecomes forwardbiased and provides a path for theinductor current to continue to flow. This allows the energy stored in L1 to be released into the load during the transistor off time.

The forward converter is always operated in continuous mode (in this case the output inductor current), since this producesverylowpeak inputand output currentsand small ripple components. Going into discontinuous mode would greatly increase these values, as well as increasing the amountof switching noisegenerated. No destabilising right hand plane zero occurs in the frequency response of the forwardin continuous mode (as with thebuck). See Ref[2]. This means thatthe control problemsthat existed with the continuous flyback are not present here. So there are no realadvantages to be gained by using discontinuous mode operation for the forward converter.

Advantages. As can beseen from the waveforms in Fig. 8, the inductor

current IL, which is also the output current, is always continuous. The magnitude of the ripple component, and hence the peak secondary current,depends uponthe size of the output inductor. Therefore, the ripple can be made relatively small compared to the output current, with the peak current minimised. This lowripple, continuousoutput currentis very easyto smooth, and so the requirements for the output capacitor size, e.s.r and peak current handling are far smaller than theyare for the flyback.

Since the transformer in this topology transfers energy directly there is negligible stored energy in the core compared to the flyback. However, there is a small magnetisation energy requiredto excite thecore, allowing it to become an energy transfer medium. This energy is very small and only a very small primary magnetisation current is needed. This means that a high primary

inductance is usually suitable, withno need forthe core air gap required in the flyback. Standard un-gapped ferrite cores with high permeabilities (2000-3000) are ideal for providing the high inductance required. Negligible energy storage means that the forward converter transformer is considerablysmaller than the flyback, and core loss is also muchsmaller forthe samethroughput power. However,the transformer is still operated asymmetrically, which means that power is only transferred during the switch on-time, and this poor utilisation means the transformer is still far bigger than in the symmetrical types.

The transistors have the same voltage rating as the discontinuous flyback (see disadvantages), but the peak current required for the same output power is halved, and this can be seen in the equations given for the forward converter. This, coupled with the smaller transformer and outputfilter capacitorrequirements means that the forward converter is suitable for use at higher output powers than the flyback can attain,and is normallydesigned tooperate inthe 100to 400Wrange. Suitable bipolars and MOSFETs for the forward converter arelisted in Table 3.

Vin

output

Inductor

current

Diode currents

TR1

current

TR1

voltage

Imag

Vce

Clamp winding

necessary

CONTROL

CIRCUIT

Id1

ton toff

Vin

n : 1

TR1

2Vin

Id2

Id3

Fig. 8 The Forward converter and waveforms.

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Disadvantages. Because of the unipolar switching action of the forward

converter, there is a major problem in how to remove the core magnetisation energy by the end of each switching cycle. If this did not happen, there would be a net dc flux build-up, leadingtocore saturation, and possible transistor destruction. This magnetisation energy is removed automatically by the push-pull action of the symmetrical types. In the flybackthis energy isdumped intothe load at transistor turn-off. However, there is no such path in the forward circuit.

Thispath is provided by adding an additionalreset winding of opposite polarity to the primary. A clampdiode is added, such that the magnetisation energyis returned to theinput supply during the transistor off time. The reset winding is woundbifilarwith the primary to ensure good coupling, and is normally made tohave thesame numberof turns as the primary. (The resetwinding wire gauge can be very small, since it only has to conduct the small magnetisation current.) The time for the magnetisation energy to fall to zero is thus the same duration as the transistor on-time. This means that the maximumtheoretical dutyratio of the forward converter is 0.5 and after taking into account switchingdelays,this fallsto 0.45.This limitedcontrol range isone of the drawbacksof usingthe forward converter. The waveform of the magnetisation current is also shown in Fig. 8. The clamp winding in the flyback is optional, but is always needed in the forwardfor correct operation.

Due to the presence of the reset winding, in order to maintain volt-sec balancewithin the transformer, the input voltage is back reflected to the primary from the clamp winding at transistor turn-off for the duration of the flow of the magnetisation resetcurrent throughD3. (There is also a voltage reversal across the secondary winding, and this is why diode D1 is added to block this voltage from the output circuit.) This means that the transistor must block two times Vin during switch-off. The voltage returnsto Vin after reset has finished, which means transistor turn-on losses will be smaller. Thetransistors musthave the same added burden of the voltage rating of the flyback, i.e 400V for 110V mains and 800Vfor 220V mains applications.

Output diode selection. The diodes in the output circuit both have to conduct the

full magnitude of the output current. They are also subject to abrupt changes in current, causing a reverse recovery spike, particularly in the freewheel diode, D2. This spike cancauseadditional turn-onswitching lossin thetransistor, possiblycausing devicefailure in the absence of snubbing. Thus, very high efficiency, fast trr diodes are required to minimise conduction losses and to reduce the reverse recovery spike. These requirements aremet withSchottky diodes for outputs up to 20V, and fast recovery epitaxial diodesforhigher voltageoutputs.It isnotnormalfor forward converter outputs to exceed100V becauseof the need for

a very large output choke, andflybacks arenormally used. Usually, both rectifiers areincluded ina singlepackage i.e a dual centre-tap arrangement. The Philips range of Schottkiesand FREDs which meet these requirements are also included in Table 3.

Two transistor forward. In order to avoid the use of higher voltage transistors, the

two transistor version of the forward can be used. This circuit, shown in Fig. 9, is very similar to the two transistor flyback and has the sameadvantages. Thevoltage across the transistor is again clamped to Vin, allowing the use of faster more efficient 400 or 500V devices for 220V mains applications. The magnetisation reset is achieved through the two clampdiodes, permitting the removalof the clamp winding.

Vin

isolated

base drive

TR2

n : 1

TR1

Fig. 9 Two transistor Forward.

The two transistor version is popular for off-line applications. It provides higher output powers and faster switching frequencies. The disadvantages are again the extracost of the higher componentcount, and the needfor an isolated drive for thetop transistor.

Although this converter has some drawbacks, andutilises the transformer poorly, itis avery popularselection forthe power range mentionedabove, and offerssimple drive for the single switch and cheap component costs. Multiple output types are very common. The output inductors are normally wound on a single core, which has the effect of improving dynamic cross regulation, and if designed correctly also reduces the output ripple magnitudes even further. The major advantage of the forward converter is thevery low output ripple thatcanbe achieved for relatively small sized LC components. This means that forward converters are normally used to generate lower voltage, high current multiple outputs such as 5, 12, 15, 28V from mains off-line applications, where lower ripple specifications are normally specified for the outputs. The high peak currents thatwould occur if a flyback was used would place an impossible burden on the smoothing capacitor.

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Output power 100W 200W 300W

Line voltage, Vin 110V ac 220V ac 110V ac 220V ac 110V ac 220V ac

Transistor requirements

Max current 2.25A 1.2A 4A 2.5A 6A 3.3A Max voltage 400V 800V 400V 800V 400V 800V

Bipolar transistors.

TO-220 BUT11 BUX85 BUT12 BUT11A --- BUT12A

Isolated SOT-186 BUT11F BUX85F BUT12F BUT11AF --- BUT12AF

SOT-93 --- --- --- --- BUW13 ---

Isolated SOT-199 --- --- --- --- BUW13F ---

Power MOSFET

TO-220 BUK454-400B BUK454-800A BUK455-400B BUK456-800A --- ---

Isolated SOT-186 BUK444-400B BUK444-800A BUK445-400B BUK446-800A --- ---

SOT-93 --- --- --- --- BUK437-400B BUK438-800A

Output Rectifiers (dual)

O/P voltage

5V PBYR2535CT --- ---

10V PBYR20100CT PBYR30100PT PBYR30100PT

20V PBYR20100CT PBYR20100CT PBYR20100CT 50V BYT28-300 BYT28-300 BYT28-300

BYV32E-100/150/200 BYV42E-100/150/200 BYV72E-100/150/200

BYV72E100/150/200

BYQ28E-100/150/200 BYV32E-100/150/200 BYV32E-100/150/200

Table 3. Recommended Power Semiconductors for single-ended forward.

Forward

Converter efficiency, η = 80%; Max duty cycle, D

Max transistor voltage, V

Maxtransistorcurrent,IC; ID=

dc voltage gain:-

Vin

or V

= 2V

η D

= nD

max

in(max)

out

maxVmin

= 0.45

Applications:- Low cost, low output ripple, multiple output supplies in the 50 to 400W range. E.g. small computer

supplies, DC/DC converters.

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Operation. To utilise the transformerflux swingfully, itis necessaryto

operate the core symmetrically as described earlier. This permits much smaller transformer sizes and provides higher output powers than possible with the single ended types. The symmetrical types always require an even numberof transistor switches. Oneof the best knownofthe symmetrical types is the push-pull converter shown in Fig. 10.

The primary is a centre-tapped arrangement and each transistor switch is driven alternately, driving the transformerin both directions.The push-pulltransformeris typically half the size of that for the single ended types, resulting in a more compact design. This push-pull action produces natural core resetting during each half cycle, hence no clamp winding is required. Power is transferred to the buck type output circuit during each transistor conduction period. Theduty ratio ofeach switch is usually less than 0.45. This provides enough dead time to avoid transistor cross conduction. The power can now be transferred to the output for up to 90% of the switching period, hence allowing greaterthroughput power thanwith the single-ended types. The push-pull configuration is normally used for output powers in the 100 to500W range.

Vin

TR1

TR2

T1 D1

n : 1

Fig. 10 Push-pull converter.

The bipolar switching action also means that the output circuitis actually operated at twice the switchingfrequency ofthepowertransistors, ascanbe seenfromthewaveforms inFig. 11. Therefore,the outputinductor and capacitor can be even smaller for similar output ripple levels. Push-pull converters are thus excellent for high power density, low ripple outputs.

Advantages. As stated, the push-pull offers very compact design of the

transformer and output filter, while producing very low output ripple. So if spaceis a premiumissue, the push-pull could be suitable. The controlof thepush-pull is similar to theforward,in thatit isagain basedon thecontinuousmode

buck. When closing the feedback control loop, compensation is relatively easy. For multiple outputs, the same recommendations given for the forward converter apply.

Clamp diodes are fitted across the transistors, as shown. Thisallows leakage and magnetisation energytobe simply channelled back to the supply, reducing stress on the switches and slightly improving efficiency.

The emitter or source of the power transistors are both at the same potential in the push-pull configuration, and are normally referenced to ground. This means that simple base drive can be used for both, and no costly isolating drive transformer is required. (Thisis not so for the bridge types which are discussed latter.)

Disadvantages. One of the main drawbacks of the push-pull converter is

the fact that each transistor must block twice the input voltage due to the doubling effect of the centre-tapped primary,even thoughtwo transistors are used. This occurs whenonetransistorisoffand theother isconducting. When both are off, each then blocks the supply voltage, this is shown in the waveforms in Fig. 11. This means that TWO expensive,less efficient800 to 1000V transistors would be required for a 220V off-line application. A selection of transistors and rectifiers suitable for the push-pull used in off-line applications is given in Table 4.

Afurther major problem with the push-pull is that it is prone to flux symmetry imbalance. If the flux swing in each half cycle is not exactly symmetrical, the volt-sec will not balance and this will result in transformer saturation, particularly for high input voltages. Symmetry imbalance can be caused by different characteristics in the two transistors such as storage time in a bipolar and different on-state losses.

The centre-tap arrangement alsomeans that extracopper isneededforthe primary, and very good coupling between the two halves is necessary to minimise possible leakage spikes. It should also benoted that if snubbersare used to protect the transistors, the design must be very precise since each tends to interact with the other. This is true for all symmetrically driven converters.

These disadvantages usually dictate that the push-pull is normally operated at lower voltage inputs such as 12, 28 or 48V. DC-DC converters found in the automotive and telecommunication industries are often push-pulldesigns. At these voltage levels, transformer saturation is easier to avoid.

Since the push-pull is commonly operated with low dc voltages,a selectionguide for suitablepower MOSFETs is alsoincluded for 48 and 96Vapplications, seenin Table 5.

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Current mode control. The introduction of current mode control circuits has also

benefited the push-pull type. In this type of control, the primary current is monitored, and any imbalance which occursiscorrectedona cycle by cycle basis by varying the duty cycle immediately. Current mode control completely

Transistor

currents

TR1

voltage

TR2 voltage

current

output

inductor

current

TR1

Vin

2Vin

ton

TR2

2Vin

Vin

ton

Fig. 11 Push Pull waveforms.

removes the symmetry imbalance problem, and the possibilities of saturation are minimised. This has meant thatpush-pull designshave becomemore popular inrecent years, with some designers even using them in off-line applications.

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Output power 100W 300W 500W

Line voltage, Vin 110V ac 220V ac 110V ac 220V ac 110V ac 220V ac

Transistor requirements

Max current 1.2A 0.6A 4.8A 3.0A 5.8A 3.1A Max voltage 400V 800V 400V 800V 400V 800V

Bipolar transistors.

TO-220 BUT11 BUX85 BUT12 BUT11A --- BUT12A

Isolated SOT-186 BUT11F BUX85F BUT12F BUT11AF --- BUT12AF

SOT-93 --- --- --- --- BUW13 ---

Isolated SOT-199 --- --- --- --- BUW13F ---

Power MOSFET

TO-220 BUK454-400B BUK454-800A BUK455-400B BUK456-800A --- ---

Isolated SOT-186 BUK444-400B BUK444-800A BUK445-400B BUK446-800A --- ---

SOT-93 --- --- --- --- BUK437-400B BUK438-800A

Output Rectifiers (dual)

O/P voltage

5V PBYR2535CT --- ---

10V PBYR20100CT PBYR30100PT --20V PBYR20100CT PBYR20100CT PBYR30100PT

50V BYT28-300 BYT28-300 BYV34-300

BYV32E-100/150/200 BYV72E-100/150/200 BYT230PI-200

BYQ28E-100/150/200 BYV32E-100/150/200 BYV42E-100/150/200

BYV72E-100/150/200

Table 4. Recommended Power Semiconductors for off-line Push-pull converter.

Output power 100W 200W 300W Line voltage, Vin 96V dc 48V dc 96V dc 48V dc 96V dc 48V dc Power MOSFET

TO-220 BUK455-400B BUK454-200A BUK457-400B BUK456-200B --- ---

Isolated SOT-186 BUK445-400B BUK444-200A BUK437-400B BUK436-200B --- ---

SOT-93 --- --- --- --- BUK437-400B ---

Table 5. Recommended power MOSFETs for lower input voltage push-pull.

Push-Pull converter.

Converter efficiency, η = 80%; Max duty cycle, D

Max transistor voltage, V

Maxtransistorcurrent,IC; ID=

dc voltage gain:-

or V

= 2V

= 2 nD

Vin

+ leakage spike.

in(max)

η D

maxVmin

= 0.9

max

out

Applications:- Compact design, very low output ripple supplies in the 100 to 500W range. More suited to low input

applications. E.g. battery, 28, 40V inputs, high current outputs. Telecommunication supplies.

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(d) The Half-Bridge.

Of all the symmetrical high power converters, the half-bridge converter shown inFig. 12 isthe mostpopular. It is also referred to as the single ended push-pull, and in principle is a balanced version of the forward converter. Again it is a derivative of the buck. The Half-Bridge has some key advantages over the push-pull, which usually makes it first choice for higher power applications in the 500 to 1000W range.

Operation. The two mains bulk capacitors C1 and C2 are connected

in series, and an artificial input voltage mid-point is provided, shown as point A in the diagram. The two transistorswitches aredriven alternately, andthis connects eachcapacitor across the single primary windingeach half cycle. Vin/2 is superimposed symmetrically across the primaryin a push-pull manner.Power is transferred directly to the output on each transistor conduction time and a maximum duty cycle of 90% is available (Some dead time is required to prevent transistor cross-conduction.) Since theprimary isdriven in both directions, (naturalreset) a full wave buck output filter (operating at twice the switching frequency) rather than a half wave filter is implemented. This again results in very efficient core utilisation. As can be seen in Fig. 13, the waveforms are identical to the push-pull, except that the voltageacross thetransistors is halved. (The device currentwould be higher for the same output power.)

Vin

TR1

isolated drive needed

TR2

Fig. 12 Half-Bridge converter.

Advantages. Since both transistors are effectively in series, they never

seegreater than thesupply voltage, Vin.When both are off, theirvoltages reach anequilibriumpoint of Vin/2.This is half the voltage rating of the push-pull (although double the

n : 1

current). This means that the half-bridge is particularly suited to high voltage inputs, such as off-lineapplications. Forexample, a 220V mains application can use twohigher speed, higher efficiency 450V transistors instead of the 800V types needed for a push-pull. This allows higher frequency operation.

Another major advantage over the push-pull is that the transformer saturation problems due to flux symmetry imbalance are not a problem. By using a small capacitor (less than 10µF) any dc build-up of flux in the transformer isblocked, and onlysymmetrical ac is drawnfrom the input.

The configuration of the half-bridge allowsclamp diodes to be added across the transistors, shown as D3 and D4 in Fig. 12. The leakage inductance and magnetisation energies are dumped straight back into the two input capacitors, protecting the transistors from dangerous transients and improving overall efficiency.

A less obvious exclusive advantage of the half-bridge is that the two series reservoir capacitors already exist, and this makes it ideal for implementing a voltage doubling circuit. This permits the useof either110V /220V mainsas selectable inputs to the supply.

The bridge circuits also have the same advantages over the single-ended types that the push-pull possesses, including excellent transformer utilisation, very low output ripple,andhighoutputpower capabilities.Thelimitingfactor inthemaximumoutput power availablefrom thehalf-bridge is the peak current handling capabilities of present day transistors. 1000W is typically the upper power limit. For higher output powers thefour switchfull bridgeis normally used.

Disadvantages. The need for two 50/60 Hz input capacitors is a drawback

because of their large size. The top transistor must also have isolated drive, since the gate / base is at a floating potential. Furthermore, if snubbers are used across the power transistors, great care must be taken in their design, since the symmetrical action means that they will interact with one another. The circuit cost and complexity have clearlyincreased,and this must be weighed upagainst the advantagesgained. Inmany cases, this normallyexcludes the use of the half-bridge at output power levels below 500W.

Suitable transistors and rectifiers for the half-bridge are given in Table 6.

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Transistor

currents

TR1

voltage

TR2 voltage

current

output

inductor

current

TR1

Vin

ton

TR2

Vin

ton

Fig. 13 Half-Bridge waveforms.

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Output power 300W 500W 750W Line voltage, Vin 110V ac 220V ac 110V ac 220V ac 110V ac 220V ac

Transistor requirements

Max current 4.9A 2.66A 11.7A 6.25A 17.5A 9.4A Max voltage 250V 450V 250V 450V 250V 450V

Bipolar transistors.

TO-220 BUT12 BUT11 --- --- --- ---

Isolated SOT-186 BUT12F BUT11F --- --- --- ---

SOT-93 --- --- BUW13 BUW13 --- BUW13

Isolated SOT-199 --- --- BUW13F BUW13F --- BUW13F

Power MOSFET

SOT-93 --- BUK437-500B --- --- --- ---

Output Rectifiers (dual)

O/P voltage

5V --- --- ---

10V PBYR30100PT --- --20V PBYR20100CT PBYR30100PT ---

50V BYT28-300 BYV34-300 BYV34-300

BYV72E-100/150/200 BYV32E-100/150/200 BYV42E-100/150/200

BYV72E-100/150/200

Table 6. Recommended Power Semiconductors for off-line Half-Bridge converter.

Half-Bridge converter.

Converter efficiency, η = 80%; Max duty cycle, D Max transistor voltage, V

Maxtransistorcurrent,IC; ID= 2

dc voltage gain:-

or V

= V

Vin

in(max)

η D

= nD

+ leakage spike.

max

out

maxVmin

= 0.9

Applications:- High power, up to 1000W. High current, very low output ripple outputs. Well suited for high input

voltage applications. E.g. 110, 220, 440V mains. E.g. Large computer supplies, Lab equipment supplies.

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(e) The Full-Bridge.

Outline. The Full-Bridge converter shown in Fig. 14 is a higher

power version of theHalf-Bridge, andprovides thehighest output power level of anyof the convertersdiscussed. The maximum current ratings of the power transistors will eventually determine the upperlimit ofthe outputpower of the half-bridge. These levels can be doubled by using the Full-Bridge, which is obtained by adding another two transistors and clamp diodes to the Half-Bridge arrangement.Thetransistors are drivenalternately inpairs, T1and T3, then T2 and T4. The transformer primary isnow subjectedtothe fullinputvoltage. Thecurrent levelsflowing are halved compared to the half-bridge for a given power level. Hence, the Full-Bridge will double the output power of the Half-Bridge using the same transistor types.

Thesecondary circuit operates inexactly the samemanner as the push-pull and half-bridge, also producing very low ripple outputs at very high current levels. Therefore, the waveforms for the Full-Bridge are identical to the Half-Bridge waveforms shown in Fig. 13, except for the voltage across the primary, which is effectively doubled (and switch currents halved). This is expressed in the dc gain and peak current equations, where the factor of two comes in, compared with the Half-Bridge.

Vin

Advantages. As stated,the Full-Bridge is ideal for thegeneration of very

high output powerlevels. The increased circuit complexity normally means that the Full-Bridge is reserved for applications with power output levels of 1kW and above. For such high power requirements, designers often select power Darlingtons, since theirsuperior currentratings and switching characteristics provide additional performance and in many cases amore cost effective design.

The Full-Bridge also has the advantage of only requiring one mains smoothing capacitor compared to two for the Half-Bridge, hence, saving space. Its other major advantages are the same asfor the Half-Bridge.

Disadvantages. Four transistors and clamp diodes are needed instead of

two for the other symmetrical types. Isolated drive for two floating potential transistors is now required. The Full-Bridge has the most complex andcostly designof any oftheconverters discussed,andshould onlybe usedwhere other types do not meet the requirements. Again, the four transistor snubbers (if required) must be implemented carefully to prevent interactions occurringbetween them.

Table 7 gives an outline of the Philips power semiconductors suitable for use with the Full-Bridge.

* Isolated drive required.

TR1

TR2

TR4

TR3

Fig. 14 The Full-Bridge converter.

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Output power 500W 1000W 2000W Line voltage, Vin 110V ac 220V ac 110V ac 220V ac 110V ac 220V ac

Transistor requirements

Max current 5.7A 3.1A 11.5A 6.25A 23.0A 12.5A Max voltage 250V 450V 250V 450V 250V 450V

Bipolar transistors.

TO-220 BUT12 BUT18 --- --- --- ---

Isolated SOT-186 BUT12F BUT18F --- --- --- ---

SOT-93 --- --- BUW13 BUW13 --- BUW13

Isolated SOT-199 --- --- BUW13F BUW13F --- BUW13F

Power MOSFET

SOT-93 --- BUK438-500B --- --- --- ---

Output Rectifiers (dual)

O/P voltage

5V --- --- --10V --- --- --20V PBYR30100PT --- ---

50V BYV34-300 BYV44-300 ---

Table 7. Recommended Power Semiconductors for the Full-Bridge converter.

BYV42E-100/150/200 BYV72E-100/150/200

Converter efficiency, η = 80%; Max duty cycle, D Max transistor voltage, V

Maxtransistorcurrent,IC; ID=

dc voltage gain:-

or V

= V

in(max)

= 2 nD

Vin

+ leakage spike.

η D

maxVmin

= 0.9

max

out

Applications:- Very high power, normally above 1000W. Very high current, very low ripple outputs. Well suited for

high input voltage applications. E.g. 110, 220, 440V mains. E.g. Computer Mainframe supplies, Large lab equipment

supplies, Telecomm systems.

Full-Bridge converter.

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Conclusion.

The 5 most common S.M.P.S. converter topologies, the flyback,forward,push-pull,half-bridgeand full-bridgetypes have been outlined. Each has its own particular operating characteristics and advantages, which makes it suited to particular applications.

Theconvertertopologyalsodefinesthevoltageand current requirements of the power transistors (either MOSFET or Bipolar).Simpleequations andcalculations used to outline the requirements ofthe transistors foreach topology have been presented.

The selection guide fortransistors andrectifiers atthe end ofeach topologysectionshows some of thePhilips devices which are ideal for usein S.M.P.S. applications.

References.

(1) Philips MOSFET Selection Guide For S.M.P.S. by M.J.Humphreys. Philips Power Semiconductor Applications group, Hazel Grove.

(2) Switch Mode Power Conversion - Basic theory and design by K.Kit.Sum. (Published by Marcel Dekker inc.1984)

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Appendix A. MOSFET throughput power calculations.

Assumptions made:The power loss (Watts) in the transistor due to on-state

losses is 5% of the total throughput (output) power. Switching losses in the transistor are negligible. N.B. At

frequencies significantly higher than 50kHz the switching losses may become important.

The device junction temperature, Tjis taken to be 125˚C. Theratio R

ds(125C˚)/Rds(25˚C)

MOSFET device. Table A1 gives the ratio for the relevant voltage limiting values.

The value of V A2.

s(min)

Device voltage limiting R

value. --------

100 1.74 200 1.91 400 1.98 500 2.01 800 2.11

1000 2.15

Table A1. On resistance ratio.

Main input Maximum dc link Minimum dc

voltage voltage link

220 / 240V ac 385V 200V 110 / 120V ac 190V 110V

Table A2. Max and Min dc link voltages for mains inputs.

isdependent on thevoltage of the

for each input value is given in Table

ds(125C)

ds(25C)

voltage

Using the following equations, for a given device with a known R topology can be calculated.

, the maximum throughput power in each

ds(125˚C)

Where:-

= Maximum throughput power.

th(max)

= maximum duty cycle.

τ = required transistor efficiency (0.05 ± 0.005)

max

Rds

s(min)

= R

(125˚C)

= minimum dc link voltage.

ds(25˚C)

x ratio.

Forward converter.

τ×V

×D

s(min)

th(max)

max

= 0.45

max

ds(125c)

Flyback Converter.

th(max)

3×τ×V

max

s(min)

4× R

= 0.45

ds(125c)

×D

max

Push Pull Converter.

τ×V

×D

s(min)

th(max)

= 0.9

max

ds(125c)

Half Bridge Converter.

τ×V

×D

s(min)

th(max)

4× R

= 0.9

max

ds(125c)

Full Bridge Converter.

τ×V

×D

s(min)

th(max)

2× R

= 0.9

max

ds(125c)

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2.1.2 The Power Supply Designer’s Guide to High Voltage Transistors

One of the most critical components in power switching converters is the high voltage transistor. Despite its wide usage, feedback from power supply designers suggests that there are several features of high voltage transistors which are generally not wellunderstood.

This section begins with a straightforward explanation of the key properties of high voltage transistors. This isdone byshowing howthe basic technology ofthe transistor leads toits voltage,current,power and secondbreakdownlimits. It is also made clear how deviations from conditions specifiedin thedata book will affect the performance of the transistor. The final section of the paper gives practical advicefordesignerson howcircuits mightbe optimisedand transistor failures avoided.

Introduction

A large amount of useful information about the characteristics of a given component is provided in the relevant data book. By using this information, a designer can usually be sure of choosing the optimum component for a particular application.

However,if aproblem ariseswiththe completedcircuit, and a more detailed analysis of the most critical components becomes necessary, the databook can become a source of frustration rather than practical assistance. In the data book,a component is often measured under averyspecific setofconditions.Very littleissaidabouthowthe component performance is affected if these conditions are not reproducedexactlywhenthecomponent isused ina circuit.

There are as many different sets of requirements for high voltage transistors asthere arecircuits whichmake use of them. Covering every possible drive and load condition in the device specification is an impossible task. There is therefore a real need for any designer using high voltage transistorstohave anunderstandingof howdeviations from theconditionsspecified inthe transistordatabook willaffect the electrical performance of the device, in particular its limiting values.

Feedbackfrom designersimpliesthatthis informationis not readily available. The intentionof thisreport istherefore to provide designers with the information they need in order tooptimise thereliability oftheir circuits.The characteristics ofhigh voltage transistors stem from their basic technology and so it is importantto begin with an overview of this.

HVT technology

Stripping away the encapsulationof the transistor reveals how the electrical connections are made (see Fig. 1). The collector is contacted through the back surface of the transistorchip,which issolderedtothenickel-plated copper lead frame. For Philips power transistors the lead frame andthe centreleg are formed froma single pieceof copper, and so the collector can be accessed through either the centre leg or any exposed part of the lead frame (eg the mounting base for TO-220 and SOT-93).

nickel-plated copper lead frame

passivated chip

aluminium wires

tinned copper leads

Base Collector Emitter

Fig. 1 High voltage transistor without the plastic case.

The emitter area ofthe transistor is contactedfrom thetop surfaceof the chip. A thinlayer of aluminium joinsall of the emitterareatoalarge bondpad.This bondpadisaluminium wire bonded to the emitter leg of the transistor when the transistor is assembled. The same method is used to contactthe base area of the chip. Fig. 2 shows the top view of a high voltage transistor chip in more detail.

Viewing thetop surface of the transistor chip,the base and emitter fingers are clearly visible. Around the periphery of the chip is the high voltageglass passivation. The purpose of this is explained later.

Taking across section through the transistor chip reveals its npn structure. A cross section which cuts one of the emitterfingersandtwoofthebasefingersisshowninFig. 3.

ultrasonic wire bonds

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Following the collector regionis the n+ backdiffusion. The n+back diffusion ensures a goodelectrical contactismade between the collector region and the lead frame/collector

base bond pad

emitter bond pad

leg, whilst also allowing the crystal to be thick enough to prevent it from cracking during processing and assembly. Thebottomsurfaceofthechip issoldered tothe leadframe.

Voltage limiting values

Part 1: Base shorted to emitter.

high voltage passivation

emitter fingersbase fingers

Fig. 2 High voltage transistor chip.

When the transistor is in its off state with a high voltage applied to the collector, the base collector junction is reverse biased by a very high voltage. The voltage supporting

depletionregion

extendsdeep intothe collector,

right up to the backdiffusion, as shown in Fig. 4.

base finger emitter finger base finger

Onthe topsurfaceofthe transistorare thealuminium tracks whichcontactthebase andemitterareas. Theemitterfinger isshownconnectedtoan n+region. Thisis theemitter area. Then+ denotes thatthis is very highly doped n type silicon. Surrounding the n+ emitter is the base, and as shown in Fig. 3 this is contacted by the base fingers, one on either side of theemitter. The p denotesthat thisis highly doped p type silicon.

Onthe other side of the base isthethick collectorn-region. The n- denotes that this is lightly dopedn type silicon.The collector region supports the transistor blocking voltage, and its thickness and resistivity must increase with the voltage rating of the device.

base finger emitter finger base finger

base

collector

back diffusion

solder lead frame

emitter

Fig. 3 Cross section of HVT.

base

emitter

n+ p

Depletion Region

collector

back diffusion

Fig. 4 Depletion region extends deep into the collector

during the off state.

Withthe base of the transistor short circuited to the emitter, or at a lower potential than the emitter, the voltage rating is governed by the voltage supporting capability of the reversebiased basecollector junction. This isthe transistor V

.Thebreakdownvoltage ofthereversebiasedbase

CESMmax

collectorjunctionisdetermined mainlybythe collectorwidth and resistivity as follows:

Figure5 shows the doping profile of the transistor.Notethe very high doping of the emitter and the back diffusion, the highdopingof the base and the low doping of the collector. Also shown in Fig. 5 is the electric field concentration throughout the depletion region for the case where the transistor is supporting its off state voltage. The electric field, E, is given bythe equation, E = -dV/dx, where -dV is the voltage drop in a distance dx. Rewriting this equation gives the voltage supported by the depletion region:

V =−

⌠

Edx

⌡

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avoided by the use ofa glasspassivation (see Fig. 6).The

Doping

E field

glasspassivation therefore allowsthe full voltage capability of the transistor to berealised.

EB C

Distance

Fig. 5 Doping profile and E field distribution.

This is the area underthe dotted line in Fig.5. During the off state, the peak electric field occurs at the

basecollectorjunctionasshownin Fig. 5.If theelectric field anywherein thetransistor exceeds 200 kVoltsper cm then avalanche breakdown occurs and the current which flows in the transistor is limited only by the surrounding circuitry. If the avalanche current is not limited to a very low value then the power rating of the transistor can easily be exceededandthetransistordestroyedas aresultofthermal breakdown. Thus the maximum allowable value ofelectric field is 200 kV/cm.

The gradient of the electric field, dE/dx, is proportional to charge densitywhich is in turn proportional to the level of doping. In the base, thegradient ofthe electricfield ishigh because of the high level of doping, and positivebecause the base is p type silicon. In the collector, the gradient of the electric field is low because of the low level of doping, and negative becausethe collector isn type silicon. In the backdiffused region, the gradient ofthe electric field is very highly negative because this is very highly doped n type silicon.

Increasing the voltage capability of the transistor can therefore be done by either increasing the resistivity (loweringthe level of doping) of thecollector region in order tomaintain a high electric field for the entire collectorwidth, or increasing the collector width itself. Both of these measures can be seen to work in principle because they increase the area under thedotted line in Fig. 5.

Thebreakdown voltage of the transistor, V

CESMmax

,is limited by the need to keep the peak electric field, E, below 200 kV/cm. Without special measures, the electric field would crowd at the edges of the transistor chip because of the surface irregularities. Thiswould limit breakdown voltages to considerably less than the full capability of the silicon. Crowding of the equipotential lines at the chip edges is

emitter

250V 600V

850V 1150V

special glass

base

Fig. 6 High voltage passivation.

Theglassusedisnegatively charged toinduce ap-channel underneath it. This ensures that the applied voltage is supported evenly over the width ofthe glass and does not crowd at any one point. High voltagebreakdown therefore occurs in the bulk of the transistor, at the base collector junction, and not at the edges of the crystal.

Exceeding the voltage rating of the transistor, even for a fraction of a second, must be avoided. High voltage breakdowneffectscanbe concentratedin a verysmall area of the transistor, and only a small amount of energy may damage the device. However,there is no danger in using thefull voltagecapability of thetransistor asthe limit under worstcase conditionsbecause thehigh voltagepassivation is extremely stable.

Part 2: Open circuit base.

With the base of the transistor open circuit the voltage capability is much lower. This is the V and it is typically just less than half of the V The reason for the lower voltage capability under open

CEOmax

of the device

rating.

CESMmax

circuit base conditions is asfollows: As the collector emitter voltage of the transistor rises, the

peakelectricfield locatedat thebasecollector junctionrises too. Above a peak E field value of 100 kV/cm there is an appreciable leakage current being generated.

In the previous case, with the base contact shortcircuited to the emitter,or held at a lower potential thanthe emitter, any holes which are generated drift from the edge of the depletion region towards the base contact where they are extracted.However, withthe base contact opencircuit, the holes generated diffuse from the edge of the depletion region towards the emitter where they effectively act as basecurrent.Thiscauses theemitterto injectelectrons into thebase, whichdiffuse towards the collector.Thus there is a flow of electrons from the emitter to the collector.

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The high electric field in the collector accelerates the electronstothe level where some have sufficient energy to produce more hole electron pairs through their collisions with the lattice. The current generated in this way adds to theleakagecurrent.Thus withthebase contactopen circuit the emitter becomes active and provides the system with

gain

, multiplying the leakage current and consequently

reducing the breakdown voltage. For a given transistor the gain of the system is dependant

on two things. Firstlyit is dependant on theprobability that a hole leaving thedepletion regionwill reachthe emitter.If the base isopen circuit andno recombination occurs then this probability is 1. If the base is not open circuit, and instead a potential below V

is applied, then there is a

BEon

chance that a hole leaving the depletion region will be extracted atthe base contact. As the voltage on the base contact is made less positive the probability of holes reaching the emitter is reduced.

Secondly, the gain is dependant on the probability of electronsleavingthe emitter, diffusing across the base and being accelerated by the high field in the collector to the level where they are able toproduce ahole electron pair in one of their collisions with thelattice. This depends on the electric field strength which is in turn dependant on the collector voltage.

Thusforagiven voltageatthe basethereis acorresponding maximum collector voltage before breakdown will occur. With the base contact shorted to the emitter, or at a lower potentialthan the emitter, the full breakdown voltage ofthe transistorisachieved(V circuit, or at a

higher

breakdown voltage is lower (V the emitter is active and it provides the breakdown

).Withthebasecontactopen

CESMmax

potential than the emitter, the

) because in this case

CEOmax

mechanism with gain. With the base connected to the emitter by a non zero

impedance, the breakdown voltage will be somewhere between the V

CESMmax

and the V

. A low impedance

CEOmax

approximates to the shorted base, ’zero gain’, case and a highimpedanceapproximates tothe openbase, ’highgain’, case. With a base emitter impedance of 47 Ω and no externally applied base voltage, the breakdown voltage is typically 10% higher than theV

CEOmax

Current limiting values

The maximum allowed DC current is limited by the size of the bond wires to the baseand emitter. Exceeding the DC limiting values I time, may blowthese bond wires. If the current pulses are short and of alow duty cycle then valuesgreatly in excess of the DC values areallowed. The I are recommendations for peak current values. For a duty cycle of 0.01 and a pulse width of 10ms these values will typically be double the DCvalues.

Cmax

and I

, for any significant length of

Bmax

and I

CMmax

BMmax

ratings

If the pulses are shorter than 10ms then even the recommended peak values can be exceeded under worst case conditions. However, it should be noted that combinations of high collector current and high collector voltagecanleadtofailureby secondbreakdown(discussed later). As the collector current is increased, the collector voltage required to trigger second breakdown drops, and soallowinglarge collector current spikes increasesthe risk of failure by second breakdown. It istherefore advised that the peak values givenin thedata bookare used as design limits in order to maximisethe component reliability.

In emitter drive circuits, the peak reverse base current is equal to the peak collector current. The pulse widths and duty cycles involvedare small, and this modeof operation iswithin the capability of all Philips high voltage transistors.

Power limiting value

The P achievable parameter because in practice it is obtainable onlyif the mountingbasetemperature can be held to25 ˚C. In practice, themaximum power dissipation capability of a givendevice is limited by the heatsink size and theambient temperature.Themaximumpowerdissipation capabilityfor a particular circuit can becalculated as follows;

jmax

sheet. The value normally quoted is 150 ˚C. T ambient temperature around the device heatsink.A typical valuein practice could be 65˚C. R resistance given inthe data sheet, but to obtain a value of junction to resistance of the mica spacer (if used), heatsink and heatsink compound should be added to this.

Themaximumpower whichcanbedissipatedunderagiven set of circuit conditions iscalculated using;

For a BUT11AF, in an ambient temperature of 65 ˚C, mounted on a 10 K/W heatsink with heatsink compound, this gives;

andhencethemaximumpowercapable ofbeing dissipated under these conditions is;

Exceeding the maximum junction temperature,T recommended. All of the quality andreliability work carried out on the device is based on the maximum junction temperaturequoted indata. IfT then the reliability of the device is no longer guaranteed.

given in device data is not generally an

totmax

isthemaximum junction temperature given in the data

is the

amb

isthedevice thermal

thj-mb

ambient

thj-a

thermal resistance, R

max

= (T

jmax-Tamb

)/R

thj-a

, the thermal

thj-a

= 3.95 K/W + 10 K/W = 13.95 K/W

= (150-65)/13.95 = 6 W

max

isexceeded inthe circuit

jmax

, is not

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