Datasheet LTC2978 Datasheet (LINEAR TECHNOLOGY)

LINEAR TECHNOLOGY

LINEAR TECHNOLOGY

LINEAR TECHNOLOGY

SEPTEMBER 2009 VOLUME XIX NUMBER 3

V

PWR

V

IN_EN

V

DD33

V

DD33

V

DD25

SDA
SCL
ALERTB
CONTROL0
CONTROL1
WDI/RESET
FAULTB00
FAULTB01
FAULTB10
FAULTB11
SHARE_CLK

V

IN_SNS

V

DACP0

V

SENSEP0

V

DACM0

V

SENSEM0

V

OUT_EN0

ASEL0
ASEL1

PWRGD

REFP

REFM

LTC2978*

PMBus

INTERFACE

TO INTEMEDIATE
BUS CONVERTER

ENABLE

V

IN

4.5V < V

IBUS

< 15V

GNDWP

V

DD33

TO µP

0.1µF

0.1µF

0.1µF

TO/FROM OTHER

CONTROLLERS

*SOME DETAILS OMITTED FOR CLARITY
ONLY ONE OF EIGHT CHANNELS SHOWN

V

IN

V

OUT

R20

R30

R10

RUN/SS

SGND

V

FB

GND

DC/DC

CONVERTER

LOAD

IN THIS ISSUE…

COVER ARTICLE
Power Management IC Digitally

Monitors and Controls Eight Supplies

...........................................................1

Andrew Gardner

Linear in the News… ...........................2

DESIGN FEATURES
PD Controller ICs with Integrated

Flyback or Forward Controllers

Meet Demands of 25.5W PoE+ .............6

Ryan Huff

Surge Stopper IC Simplifies Design of
Intrinsic Safety Barrier for Electronics
Destined for Hazardous Environments

...........................................................9

Murphy Pickard, Hach Co.

Consider New Precision Amplifiers for
Updated Industrial Equipment Designs

.........................................................16

Brian Black

Analog VGA Simplifies Design
and Outperforms Competing

Gain Control Methods ........................19

Walter Strifler

Power Management IC Digitally Monitors and Controls Eight Supplies

by Andrew Gardner

Introduction

Today’s high reliability systems require
complex digital power management
solutions to sequence, supervise,
monitor and margin a large number of
voltage rails. Indeed, it is not unusual
for a single application board to have
dozens of rails, each with its own
unique requirements. Typically the
power management solutions for these
systems require that several discrete
devices controlled by an FPGA or a
microcontroller are sprinkled around
the board in order to sequence, super­vise, monitor and margin the power
supply array. In this scheme, signifi-

cant time is required to develop the
necessary firmware, and the tendency
to underestimate the complexity and
duration of that task is well known.

The LTC®2978 octal PMBus power
supply monitor and controller with
EEPROM offers power supply system
designers an integrated, modular
solution that reduces debugging time
and effort over microcontroller solu­tions. The LTC2978 can sequence
on, sequence off, monitor, supervise,
margin and trim up to eight power
supplies. Multiple LTC2978s can be

continued on page 3

Accurate Silicon Oscillator Reduces
Overall System Power Consumption

.........................................................22

Albert Huntington

Easy Multivoltage Layout with
Complete Dual and Triple Output
Point-of-Load µModule® Regulators

in 15mm × 15mm Packages...............24

Eddie Beville and Alan Chern

DESIGN IDEAS

....................................................27–40

(complete list on page 27)

New Device Cameos ...........................41

Design Tools ......................................43

Sales Offices .....................................44

Figure 1. Octal power supply controller with PMBus communication. One channel is shown.

L

, Linear Express, Linear Technology, LT, LTC, LTM, BodeCAD, Burst Mode, FilterCAD, LTspice, OPTI-LOOP, Over-The­Top, PolyPhase, SwitcherCAD, µModule and the Linear logo are registered trademarks of Linear Technology Corporation.
Adaptive Power, Bat-Track, C-Load, DirectSense, Easy Drive, FilterView, Hot Swap, LTBiCMOS, LTCMOS, LinearView,
Micropower SwitcherCAD, Multimode Dimming, No Latency ∆Σ, No Latency Delta-Sigma, No R
PanelProtect, PowerPath, PowerSOT, SafeSlot, SmartStart, SNEAK-A-BIT, SoftSpan, Stage Shedding, Super Burst, ThinSOT,
TimerBlox, Triple Mode, True Color PWM, UltraFast and VLDO are trademarks of Linear Technology Corporation. All other
trademarks are the property of their respective owners.

, Operational Filter,

SENSE

L LINEAR IN THE NEWS

Linear in the News…

On the Road in China

For the past several years, Linear has participated in the
IIC China Conference. Traditionally, this has been an
opportunity for major electronics companies to showcase
their product capabilities in the major Chinese centers in
Beijing, Shenzhen and Shanghai. This year, for the first
time, the IIC is also holding a trade show in the remote
area of Wuhan, since this area is a growing technology
center, and Linear will participate. This follows Linear’s
participation at the IIC Conference in February/March in
Shenzhen, Beijing and Xian.

At the IIC in Wuhan on September 14–15, Linear will
focus on products for the automotive, industrial and telecom
markets. Some of the product highlights include:

q

LED drivers for a range of applications

q

µModule receiver products for cellular basestations

q

DC/DC µModule regulators, providing easy-to-

implement power solutions

q

Battery stack monitors for hybrid and electric

vehicles

At its booth, Linear will run a demo of the LTC6802
Battery Stack Monitor, showing automotive electronics
designers how to use the device to precisely monitor
every cell in long strings of series-connected lithium-ion
batteries.

Linear Debuts Isolated µModule
Transceiver with Power

Leveraging its experience in µModule technology, Linear
has just announced the first product in a new family of
galvanically isolated µModule products aimed for use in
industrial networks. The LTM®2881 is a complete isolated
RS485/RS422 solution and the first transceiver product
to utilize Linear’s isolator µModule technology, integrating
a 2500V
transceiver and all necessary power components into low
profile LGA and BGA packages. No external
components are required, eliminating issues
with sourcing transformers. In addition, the
LTM2881’s 1W DC/DC converter provides
surplus current for powering external ICs and
LEDs via a 5V regulated output. The LTM2881
exhibits high common mode transient im­munity, >30kV/µs, allowing the LTM2881 to
continue communicating, rather than merely
holding a data state, through severe transient
events.

The features of the LTM2881 make it suit­able for a wide range of applications, including
breaking ground loops, working with large com­mon mode voltages and when using multiple
unterminated line taps. Integrated selectable
termination allows cables to be properly termi­nated to avoid signal reflections and distorted

galvanic isolation barrier, a high performance

RMS

waveforms, with the flexibility to add or remove termination
anywhere onto the bus via a software switch. Users will
appreciate how the self-powered LTM2881 takes many
precautions to guarantee safe and reliable communica­tions in RS485 or RS422 systems.

Solar Power Battery Charger
Improves Panel Efficiency

For a given amount of light energy, a solar panel has a
certain output voltage for peak output power production.
Bypass diodes inside a panel can create complex power
versus current characteristics that are not easily optimized
when partial shading exists on the panel. However, virtu­ally all of the 12V system solar panels currently on the
market that are specified with maximum output power
less than 25W are constructed from a simple series cell
arrangement with no bypass diodes. This type of arrange­ment yields peak output power within a narrow band of
panel output voltages, regardless of lighting conditions.
Peak power in excess of 95% may be produced from panel
voltages of 12.5V–18.5V, depending on the characteristics
of the panel.

ger, the LT®3652, designed to provide an elegant electrical
operating characteristic while extracting the maximum
available power from the solar panel. The LT3652 employs
a simple but innovative input voltage regulation loop,
which controls charge current to hold the input voltage at
a programmed level. This input regulation loop maintains
the panel at the output voltage corresponding to the peak
output power point for the particular solar panel used. The
specific desired peak-power voltage is programmed via a
resistor divider. This method yields charging efficiencies
virtually the same as more costly maximum peak power
tracking (MPPT) solar charging techniques.

Linear has just announced a solar power battery char-

L

2

2

Linear Technology Magazine • September 2009

DESIGN FEATURES L

LTC2978 #1

SHARE_CLK

FAULT

LTC2978 #2

SHARE_CLK

FAULT

LTC2978 #N

SHARE_CLK

FAULT

•

•

•

POWER SUPPLY

ARRAY

V

OUTn

500mV/DIV

200ms/DIVSTART UP
8 SUPPLIES
IN ANY ORDER

SHUT DOWN

8 SUPPLIES

IN ANY ORDER

INDIVIDUAL MARGINING

FOR 8 SUPPLIES

LTC2978, continued from page 1

easily cascaded using the 1-wire share­clock bus and one or more bidirectional
fault pins (Figure 1 shows a typical
application).

In addition, the LTC2978 uses a
protected block of nonvolatile memory
to record system voltage and fault
information in the event of a critical
system failure. Preserving critical
system data in nonvolatile memory
allows users to identify a failing volt­age rail and isolate the cause of board
failures during system development,
test debug or failure analysis.

A free, downloadable graphical
PC interface is available to facilitate
interaction with the part in design
and testing. The LTC2978 utilizes the
industry standard PMBus command
protocol in order to simplify firmware
development. The LTC2978’s most
important feature, though, is that its
precision integrated reference and 15­bit ∆Σ ADC delivers ±0.25% absolute
accuracy when measuring or adjusting
power supply voltages.

Improve Manufacturing Yields
with Precision Margin Testing

Margin testing of system voltages is
an effective means of weeding out
premature failures in high reliability
systems. Typically, voltages are mar­gined at least ±5% in order to guarantee

Linear Technology Magazine • September 2009

Figure 3. Multiple LTC2978s can be cascaded using only two connections.

Figure 2. The LTC2978 offers flexible
sequencing and precision margining.

that the system under test is robust
enough to operate reliably in the field.
Depending on system tolerances,
however, this approach can lead to
excessive test fallout. Many of these
test rejects might have been avoided
if the tolerances of the supply voltages
in question were tighter.

With its precision reference, mul­tiplexed 15-bit ∆Σ ADC, eight margin
DACs and integrated servo algorithm,
the LTC2978 offers a relatively easy­to-use, yet powerful, solution to this
problem (see Figure 4 for the LTC2978
bock diagram). By simply writing an
I2C command to either trim or margin
to a specific voltage, the LTC2978
adjusts the DC/DC point-of-load con­verter within the prescribed software

and hardware limits to deliver the

commanded output voltage to ±0.25%
absolute accuracy.

The margin DAC outputs are con­nected to the feedback nodes or trim
inputs of the DC/DC POL converters
via a resistor. The value of this resistor
sets a limit on the range over which
the output voltage can be margined,
an important limitation for power sup­plies under software control. Another
significant benefit of the 10-bit margin
DACs is that they enable very fine
resolution when margining voltages.
This makes it possible to extract useful
data from failure testing, as opposed
to a trashcan full of failed, but not well
understood, boards.

Flexible Power Sequencing
and Fault Management

Many traditional power sequencing
solutions rely on comparators and
daisy-chained PCB connections. While
relatively easy to implement for a hand­ful of supplies, this approach quickly
becomes complicated as the number
of voltage rails grows, and is relatively
inflexible in the face of specification
changes. It’s also extremely difficult
to implement turn-off sequencing with
this type of approach.

The LTC2978 makes sequencing
easy for any number of supplies. By
using a time-based algorithm, users
can dynamically sequence on and
sequence off in any order (see Fig­ure 2). Sequencing across multiple
LTC2978s is also possible using the
1-wire share-clock bus and one or
more of the bidirectional fault pins
(see Figure 3). This approach greatly
simplifies system design because
channels can be sequenced in any
order, regardless of which LTC2978
provides control. Additional LTC2978s
can also be added later without hav­ing to worry about system constraints
such as a limited supply of daughter
card connector pins.

On sequencing can be triggered in
response to a variety of conditions. For
example, the LTC2978s can auto-se­quence when the downstream DC/DC
POL converters’ intermediate bus
voltage exceeds a particular turn-on
voltage. Alternatively, on sequencing

3

L DESIGN FEATURES

15

3V REGULATOR

INTERNAL

TEMP

SENSOR

REFERENCE

1.232V
(TYP)

MASKING

CLOCK

GENERATION

OSCILLATOR

UVLO

V

DD

OPEN-DRAIN

OUTPUT

EEPROM

NONVOLATILE MEMORY

RAM

ADC_RESULTS

MONITOR LIMITS

SERVO TARGETS

PMBus
INTERFACE
(400kHz I

2

C

COMPATIBLE)

CONTROLLER

PMBus ALGORITHM

FAULT PROCESSOR

WATCHDOG
SEQUENCER

V

IN

V

DD

V

SENSEP0

V

SENSEM0

V

OUT

V

PWR

17

2.5V REGULATOR

V

IN

V

OUT

V

DD33

36

V

SENSEP0

37

V

SENSEM0

V

SENSEP1

V

SENSEM1

V

SENSEP2

V

SENSEM2

V

SENSEP3

V

SENSEM3

V

SENSEP4

V

SENSEM4

V

SENSEP5

V

SENSEM5

V

SENSEP6

V

SENSEM6

2

V

SENSEP7

3

V

SENSEM7

V

DACP0

V

DACP1

V

DACP2

V

DACP3

V

DACP4

V

DACP5

V

DACP6

V

DACP7

V

DACM0

V

DACM1

V

DACM2

V

DACM3

V

DACM4

V

DACM5

V

DACM6

V

DACM7

4

V

OUT_EN0

5

V

OUT_EN1

6

V

OUT_EN2

7

V

OUT_EN3

8

V

OUT_EN4

9

V

OUT_EN5

10

V

OUT_EN6

11

V

OUT_EN7

12

V

IN_EN

18

V

DD25

65

GND

28

SCL

27

SDA

29

ALERTB

32

ASEL0

33

ASEL1

30

CONTROL0

19

WP

31

CONTROL1

WDI/RESET

22

23

FAULTB00

24

FAULTB01

25

FAULTB10

26

FAULTB11

20

21

SHARE_CLK

PWRGD

16

V

DD33

14

V

IN_SNS

REFP

REFM

3R

R

V

SENSEP1

V

SENSEM1

V

SENSEP2

V

SENSEM2

V

SENSEP3

V

SENSEM3

V

SENSEP4

V

SENSEM4

V

SENSEP5

V

SENSEM5

V

SENSEP6

V

SENSEM6

V

SENSEP7

V

SENSEM7

15-BIT

∆∑ ADC

ADC

CLOCKS

V

DD

+
–

+
–

MUX

34

35

10-BIT

VDAC

+

–

+
+

–
–

+

–

SC

CMP0

CMP0

VBUF0

IDAC0

10 BITS

42

43

46

47

48

49

52

53

62

63

64

1

39

40

44

50

55

56

60

38

41

45

51

54

57

58

61

59

4

Figure 4. Block diagram of the LTC2978

Linear Technology Magazine • September 2009

DESIGN FEATURES L

can initiate in response to the rising- or
falling-edge of the control pin input.
Sequencing can also be initiated by a
simple I2C command. The LTC2978
supports any combination of these
conditions.

The bidirectional fault pins can
be used for various fault response
dependencies between channels.
For instance, on sequencing can be
aborted for one or more channels in the
event of short-circuit. Once a rail has
powered-up, the undervoltage supervi­sor function is enabled (the overvoltage
function is always enabled). The
overvoltage and undervoltage thresh­olds and response times of the voltage
supervisors are all programmable. In
addition, input voltage and tempera­ture are also monitored. If any of these
quantities exceed their over- or under­value limits, the customer can select
from a rich variety of fault responses.
Examples include immediate latchoff,
deglitched latchoff, and latchoff with
retry.

An integrated watchdog timer is
available for supervising external
microcontrollers. Two timeout inter­vals are available: the first watchdog
interval and subsequent intervals.
This makes it possible to specify a
longer timeout interval for the micro
just after the assertion of the power
good signal. In the event of a watchdog
fault, the LTC2978 can be configured
to reset the micro for a predetermined
amount of time before reasserting the
power good output.

Multifaceted Telemetry

The LTC2978 serves up a variety of
telemetry data in its registers. The
multiplexed, 15-bit ∆Σ ADC monitors
input and output voltages and on-chip
temperature, storing minimum and
maximum values for all voltages and
temperature readings. In addition, the
ADC inputs for odd-numbered output
channels can be reconfigured to mea­sure sense resistor voltages. In this
mode, the ∆Σ ADC can resolve voltages

down to 15.3µV, which is invaluable
when attempting to measure current
with inductor DCR circuits.

Although the LTC2978 can be
directly powered from a 3.0V to 3.6V
supply, the ADC is capable of accepting
input voltages of up to 6V—no need
to worry about body diodes or exotic
standby supply voltages. The LTC2978
can also run off of a 4.5V to 15V input
supply using its internal regulator.
A separate high voltage (15V max)
sense input is provided for measur­ing the input supply voltage for the
DC/DC POL converters controlled by
the LTC2978.

Black Box Data Recorder

In the event a channel is disabled in
response to a fault, the LTC2978’s
data log can be dumped into protected
EEPROM. This 255-byte block of data
is held in NVM until it is cleared with
an I2C command. The data block
contains output and input voltages
and temperature data for the 500ms
preceding the fault as well as the cor­responding minimum and maximum
values. Status register values and total
up time since the last system reset are
also stored in the log.

Figure 5 shows the data log con­tents viewed in the PC-based LTC2978
interface. In this way, the LTC2978
provides a complete snapshot of the
state of the power system immediately
preceding the critical fault, thus mak­ing it possible to isolate the source of
the fault well after the fact. This is an
invaluable feature for debugging both
prerelease characterization or in-field
failures in high reliability systems.

Figure 5. The LTC2978 comes with free software that allows easy data monitoring and
cofiguration. The data log shows monitor readings just before a failure for debugging analysis.

Linear Technology Magazine • September 2009

Graphical User Interface
and PMBus

Linear T echnology’s easy-to-use
PC-based graphical user interface
(GUI) allows users to configure the
LTC2978 via a USB interface and a
dongle card. The GUI, which is free
and downloadable, takes much of the
coding out of the development process
and improves time-to-market by al­lowing the designer to configure all
device parameters within an intuitive
framework. Once the device configura-

continued on page 18

5

L DESIGN FEATURES

100k

20Ω

3.01k
1%

27.4k
1%

10µF

39k

1µF

BAS21

1.2k

38.3k

33pF

1.5nF

10µF
100V

0.1µF

t

ON

12k

PGDLYV

NEG

SYNC

R

CLASS

SHDN

V

CMP

R

CMP

ENDLY OSC SFST

LTC4269-1

GND

UVLO

V

PORTP

V

CC

T2P

T2P

FB

C

CMP

+

+

V

PORTN

383k
1%

3.01k
1%

10k

1nF

3.3nF

•

•

•

•

33mΩ
1%

FDS2582

10k

15Ω

150Ω

PE-68386

BAT54

100Ω

2200pF

MMBT3906 MMBT3904

1µF

1µF
16V

T1

PA2369NL

SENSE

–

SENSE

+

SG

PG

L1

0.18µH

22pF

100µF

5.1Ω

•

FDS8880

47µF

5V
5A

+

SMAJ58A

30.9Ω

24k

107k

10k

S1B

B1100 s 8 PLCS

2.2µF
100V

10µH

DO1608C-103

0.1µF
100V

36V
PDZ36B

BSS63LT1

V

PORTP

48V

AUXILIARY

POWER

–54V FROM

DATA PAIR

–54V FROM

SPARE PAIR

+

–

PD Controller ICs with Integrated Flyback or Forward Controllers Meet Demands of 25.5W PoE+

Introduction

The IEEE 802.3af Power over Ether­net (PoE) standard allows a powered
device (PD), such as an internet
protocol (IP) telephone, to draw up to

12.95W from an Ethernet cable. When
the 802.3af standard was drafted,

12.95W appeared sufficient to cover
the immediately imaginable range of
PD products (primarily IP phones).
Of course, application developers
are always far more innovative than
standards committees anticipate, so
new power-hungry applications for
PoE immediately started to appear,
such as dual-radio IEEE 802.11a/g
and 802.11n wireless access points,
security cameras with pan/tilt/zoom
motors, and color LCD IP video
phones. 12.95W was suddenly not
enough. The IEEE committee re­sponded with the 802.3at standard,
which raises the available PD power

to 25.5W. The new “at” standard, com­monly referred to as PoE+, also adds
a “handshaking” communications
requirement between PDs and power
sourcing equipment (PSEs), while al­lowing backward compatibility with
the legacy “af” standard.

New power control ICs are required
to take advantage of these expanded
requirements. The DC/DC conversion
and control schemes used for legacy
“af” PDs are not optimized for the in­creased power capability and feature
requirements of PoE+. For instance,
in both standards the 37V to 57V PoE
voltage is converted to lower voltages
that digital circuitry can tolerate.
This DC/DC conversion is handled
in the lower power 12.95W standard
with a conventionally rectified (i.e.,
diode rectified) flyback converter. The
higher power 25.5W standard is better

by Ryan Huff

served by a synchronously rectified
(i.e. MOSFET rectified) flyback or a
forward power supply topology.

To meet the new performance
requirements of PoE+, including
handshaking, Linear Technology offers
a new family of PD controller ICs that
integrate a front-end PD controller with
a high performance synchronously
rectified flyback (LTC4269-1) or a
forward (LTC4269-2) power supply
controller.

Features

Both parts combine a PD control­ler—which includes the handshaking
circuitry, Hot Swap™ FET, and input
protection—with a DC/DC power
supply controller. While the power
supply sections of the two parts are
very different, the PD controller in
both is identical.

6

Figure 1. LTC4269-1-based synchronous flyback converter

Linear Technology Magazine • September 2009

+

T2P

T2P

V

NEG

V

PORTN

SHDN

SMAJ58A

R

CLASS

V

PORTP

PGND GND BLANK DELAY

82k

30.9Ω24k

158k 332k

133Ω

BAS516

BAS516

PA2431NL

BAS516

10k

IRF6217

FDS8880

FDS8880

5.1Ω

158k

22.1k

33k

1.5k

50mΩ

2k

5.1Ω

1.2k

TLV431A

PS2801-1-L

V

CC

11.3k

3.65k

22k

0.22µF

0.1µF

R

OSC

V

REF

FB

COMP

I

SENSE

OC

SS_MAXDC

FDS2582

SD_V

SEC

V

IN

S

OUT

LTC4269-2

0.1µF

18V

PDZ18B

10µF

16V

V

CC

33k

237k

107k

10.0k

S1B

B1100 s 8 PLCS

2.2µF

100V

+

10µF

100V

10µH

DO1608C-103

1mH

DO1608C-105

6.8µH
PG0702.682

10.0k

OUT

4.7nF

1nF

5.1Ω

1nF

0.1µF

100V

4.7nF

250V

10nF

+

220µF

6.3V

PSLVOJ227M(12)A

5V

5A

36V

PDZ36B

BSS63LT1

V

PORTP

–48V

AUXILIARY

POWER

–54V FROM

DATA PAIR

–54V FROM

BC857BF

•

•

•

EFFICIENCY (%)

LOAD CURRENT (A)

50.5

95

65

1 1.5 2 2.5 3 3.5 4 4.5

70

75

85

80

90

VIN = 42V
VIN = 50V
VIN = 57V

In the LTC4269, handshaking cir­cuitry, also known as the “High Power
Available,” “Two Finger Detect,” or
“Ping Pong” indicator, allows the PD
to take full advantage of a new PSE’s
full 25.5W of available power. Both
parts include an integrated Hot Swap
MOSFET for a controlled power up of
the PD. The switch has a low 700mΩ
(typical) resistance and a robust 100V
max rating, thus meeting the needs of
a wide range of applications. Auxiliary
power supplies (“wall warts”) can be
accommodated by interfacing to the
SHDN pin to disable the PoE power
path. Setting a programmable clas­sification current allows different
power leveled PDs to be recognized
by the PSE. Achieving this is as easy
as choosing the proper resistor and
placing it from the R
pin. The ICs are chock-full of protec­tion features, including overvoltage,
undervoltage, and overtemperature to
name a few. Finally, complementary
power good indicators signal that the

CLASS

pin to V

PORTN

PD Hot Swap MOSFET is out of the
inrush limit and ready to draw full
power.

The power supply controllers of the
LTC4269s also share some features.
Both offer programmable switching
frequency, which allows the designer
to optimize the trade-off between ef­ficiency and size, or the designer can
choose a specific frequency to meet
application specific EMI require­ments. The power supply soft-start
time is also adjustable to prevent
the PSE from dropping out its power
due to excessive inrush current and
virtually eliminate any power supply

DESIGN FEATURES L

Figure 3. LTC4269-2-based self-driven synchronous forward converter

Figure 2. Efficiency of the circuit in Figure 1

Linear Technology Magazine • September 2009

7

L DESIGN FEATURES

EFFICIENCY (%)

LOAD CURRENT (A)

50.5

95

65

1 1.5 2 2.5 3 3.5 4 4.5

70

75

85

80

90

VIN = 42V
VIN = 50V
VIN = 57V

output voltage overshoot. Both parts
include short circuit protection with
automatic restart.

LTC4269-1 Synchronous
Flyback for Optimized
Combination of Efficiency,
Simplicity, Size and Cost

A synchronous flyback supply utiliz­ing the LTC4269-1 offers the best
combination of efficiency, simplicity,
size and cost. See Figures 1 and 2 for
the schematic and efficiency curves,
respectively, for an LTC4269-1-based
PD power supply capable of a 5V out­put voltage at 5A.

The flyback parts count is low for a
few reasons. There is no need for the
large output inductor that a forward
converter (see Figure 3) needs, for this
function is rolled into the isolation
transformer (T1). A small, inexpensive
second-stage filter inductor (L1) is
used in the flyback in order to reduce
output voltage ripple, but it should not
be confused with a traditional output
inductor.

In the case of the LTC4269-1, nei­ther a secondary side reference nor an
optocoupler are needed to transmit the
output voltage regulation information
across the isolation boundary. This is
because the IC uses the third (bias)
winding on the transformer, T1, to get
the output voltage information across
the boundary. Finally, the synchro­nous flyback topology requires half
of the switching MOSFETs (only two)
needed by the forward converter.

Performance, in terms of effi­ciency, tops out at above 90% for the

Figure 4. Efficiency of the circuit in Figure 3

LTC4269-1 synchronous flyback. As
a contrast, typical PoE efficiencies at
the “af” power level for a convention­ally rectified flyback were in the lower
half of the 80%’s. This higher effi­ciency is due to the IC’s well-controlled
implementation of the synchronous
rectifier’s gate drive. This efficiency
is not attainable with an uncontrolled
self-driven synchronous rectification
scheme that is sometimes used.

Regulation over the full PoE+ input
voltage range and 0A to 5A output cur­rent range for the LTC4269-1 is better
than ±1%. Output voltage ripple for
the fundamental switching frequency
is less than 30mV peak-to-peak.

LTC4269-2
Synchronous Forward
to Maximize Efficiency

If the efficiency of a PoE+ power supply
is paramount, an LTC4269-2-based
synchronous forward supply is the an­swer at 92.5% efficiency. The increased
efficiency comes with the trade-off of
increased circuit size and complexity.

Figure 3 shows a complete PD power
supply. Figure 4 shows efficiency,
and Figure 5 compares the physical
size of the flyback (LTC4269-1) versus
the forward (LTC4269-2). The forward
supplies 5V at 5A.

The increase in the forward’s ef­ficiency comes about in part from
decreased RMS currents in the second­ary side MOSFETs and in part from
separating the transformer and output
inductor. Both of these changes from
the flyback reduce resistive losses.
The forward supply uses twice the
number of MOSFETs as a flyback so
each switch handles just a portion
of the current that the switches in
the flyback do, thus reducing the I2R
power losses. By separating the isola­tion transformer and output inductor,
instead of using the transformer for
both as in the flyback, the same power
is processed through two components
instead of one. The net effect is more
copper, thus less resistance and lower
resistive losses.

The cost of the circuit obviously
increases with the addition of larger
and more expensive power path
components. Complexity also goes
up with the need to control twice as
many MOSFETs. Also, the forward
topology does not lend itself to the
third winding feedback method. This
means extra complexity in the design
and compensation of a secondary side
reference and opto-coupler circuitry.

Other than the ultra high efficiency
of the LTC4269-2’s synchronous
forward, the solution has similar per­formance to the flyback. The output
ripple of the fundamental switching
frequency is about 40mV peak-to­peak. The regulation over the entire
input voltage and load current range
is well under ±1%.

8

Figure 5. LTC4269-1 and -2 solutions

Conclusion

Two new highly integrated PD control­ler ICs are fully compliant with, and
take full advantage of, the upcoming
IEEE 802.3at PoE+ standard. The
LTC4269 family of parts support the
preferred high performance power
supply topologies for use in the new
standard.

Linear Technology Magazine • September 2009

L

DESIGN FEATURES L

IGNITION SOURCE

THERMAL OR

ELECTRIC

OXIDIZER

AIR OR

OXYGEN

FUEL

GAS, VAPOR
OR POWDER

COMBUSTION

Surge Stopper IC Simplifies Design of
Intrinsic Safety Barrier for Electronics
Destined for Hazardous Environments

by Murphy Pickard, Hach Co.

Introduction

As applications for electronic instru­mentation proliferate, an increasing
number of applications require equip­ment safe enough to operate in
hazardous environments. Chemical
plants, refineries, oil/gas wells, coal,
and textile operations are all examples
of potentially explosive environments
that use electronic instrumentation.
In order to operate safely in such en­vironments, instrumentation must be
made explosion proof.

Companies that supply appara­tus to these markets must integrate
protection into the design. It falls to
the electronic designer to consider
available safety measures and imple­ment them with minimum cost and
impact on proper circuit operation.
This is a daunting task from a design
standpoint, made even more difficult
by the number of hazardous environ­ment standards that must be met to
satisfy global or domestic markets.
Although the various standards are
moving slowly to harmonization, in
some cases they still contradict them­selves and each other.

This article discusses the essential
requirements of safety standards, and
methodologies for meeting these re-

Table 1. Established protection techniques

‘Ex’ Designation Technique Description Application

‘p’ Separation: Gas Pressurization Equipment Rooms
‘o’ Separation: Liquid Oil Fill Transformers
‘q’ Separation: Semi-Solid Sand Fill Instrumentation

‘m’ Separation: Solid Encapsulation Instrumentation

‘n’ Construction Nonincendive Switchgear

‘e’ Construction Increased Safety Lighting, Motors

‘d’ Containment Flameproof Pumps

‘i’ Electrical Design Intrinsic Safety Instrumentation

Figure 1. The ignition triangle

LT4356 series surge stopper

IC can be used to design

an active barrier with

parameters that can be

easily altered to quickly

produce custom barriers.

Since the fundamental

circuit topology won’t

be changing much, once

such an active design is

approved, it will be more

readily approved when only

component value changes

are made.

About the Author

Murphy Pickard is an Electronic
Engineer in the Flow & Sampling
Business Unit of Hach Company
(www.hach.com) of Loveland, CO.
If you have questions about this
article or intrinsic safety bar ­rier design, feel free to contact
the author at 800-227-4224 or
[e-mail].

quirements. In particular, the LT4356
series of overvoltage/overcurrent pro­tection devices offers an efficient and
elegant means of creating protection
barriers in electronic apparatus. To
fully understand the requirements
and solutions, one must become mod­erately acquainted with the standards
themselves, and the agencies that
enforce them.

Intrinsic Safety and the
Classification of Hazardous
Environments

Simply put, in a hazardous environ­ment, the designer’s task is to prevent
an ignition source from meeting an
explosive atmosphere. There are sev­eral techniques for achieving this end,
and this article focuses on a design
discipline referred to as intrinsically
safe (IS) design. Figure 1 depicts the
ignition triangle, illustrating that a
fuel, an oxidizer and an ignition source
must all be present for an explosion to
occur. Several techniques simply pre­vent an existing ignition source from
contacting an explosive atmosphere,
while Intrinsically Safe design actually
eliminates the ignition source. The
principal protection techniques are
listed in Table 1.

Separation techniques are well
suited for many applications but
require special sealing methods and

Linear Technology Magazine • September 2009

9

L DESIGN FEATURES

substances, often creating a perma­nent barrier, making repair or service
impossible. Construction techniques
are mechanical approaches, and again
require special materials.

Only the Intrinsic Safety technique
allows normal instrument fabrication
methods and materials and requires
no exotic construction or packag­ing. Additionally, IS circuits may be
serviced with power present, and are
generally the lowest cost approach
to gaining certification. Further, only
IS certified equipment is allowed in
ATEX Zone 0 areas (Directive 94/9/
EC ATEX “Atmosphères Explosibles”).
This is true because the instrument
design ensures that there is not
enough electrical (spark) or thermal
energy present to serve as an ignition
source. Specifically, an Intrinsically
Safe circuit is one in which any spark
or any thermal effect produced in the
conditions specified in the principal
Standard (IEC 60079-2006), which
includes normal operation and speci­fied fault conditions, is not capable of
causing ignition of a given explosive
gas atmosphere.

Several bodies oversee compliance
to standards and issue certifications
to manufacturers. In North America
FM, UL and CSA govern IEC-79 series
standard certification, while ATEX
standard compliance in the European
Union is certified principally by DEM­KO. The level of protection required
depends on the environment in which
the instrument will operate. Interna­tional Standards and Codes of Practice
classify environments according to
the risk of explosion. The type and
the volatility of the gas/vapor/dust
present and the likelihood of its pres­ence determine such risk. Depending
on the jurisdiction, the classification
system is by Class/Division (North
America) or Zone (EU). These systems
are generally compatible, and for the
purposes of this article, we concentrate
on the Class/Division system as many
countries have adopted IEC79 series
Standards, the most fully utilized and
harmonized of all standards extant.

When electrical equipment and
flammable materials are present si­multaneously, both the equipment and

Table 2. Hazardous environment classification systems

Class Hazard

I Gas/Vapor

II Dust

III Particles/Fibers/Filings

Division

(North America)

1 Likely

2 Unlikely 2 Unlikely

Gas Group Industry

I Underground

II Surface

Apparatus Group Representative Gas

IIA Propane
IIB Ethylene
IIC Hydrogen

Temperature Code Maximum Surface Temperature °C (40°C Ambient)

T1 450
T2 300
T3 200
T4 135
T5 100
T6 85

Presence

explosive atmospheres must be clas­sified. The level of protection provided
must be the same or better than that
required by the standards for use in
such environment. The environment,
or “plant,” is classified according to the
type (Class and Group) and probability
of presence (Division) of the explosive
atmosphere. The equipment is classi­fied according to the maximum surface
temperature (Temperature Code) of
any component of the equipment ex­posed to the hazardous atmosphere,
and by the maximum amount of energy
(Apparatus Group) it can produce or
release in a spark event. It is important
to understand that there is no relation­ship between the surface temperature
and the spark ignition energy neces­sary to ignite a given gas. These limits

Zone

(Europe)

0 Continually
1 Likely

Presence

The Role of Electronic Design
in Intrinsic Safety

An IS circuit is defined in Standard
IEC79-11 as:

“A circuit in which any spark or
thermal effect produced in the con­dition specified in this International
Standard, which include normal op­eration and specified fault conditions,
is not capable of causing ignition in a
given explosive gas atmosphere.”

Thus, a circuit must contain safety
components that prevent spark or heat
energy of a sufficient level to cause an
explosion under fault conditions. It
is the responsibility of the circuit de­signer to incorporate these protective
components into the design while still
maintaining proper circuit operation.
This is seldom an easy task.

are summarized in Table 2.

10

Linear Technology Magazine • September 2009

Any device designed for use in

R

V

OC

I

SC

INTRINSICALLY

SAFE

EQUIPMENT

HAZARDOUS AREA NON-HAZARDOUS AREA

APPROVED APPROVED

INTRINSIC

SAFETY

BARRIER

CONTROL

EQUIPMENT

ROOM

hazardous environments may be
categorized as either a simple or non­simple apparatus. Without going into
detail, a simple apparatus requires no
agency certification if it contains pas­sive components, does not generate or
store significant energy greater than

1.5V, 100mA, and 25mW. Examples of
simple apparatus are resistors, diodes,
LEDs, photocells, thermocouples,
switches, terminal blocks and the like.
For obvious reasons we will not dwell
on this class of equipment.

A non-simple IS apparatus, with
which electronic instrument design­ers are concerned, are categorized as
either “Ex ib,” which may have one
countable fault, and “Ex ia,” which may
have two countable faults. Countable
faults refer to arbitrary faults imposed
by the examiner to analyze efficacy of
protection against thermal and spark
ignition faults. A non-countable fault
occurs not from component failures,
but from circuit spacing issues such
as creepage/clearance, improper
component voltage/current/power
rating or component construction. It
is the designer’s job to ensure that
his component selection and circuit
layout do not contain any non-count­able faults or he may fail certification
from these alone.

During the compliance examination
the assessor is allowed to fail one (Ex
ib) or two (Ex ia) protective compo­nents and explore the implications for
safety of these failures. If these failures
do not degrade the circuit’s safety
features, the apparatus is awarded
a hazardous location certification.
Referring to Table 2, a certification
to Class I, Division 1, Group IIC, T6
allows operation in any hazardous
environment, including ATEX Zone 0

Linear Technology Magazine • September 2009

Figure 2. Isolation/protective barrier location

areas. Clearly, Ex ia is the most dif­ficult certification to obtain, and the
manufacturer should determine that
he must have this level of protection
before incurring the cost of doing so.
Most applications require only Class
I/Div 1 or 2 (Zone 1) certification.

The Barrier Concept

A barrier that limits power/voltage/
current to safe levels for the par­ticular environment must moderate
any power or signaling flow between
a hazardous location and a non­hazardous location. Such a barrier
is termed an Associated Apparatus
in the Standards. It is important to
realize that an IS barrier, containing
protective components, resides in the
non-hazardous area and supplies
power to the IS certified apparatus in
the hazardous area, including Simple
Apparatus. Both pieces of equipment
must comply with IS rules. That is
to say that for an Ex ia certification,
both units must be approved to suffer
double faults while maintaining safety
from ignition as Figure 2 illustrates.
Proper or merchantable operation
of the apparatus is irrelevant to the
examiner, as long as it is safe.

The concept of a barrier is a powerful
tool in gaining compliance. It is clear
that the non-hazardous area barrier
in Figure 2 must limit the total power
available to the IS apparatus in the
hazardous area. However, multiple
barriers may also exist within the

Figure 3. Simple passive component barrier

DESIGN FEATURES L

hazardous area apparatus. Internal
barriers may be used to further limit
power to sub-circuits within the equip­ment to prevent application of multiple
countable faults.

In the broadest terms, protective
components are either series type or
shunt type. A current-limiting resistor
is the most common series protective
device, while a voltage-limiting Zener
diode is the most common shunt
protective device. When used in com­binations to limit power, protective
devices are referred to as barriers.
Barriers in which true galvanic isola­tion is maintained are referred to as
“isolators.” Examples of isolators are
transformers, capacitive couplers and
optical couplers. Isolators however will
not provide DC power or transfer DC
signals and are not germane to this
discussion. We will not delve into the
use of resistors or diodes to isolate
energy-storing components to provide
spark ignition protection, but this is
provided for in the Standards and
is a different concept from galvanic
isolators.

Safety Components
and Barrier Design

Barriers can be categorized as either
passive or active according to the
components used to design them.
Passive barriers have the advantage
of conceptual simplicity, ease of
design and ready availability in the
market. However, the protected field
apparatus must suffer the voltage
burden imposed by the barrier and
still function properly. Passive barriers
are energy inefficient and bulky. If any
significant power must be transferred
to the field device beyond a few mil­liwatts, the safety components become
very large.

Active barriers have a tremendous
advantage in efficiency and component
size, but are generally more difficult to
design and may be more expensive to
produce. Additionally, these are typi­cally custom designs that are not easily
reused. The most serious disadvantage
of active barriers is not conceptual,
but bureaucratic. The examiners
who analyze the barrier design are
completely familiar with common pas-

11

L DESIGN FEATURES

SHORT CIRCUIT CURRENT (mA)

OPEN CIRCUIT VOLTAGE (V)

1k10

10k

10

100

100

1k

Grp-A&B: Acetylene,Hydrogen
Grp-C: Ethylene
Grp-D: Propane
Grp-D: Methane

–

+

–

+

V

CC

SHDN

IN+

AMPLIFIER

TIMER

IA

1.25V

50mV

1.25V

SNS

OUTPUT
TO LOAD

SUPPLY

INPUT

TMRGND

GATE

A

OUT

OUT

FLT

EN

FB

+

–

30MA

VA

LT4356-1

sive designs, and may require actual
spark testing (at your expense) before
approving active designs. However, as
we will see, the LT4356 series surge
stopper IC can be used to design an
active barrier whose parameters can
be easily altered to quickly provide
custom barriers. Since the fundamen­tal circuit topology won’t be changing
much, once such an active design is
approved, it will be more readily ap­proved when only component value
changes are made. If the IS instrument
supplier is performing even a few IS
barrier designs, significant savings are
realized in energy efficiency, barrier
size and cost.

A passive design for associated ap­paratus, the barrier, that supplies DC
power to the field apparatus utilizes
three venerable passive devices to
implement protection: fuses, resistors
and Zener diodes. Safety factors of

1.5 or 1.7 are applied to these device
parameters. Furthermore, for double­fault protection at ‘ia’ protection level,
multiply redundant components are
necessary. Figure 3 shows the most
common type of passive barrier design
as an example.

Only the Zener diodes can limit open
circuit voltage and only the resistor
and fuse can limit current. Fuses are
not considered as a spark-ignition
energy limit device because of its
slow reaction time. In each case, the
devices dissipate power and must be

12

at a worst-case barrier performance,
always erring on the side of safety.

The barrier is assumed to pass a

maximum power of V

OC

• I

SC

= P

MAX

/2
when the field apparatus impedance is
equal to the barrier source impedance,
the point of maximum power transfer.
For this analysis the resistor value is
assumed to be (R – %tolerance) and V

OC

at (Vz + %tolerance). Any component
in the field apparatus must be able

Figure 4. Resistive circuit
spark ignition curves

to tolerate P
lower values by secondary means. If
we assume that the field apparatus is

/2 unless protected at

MAX

nothing more than an LED, the LED

properly rated. The Zeners actually do
sink some reverse leakage current even
though they are not fully on.

The examiner assumes the Zener
voltage knee to occur at the high
end of its tolerance, usually 5%. The
Zener must be rated at 1.5 times the
maximum power of the barrier, the
resistors must be rated at 1.5 times
the maximum power and the fuse is
presumed to pass 1.7 times its rated
current. The resistor is presumed
to be at the low end of its tolerance
range. All active and passive devices
must also have an absolute maximum
breakdown voltage specification that
is 1.5 times the maximum operating
voltage they will encounter in normal
or fault conditions. These presump­tions are imposed not to frustrate
the electronic designer, but to arrive

must be able to dissipate P

MAX

/2 with­out exceeding the apparatus Surface
Temperature code, such as 85°C for
a T6 rated product.

In practical barrier designs, pro­tective component redundancy is
necessary for compliance, especially
for Zener diodes. Two Zeners in parallel
are required for Ex ib rated equipment,
and three parallel Zeners for Ex ia
protection level. Note that the Zener
power dissipation rating depends on
the fuse clearing. If the fuse were not
present, proof must be supplied that
the Zener can dissipate the full barrier
power indefinitely without failing or
exceeding the temperature rating of
the apparatus. In addition, the IEC79
Standard requires that all fuses not
contained in approved holders must
be encapsulated. Further require­ments exist for the protective resistor:
it must be “infallible.” If two resistors
are used in series, each resistor must
be of a high enough value as to limit
current if one of them fails short. If
two resistors are used in parallel,
each must be specified to dissipate the
maximum fault power if one resistor
fails open. An infallible resistor is one
of metal film, ceramic glazed wire­wound, or thick film SMD type with
a conformal coating, all with suitable
creepage/clearance spacing to avoid
a non-countable fault. The infallible
resistor is considered to fail only to

Figure 5. Simplified block diagram of the LT4356

an open circuit. The examiner may
take this as one countable fault, but
unless it reveals failures downstream
of the resistor, it does not inform the
analysis.

Linear Technology Magazine • September 2009

DESIGN FEATURES L

C

TMR

0.1µF

R

SNS

10mΩ

Q1

IRLR2908

Q2

IRLR2908

V

IN

12V

V

OUT

12V, 3A
CLAMPED
AT 16V

LT4356S

GND TMR

OUTSNS

SHDN

A

OUT

IN

+

V

CC

EN

FLT

FB

D2*

SMAJ58CA

R2

4.99k

R1
59k

GATE

R7
10k

R5

1M

Q3

2N3904

D1

1N4148

R3
10Ω

R4
10Ω

*DIODES INC.

Despite their simplicity, passive
barriers exact a high price in power
loss and size. Maximum power is
transferred to field apparatus only
when its input impedance is equal to
the resistance of the current limiting
resistor in the barrier, and this is only
half of the power supplied to the bar­rier. If more than a few milliwatts are
required in the field apparatus, the
barrier resistor may become physically
large. Such resistors are understand­ably expensive, have a limited value
range and are difficult to source and
mount. If a fuse is not included in
the design, the Zener diodes likewise
become bulky and expensive. The fact
that the fuse must be encapsulated
(Paragraph 7.3) usually dictates that
the entire barrier is encapsulated,
making it impossible to service as
well as messy and more expensive to
manufacture.

Determining Maximum Safe
Field Apparatus Power Limits

The actual power that may be trans­ferred to a field apparatus through
the associated apparatus barrier is
determined entirely by the level of
certification the instrument supplier
is seeking. This in turn is determined
entirely by the environment it will
encounter.

The Class and Division rating de­sired is easily determined. However,

Linear Technology Magazine • September 2009

Figure 6. Redundant pass transistors

the flammable gas/dust type is what
determines the Apparatus Group and
T code. The fact that hydrogen has a
relatively high ignition temperature
(560°C) and very low spark ignition
energy (20µJ) demonstrates that
careful thought must given to these
parameters before seeking certification
testing. Here we confine our discus­sions to Class I locations, gasses and
vapors in surface operations, Group II.
To determine how much power can be
available at the output of a barrier, and
still be safely faulted open or shorted,
we utilize the empirically determined
gas ignition curves published in the
standards. These curves indicate the
maximum voltage and current allow­able for a given gas group.

There are three charts published in
the standards, one for resistive, induc­tive and capacitive circuits. Figure 4
shows the curve for a simple resistive
circuit. For sake of discussion, we
assume that we are dealing with the
worst environment for spark ignition,
acetylene, Group IIA. Referring to Fig­ure 4, at 20V

it appears that up to

OC

400mA ISC is allowed without danger
of ignition. Additionally, this power
must not permit a corresponding
surface temperature rise high enough
to thermally ignite the gas in normal
or fault conditions.

Some authorities recommend derat­ing the voltage V

by 10% and the

OC

current ISC by 33%. This is stated in the
standards (IEC 60079-11, 10.1.4.2)
under safety factors. The calculated
value of the current limiting series
resistor is simply V

OC

/I

= 20/0.4

SC

= 5Ω. The power the resistor must
dissipate is V
(V

)2/R, whichever is highest dur-

OC

OC

• I

or (ISC)2/R or

SC

ing circuit operation or fault. Simple
calculations show that even small
amounts of power may require rather
physically large current-limiting resis­tors. A final note: the Standards state
that from empirical and analytical
data, a T4 (135°C) temperature code
is automatically awarded to any circuit
using 1.3 watts or less.

Using the LT4356
Surge Stopper as an
Intrinsic Safety Barrier

The LT4356 series of overvoltage/
overcurrent limiters are excellent
choices for designing active protective
barriers with minimum parts count
and wasted power. Recognizing this
fact, Linear Technology offers the
IC in a 16-lead SO package with pin
spacing sufficient to avoid penalizing
the design with a non-countable fault
when encapsulated. For voltages up
to 10V, some Standards require a

1.5mm (59.1mil) creepage spacing,
and 2.0mm (78.7 mil) for up to 30V.
Before the 2006 79 series Standard,
the IC must be encapsulated to meet
these requirements because of the 50
mil (1.2mm) lead spacing of the 16­lead SO package, but encapsulation
has the added advantage of raising
the thermal limits on any associated
components in the circuit.

However, the latest version of the
harmonized Standard, IEC60079-11
(5th edition 2006-07) dramatically
reduces these creepage requirements
on printed circuit boards when the
apparatus is enclosed in such a way
as to meet ingress protection stan­dards. These standards, known as
IP levels, prevent ingress of dust or
moisture, thereby guaranteeing a pol­lution degree of 2 or less. The idea is
that the cleaner and drier the circuit
board stays, the lower the board’s CTI
(Comparative Tracking Index) and the
less likely leakage current will occur.

13

L DESIGN FEATURES

C

TMR

0.1µF

R

SENSE

3 s 0.5Ω

IN PARALLEL

Q1

IRLR2906

INPUT TEST

VOLTAGE STEP

0V TO 15V

VINV

IN

V

OUT

9.9V
300mA

LT4356CDE

GNDTMR

OUTSNS

SHDNSHDN

A

OUT

IN

+

V

CC

EN

FLT

FB

C

TMR

0.1µF

4.99k

GATE

22µF
25V

100k

10Ω

SMAJ58A*

*DIODES INC.

UNDERVOLTAGE LOCKOUT

LED SUPPLY

C

SNUB

0.1µF
100V

1M

1k

BZT52C16T

BZT52C5V6T

1M
OPT

R

SNUB

10Ω

3.9k

250mW

LED

GREEN

3.9k

3.9k

22k

34.8k

LED
GREEN

LED
RED

LED

GREEN

DRAN

DRAN

+

MMBT5551

Figure 7. Schematic of a modified DC1018A evaluation board

Annex F of 79-11 therefore allows only

0.2mm creepage all the way up to 50V
for Class I environments. Since most
instrumentation is enclosed anyway,
it behooves the designer to use an
enclosure with a high IP rating, such
as IP67 or IP68 to avoid encapsulation
requirements. Unless encapsulation
is necessary to meet thermal limits,
its cost and associated problems are
best avoided.

Figure 5 is a simplified block dia­gram of the LT4356 IC. The LT4356
monitors both current and voltage
continually and turns off the series
pass MOSFET quickly if a fault oc­curs. Both current and voltage limits
are set by external components, so
limits may be changed easily. The
current shunt resistor and the voltage
feedback resistors should be made in­fallible to achieve certification. Usually
the feedback resistors can be made
arbitrarily large so that a MOSFET
fault that shorts input power directly
to the feedback resistors cannot cause
significant power dissipation.

Nevertheless, two cautionary notes
are in order. The first is that active
devices (controllable semiconductors)
can be used in Ex ib situations for
power limitation (thermal ignition)

14

but not for spark ignition protection.
See paragraphs 7.5.2 and 7.5.3 in the
Standards. Some interpretations may
allow active barrier use in Zone 0, but
only in triplicate form. The second
caution is that, as with any IS barrier,
even for Ex ib (single fault) applica­tions, barrier failure usually results
in non-countable thermal fault failure
downstream of the barrier. Therefore,
redundancy is required in case one of
the barriers fails.

The LT4356 provides for two series
pass transistors, typically for reverse
polarity protection. Protection against
polarity reversal is required “where this
could occur.” A single diode is deemed
acceptable to satisfy this requirement,
but two pass transistors offer better
protection from countable faults with­out a significant voltage drop.

For Ex ib environments, the exam­iner can use his single countable fault
to internally short all the pins on the
IC to analyze resultant failures. While
properly rated redundant Zeners could
be positioned at the output of the
LT4356 to provide a voltage limit, at
any appreciable power level the cost
and difficulty of specifying these Zeners
makes it more cost effective to simply
duplicate the entire barrier. Note that

for Ex ia applications, either triplicate
barriers, or two barriers with a series
infallible resistor are required to meet
the double-fault analysis rule.

From here on, we assume that
spacing and thermal rise, component
ratings, PCB tack width and redun­dancy rules are followed and the circuit
cannot be failed with either countable
or non-countable faults. The remain­ing question is that of spark ignition
energy. For this purpose, the LT4356
may not prove useful, depending on
the application.

The LT4356 reacts to both current
and voltage faults by turning off the
pass transistor(s). However, since it
does not shut down instantaneously,
some amount of energy squirts
through the barrier. In the standards
this is termed the let-through energy,
and is usually assessed using oscil­loscope measurements and/or an
actual spark ignition test in a cham­ber. If this energy is enough to ignite
the subject gas, the barrier has failed
certification. Acceptable let-through
energy is summarized in Table 3.

Bench tests reveal that the LT4356 is
much more than adequate for even Ex
ia thermal ignition applications. Bench
testing was done using a modified

Linear Technology Magazine • September 2009

LT4356 evaluation board DC1018A.

0V

TRIGGER ON = SHORT CIRCUIT

0V

V

TRIGGER

20V/DIV

V

OUT

2V/DIV

20µs/DIV

0V

0V

V

IN

5V/DIV

V

OUT

2V/DIV

20ms/DIV

V

OUT(CLAMP)

= 9.9V

The schematic for the setup appears in
Figure 7. The feedback resistors were
selected for an IS-specific 9.9V voltage
limit and the current sense resistor
value was changed to allow a 300mA
current limit. Both overvoltage and
overcurrent limit performance were
tested. The voltage limit was evaluated
by a step change in input from zero to
15V. The current limit was evaluated
by applying a direct short to the output
ground through a low R
driven by a 5V square wave.

fault recovery options using fault
timers that may be exploited by the
designer of IS apparatus, depending
on the application, but these are not
discussed here. The automatic fault
reset enabled on the evaluation board
is left enabled for testing.

the voltage clamping action when the
evaluation board is powered up with
a 15V supply and a 9.9V clamp limit.
The action of the fault reset timer is
obvious.

the current fault action. It shows that
when the short circuit is applied, by
turning on the load MOSFET, voltage
is clamped to ground in less than 6µs.
Channel 1 is the trigger pulse and
Channel 2 is the barrier output voltage.
Although not shown, the current is also
declining , though not as rapidly as
voltage. The slew rate of the current is
dependent on the power supply source
impedance, the circuit inductance and
the MOSFET gate capacitance, among
other variables. In general, as small a
MOSFET die size as possible should
be used, and it may be necessary to
use a low value resistor in series with

Linear Technology Magazine • September 2009

Figure 8. Overvoltage fault operation

DS(ON)

MOSFET

the barrier output to stay below the
spark ignition thresholds.

The IC series offers a number of

To properly calculate the let­through energy, the power profile
must be derived from both current
and voltage curves and then integrated
over time. Spark ignition testing is only
done on connections that may be bro­ken without opening the instrument
enclosure. That is, cables or connec-

Figure 8 shows a scope trace of

tors to devices outside and beyond the
barrier itself. The examiner may cut
the cable or disconnect connectors
to measure spark ignition potential.
Within the enclosure, only thermal
ignition potential must be assessed.

More importantly, Figure 9 shows

Conclusion

Any supplier wishing to sell equipment
into markets and environments that
may be explosive must follow design
rules that make their operation in such
environments nonincendive. That is,
they must not be capable of provid­ing either thermal or spark ignition
sources. Several standard methods
exist for providing such protection,
but for electronic instrumentation, the
preferred and least costly approach is
usually Intrinsic Safety. The Interna­tional Standards that govern electrical
devices in explosive atmospheres are
convoluted and in many cases vague

Table 3. Permissible let-through energy by IEC/NEC gas group

Apparatus Group Classification Let-Through Energy

Class I Group IIC = Ethylene 20µJ
Class I Group IIB = Hydrogen 80µJ
Class I Group IIA = Acetylene 160µJ

Class I Group I = Methane 226µJ

DESIGN FEATURES L

Figure 9. Overcurrent fault operation

as to the design methods necessary to
achieve compliance. In today’s safety
conscious world, both governments
and markets demand that the appa­ratus be certified to compliance with
the standards. Certification is done by
a number of regulatory bodies known
as Nationally Recognized Test Labo­ratories, and a thorough and detailed
analysis process is performed before
certification is awarded.

Obtaining certification of instru­mentation for IS environments is
greatly eased by proper protective
barrier design. While passive barriers
are simple to design, they exact heavy
penalties in size and cost when more
than a few milliwatts are needed for
proper operation. Active barriers can
achieve safe operation while delivering
several watts of energy, but the design
rules are more complex.

Integrated circuits such as the
LT4356 make active barrier designs
considerably easier to certify if ba­sic rules are followed. The superb
response times of the LT4356 series
voltage/current clamps are key to
meeting regulatory requirements
for limiting power that could cause
thermal ignition. Careful design, and
possibly additional fast clamps may
be needed if the LT4356 is to be used
to limit spark ignition also.

This article does not cover all of the
details necessary for a compliant bar­rier design and thorough study of the
applicable standards is still required
of an IS designer. Nevertheless, certi­fication-ready active barriers are now
very accessible, giving designers and
their companies an unprecedented
opportunity to expand into heretofore,
relatively closed markets.

L

15

L DESIGN FEATURES

Consider New Precision Amplifiers for
Updated Industrial Equipment Designs

Introduction

Industrial equipment is designed for
long life cycles, so the electronic com­ponents used in industrial applications
are often chosen with significant em­phasis on proven performance, quality
and reliability. Precision amplifiers are
no exception. Even if new and innova­tive amplifiers become available over a
product’s lifetime, a redesigned board
is often built using the same proven
op amps in the old board. Even for
entirely new applications, designers
will choose amplifiers that have proven
their mettle in other circuits, making a
choice based more on familiarity than
performance.

Although an amplifier may have
been tried and proven in a design, it
is not necessarily the best solution for
every new design. Many can benefit
from using more recently released
amplifiers, which can improve overall
system performance, reduce power
consumption, shrink the board real
estate and expand the capability of
the system while reducing component
count.

Table 1 shows is a list of high per­formance amplifiers and their features.
Many are pin compatible with older
amplifiers, making it easy to swap
them into existing designs to update
industrial applications.

Table 1. Comparison of old and new high performance industrial amplifiers

Industry

Standard

Amplifiers

LT1078
LT2078

LT1012
LT1097

LT1112
LT1114

LT1494

LT1008

LT1055 Family

LT1169

LT1013
LT1014

Features

❏ Precision
❏ Micropower
❏ Single Supply

❏ Precision
❏ Low Noise
❏ Stable with any C-Load

❏ Low Power
❏ Matching Specs
❏ C-Load Stable

❏ Ultralow Power
❏ Rail-to-Rail
❏ Precision

❏ Picoamp Input

Bias Current

❏ Low Offset

Alternative

Amplifiers

LTC6078*

LT1880

LT1881 Family
LT6010 Family

LT6003*

LTC6240 Family*
LTC6084 Family*
LTC6088 Family*

LT1490A
LT1491A

by Brian Black

Feature

Improvements

❏ Higher Precision
❏ Lower Noise
❏ Faster

❏ Rail-to-Rail Out

❏ Higher Precision
❏ Rail-to-Rail Out

❏ Lower Power
❏ Lower Supply

Range

❏ Smaller Package

❏ Lower Power
❏ Lower Noise
❏ Higher Precision
❏ Faster
❏ Rail-to-Rail Out

❏ Rail-to-Rail In/Out
❏ Over-The-Top
❏ Lower Noise

Old and New Amplifiers Go
Head-to-Head

What follows is a comparison of some
old and new amplifiers, where the
new can easily be swapped in for the
old. Figures 1 and 2 show two ap­plications that can benefit from the
updated features offered in recently
released amps.

Rugged LT1494 vs
Miniscule LT6003

The LT1494 (introduced in 1997) is a
precision micropower (375µV offset
voltage at 1.5µA supply current) rail­to-rail input and output amplifier ideal

16

❏ Low Noise

LT1028

LT1007
LT1037

LT1124 Family

LTC1050

Family

* Maximum supply voltage is lower than predecessor

❏ Low Drift
❏ Unity Gain Stable

❏ Low Noise

❏ Low Noise
❏ Low 1/f Corner
❏ Precision

❏ Zero Drift
❏ No External Capacitors

LT6200 Family*
LT6230 Family*

LT1677 Family

LT6202 Family*
LT6233 Family*

LTC2050 Family*

Linear Technology Magazine • September 2009

❏ Lower Power
❏ Faster
❏ Rail-to-Rail In/Out

❏ Rail-to-Rail In/Out

❏ Lower Power
❏ Lower Noise
❏ Faster
❏ Rail-to-Rail In/Out

❏ Shutdown
❏ Lower Offset/Drift

DESIGN FEATURES L

+

–

0V TO 10V

INPUT

10kHZ

TRIM

5k

4.7k
15V

2N3906

15V

–15V

–15V

= 1N4148

3M

0.001 (POLYSTYRENE)

0.1MF

THE LOW OFFSET VOLTAGE OF LT1056
CONTRIBUTES ONLY 0.1Hz OF ERROR
WHILE ITS HIGH SLEW RATE PERMITS
10kHz OPERATION.

0.1MF

22k

75k

1.5k

LM329

3.3M

7

6

4

3

2

33pF

*1% FILM

OUTPUT
1Hz TO 10kHz

0.005%
LINEARITY

LT1056

Table 2. LT1056 vs LTC6240HV

Feature: LTC1056 LTC6240HV

Rail-to-Rail Outputs NO

Minimum Supply Voltage 10V

Maximum Supply Voltage

Single Supply NO

Supply Current 7mA

V

OS

I

B

Noise Voltage Density 22nV/√Hz

GBW 5.5MHz

Slew Rate

Settling Time

for low power battery operated ap­plications. Its rugged design includes
reverse battery protection along with
Linear Technology’s Over -The-Top

®

feature, which allows inputs to oper­ate above the voltage rails without
affecting the amplifier.

For handheld systems where reduc­ing space and extending battery life
are top design priorities, the LT6003

Linear Technology Magazine • September 2009

Figure 1. Precision: 1Hz to 10kHz voltage-to-frequency converter

L

YES

L

2.8V

L

40V

12V

L

YES

L

3.3mA

800µV

150pA

L

250µV

L

1pA

L

10nV/√Hz

L

18mHz

L

14V/µs

L

600ns

10V/µs

900ns

can be swapped for the LT1494.
The LT6003 is designed specifically
with handheld devices in mind with
higher integration, a smaller package,
and a lower supply voltage than the
LT1494.

The LT6003 also has a lower mini­mum supply voltage, 1.6V vs 2.2V for
the LT1494. This feature allows the
LT6003 to operate on a wider range

of supplies and allows for a deeper
discharge of alkaline batteries (known
for the steep dropoff in battery voltage
when depleted). The LT6003 further
extends battery life with a lower sup­ply current of 1µA vs 1.5µA for the
LT1494. Consistent rail-to-rail inputs
and outputs preserve dynamic range
even at low supply voltages.

Furthermore, the LT6003 is offered
in a tiny 2mm × 2mm DFN package,
which is three times smaller than the
LT1494’s MSOP package. The LT1494
still has the advantage of higher
maximum supply voltage of up to 36V
vs the 18V of the LT6003. Also, the
Over-The-Top inputs of the LT1494
make it a great choice for applications
in which the inputs may go above the
positive supply.

The LT1677 Updates the LT1007
with Rail-to-Rail Inputs and
Outputs

The LT1007, introduced in 1985 as
one of Linear Technology’s first prod­uct releases, is a precision low noise
40V amplifier with a great combina­tion of DC performance, high gain,
and low noise performance, making
it ideal for small signal applications.
However, since neither the inputs
nor the outputs are rail-to-rail, the
designer must take care to consider
the headroom required for the part to
function properly. Systems that can
benefit from rail-to-rail inputs and
outputs as a way to increase dynamic
range, to reduce the supply voltage,
or to eliminate the negative supply
rail altogether, should consider using
the LT1677.

The LT1677 is a single supply
drop-in update to the LT1007 with the
added benefits of rail-to-rail inputs and
outputs. An important feature in low
voltage (as low as 3V), single-supply
applications is the ability to maximize
the dynamic range. The LT1677’s input
common mode range can swing 100mV
beyond either rail and the output is
guaranteed to swing to within 170mV
of either rail. This rail-to-rail benefit
comes with minimal impact on noise
and DC precision.

17

L DESIGN FEATURES

–

+

–

+

S S

S

INVERTING

INPUT

1M

2

3

1

6

5

7

A

1/2 LT1078

10.1k 1M

10.1k

NONINVERTING

INPUT

B

1/2 LT1078

4

8

3V (LITHIUM CELL)

S

OUT

–

+

The LT1112 and LT1114 vs LT1881
Family and LT6010 Family

The LT1112 and LT1114 have a wide
supply range of 2V to 40V, high pre­cision and very low noise; there is
not much missing from these older
standards. An alternative to these
parts is the LT1881 family, which adds
rail-to-rail outputs. The LT1881 family
brings the performance of the LT1112
to applications that need the wide
dynamic range. Another option is the
LT6010 family, which achieves higher
precision than the LT1112/LT1114
and includes rail-to-rail outputs. It
is especially attractive for low power
applications due to its lower supply
current and shutdown capability.

Conclusion

Amplifiers are highly versatile building
blocks that can often be reused from
one system design to the next, which
can simplify redesign. The pitfall of
reuse is that designers can miss out on
the benefits offered by newer amplifi­ers, sometimes settling for sub-optimal
performance, higher costs and larger
system size, when a better solution is
just as easy to use. Not only are most
of the newer devices pin-to-pin func­tional equivalents, they offer additional
benefits such as lower power, smaller
size, or rail-to-rail outputs which can
help next generation designs achieve
longer battery life, better precision and
smaller form factors.

L

Table 3. LT1078 vs LTC6078

Feature: LT1078 LTC6078

Rail-to-Rail Outputs NO

Minimum Supply Voltage

Maximum Supply Voltage

L

2.3V

L

44V

Shutdown Mode NO

Supply Current

V

OS

I

B

L

50µA

120µV

10nA

Noise Voltage Density 28nV/√Hz

GBW 200kHz

Figure 2. AC speed: single battery, micropower, gain = 100 instrumentation amplifier

L

YES

2.7V

6V

L

YES

72µA

L

25µV

L

1pA

L

16 nV/√Hz

L

750kHz

LTC2978, continued from page 5

tion has been selected, the designer
can save the parameters to a file and
upload it to the LTC factory. LTC can
use the file to pre-program parts, thus
allowing the customer to bring up their
boards with minimum hassle.

The LTC2978 utilizes the industry
standard PMBus interface protocol
which is a superset of the I2C compat­ible SMBus standard. PMBus is an
open and widely adopted standard that
clearly defines the protocols for digi­tal power management of individual
DC/DC POL converters. The LTC2978
supports a large number of the PMBus
commands. It also features a number

18

of DC/DC converter manufacturer­specific commands to keep complexity
low and versatility high.

configuration to the LTC factory. From
this, Linear Technology can provide
ready-to-use, pre-programmed de­vices, customized for the particular

Conclusion

With its unprecedented parametric ac­curacy, rich feature set, and modular
architecture, the LTC2978 is an ideal
solution for managing large arrays of
DC/DC POL converters.

The industry standard PMBus in­terface, free PC-based graphical setup
software, and integrated EEPROM
make it easy to customize the LTC2978
for any application. Designers can
use the PC-based graphical interface

application.

Other features include an integrated
precision reference, a multiplexed
15-bit ∆Σ ADC, eight 10-bit voltage­buffered IDACs, eight overvoltage and
undervoltage 10-bit voltage supervi­sors with programmable thresholds
and response times, and an integrated
EEPROM for storing configuration
parameters and fault-log information.
The LTC2978 is offered in a 64-lead
9mm × 9mm QFN package.

L

to configure a device and upload the

Linear Technology Magazine • September 2009

DESIGN FEATURES L

9

2

BUFFER/

OUTPUT

AMPLIFIER

ATTENUATOR

CONTROL

REFERENCE AND BIAS CONTROL

+V

G

+IN

3

–IN

4

5

V

CM

V

CM

10

V

REF

11 1

–V

G

GND8GND12GND15GND

21

EN

222419136

SHDN

18

GND

20

GNDV

CC

VCCV

CC

V

CC

23

GND

DECL2

25

14

DECL1

7

–OUT

16

+OUT

17

EXPOSED

PAD

REFERENCE AND

BIAS CONTROL

• • •

• • •

• • •

+VG OR –VG VOLTAGE (V)

0

GAIN (dB)

–5

0

5

0.6

1.0

–10

–15

–20

0.2 0.4 0.8

10

15

20

1.2

–40°C
25°C
85°C

–VG: NEGATIVE
SLOPE MODE

+VG: POSITIVE
SLOPE MODE

FREQ = 140MHz

FREQUENCY (MHz)

–20

GAIN (dB)

–10

0

10

20

1 100 1000 10000

–30

10

G

MAX

G

MIN

–VG VOLTAGE (V)

0

–5

GAIN CONFORMANCE ERROR (dB)

–3

–1

1

0.2

G

MAX

G

MIN

0.4

0.6 0.8 1.0

3

5

–4

–2

0

2

4

1.2

FREQ = 140MHz

–40°C

85°C

25°C

Analog VGA Simplifies Design
and Outperforms Competing
Gain Control Methods

Introduction

Variable gain amplifiers (VGAs) are
widely used in communications and
imaging applications such as cel­lular radio, satellite receivers, global
positioning, radar, and ultrasound ap­plications. Most of these applications
involve transmit and receive signals
of varying amplitude that need to be
managed within the constraints of the
overall system design. On the transmit
side, the signal amplitude is usually
adjusted near a maximum limit im­posed by the transmit power amplifier
or below a power limit imposed by the
receivers or reflectors of the signal. On
the receive side, the signal amplitude is
usually amplified and tailored to take
optimum advantage of the demodula­tor or ADC that decodes the signal. In
both the transmit and receive case, the
optimum signal gain targets change
over time and temperature, so most
systems share a common require­ment of controlling signal amplitude
through the use of adjustable gain
stages commonly known as variable
gain amplifiers.

This article introduces the LTC6412,
Linear Technology’s first high fre­quency, analog-controlled VGA—now
added to Linear Technology’s existing
portfolio of digitally controlled VGAs.
The design considerations for analog

vs digital control are also discussed.
This is followed by a brief introduction
to the important design and perfor­mance features of the LTC6412 along
with a discussion of a few application
examples.

Analog vs Digital Control
of VGAs

The vast majority of modern com­munication and imaging equipment
contains significant digital hardware
in the form of microprocessors, con­trollers, memory, data busses and the
like, so the choice of analog vs digital
system control would seem to be a
forgone conclusion in favor of the digi-

Figure 1. Block diagram of the LTC6412

by Walter Strifler

tally controlled VGA. While this trend
statement is largely true, it overlooks
important distinctions between the
two types of VGA control.

The digitally controlled VGA is
a natural choice when the system
parameters that determine optimum
gain are known to the digital control
system and are readily available across
a data bus. This information is piped
to the data inputs of the VGA, and the
desired gain is step-adjusted during
noncritical periods in the time-slotted
signal.

The digital control scenario is the
goal of most system designs, but
it leaves many application gaps for

Figure 2. LTC6412 gain vs frequency over gain
control range

Linear Technology Magazine • September 2009

Figure 3. Differential gain vs control voltage
over temperature for the LTC6412

Figure 4. LTC6412 gain conformance error vs
control voltage over temperature

19

L DESIGN FEATURES

RF

OUT

50Ω

PEAK

RF

OUT

= 4dBm

–V

G

0.5V/DIV

0.5µs/DIV

V

CC

GND

SHDNEN

–V

G

V

REF

+V

G

–OUT

+OUT

–OUT

+OUT

–IN

+IN

–IN

+IN

10nF

10nF

10nF

C

F

1000pF

10nF

IF IN IF OUT

–IN

+IN

LTC6412 LTC6400-8

LT5537

V

CC

EN

GND

220Ω

10nF

2.2nF

470pF

10nF

1k 1k1k 1k

180nH 180nH

3.3V

3.3V

3.3V

3.3V

3.3V

3.3V

–

+

½LTC6244

590k

33k

2k

100Ω

AGC SET

–20dB TAP

0.1µF

OUT

INPUT

200mV/DIV

OUTPUT

200mV/DIV

20µs/DIV

INPUT

200mV/DIV

C=330pF

C=1000pF

C=4400pF

20µs/DIV

OUTPUT (200mV/DIV)

clever analog solutions. For example,
what if the information needed to
control the amplifier gain is not known
to the digital control system or no
practical data bus is available? What
if the RF signal through the ampli­fier chain cannot tolerate any step
disturbance in amplitude or phase?
These kinds of situations arise often
enough to sustain a healthy market
for analog-controlled VGAs. A few
such applications are discussed later
in this article.

Design Features

The LTC6412 is an 800MHz analog­controlled VGA manufactured on an
advanced silicon-germanium (SiGe)
BiCMOS process that offers the speed
and performance of a complementary
SiGe bipolar process along with the
flexibility and compactness of a CMOS
process. The term SiGe refers to the
material composition of the bipolar
base layers whereby a SiGe semicon­ductor alloy is used to create critical
bandgap discontinuities and drift
fields within the bipolar devices to
improve high speed performance.

Figure 5. LTC6412 gain control 10dB step
response at IF = 70MHz

Figure 1 shows a block diagram of
the LTC6412. The design employs an
interpolated, tapped attenuator circuit
architecture to generate the variable
gain characteristic of the amplifier.
The tapped attenuator is fed to a
buffer and output amplifier to com­plete the differential signal path. The
circuit architecture provides good RF
input handling capability along with
a constant output noise and output
IP3 characteristic that are desirable for
most IF signal chain applications.

The internal circuitry takes the gain
control signal from the ±VG terminals
and converts this to an appropriate set
of control signals to the attenuator lad-

der. The attenuator control preserves
OIP3 through the interpolated transi­tions and ensures that the linear-in-dB
gain response is continuous and
monotonic over the 31dB gain range
for both slow and fast moving input
control signals, all while maintaining a
fixed input and output terminal imped­ance. The control terminal inputs can
be configured for positive or negative
gain slope mode by connecting the
unused control terminal to the V

REF

pin provided.

The output amplifier employs an
open-collector topology and linearizing
techniques similar to the LT5554. En­hanced clamping circuits provide fast
overdrive recovery up to 15dB signal
compression. The entire circuit runs
off a 3.3V supply at a nominal total
supply current of 110mA.

Electrical Performance

The LTC6412 is a fully differential VGA
designed for AC-coupled operation in
signal chains from 1MHz–500MHz and
provides a typical maximum gain of
17dB and minimum noise figure (NF)
of 10dB over this frequency range.

Figure 6. Analog control loop application circuit at IF = 240MHz. LTC6412 bypass capacitors to
ground omitted for clarity.

20

Figure 7. Measured analog control loop circuit
response to 6dB step changes in input signal
amplitude for CF = 1000pF

Figure 8. Measured analog control loop
response to 6dB step changes in input signal
amplitude over a range of CF values

Linear Technology Magazine • September 2009

DESIGN FEATURES L

GAIN AT 70MHz (dB)

TEMPERATURE (°C)

100–60

20

–15

–40 –20 0 20 40 60 80

15

10

5

0

–5

–10

POT R1: SLOPE ADJUST

POT R2: GAIN ADJUST

0.080dB/°C

0.064dB/°C

0.048dB/°C

0.032dB/°C

0.016dB/°C

TO LTC6412
–VGPIN

0.1µF

0.1µF

3.3V

3.3V

R

2

GAIN
100k

100k

390k

200k

10k30k

MAXMIN

–

+

½LTC6078

R

1

SLOPE

100k

MAX

MIN

10mV/°C

TEMPERATURE

SENSOR

V

CC

GND

SHDNEN

–V

G

V

REF

+V

G

–OUT

+OUT

–IN

+IN

10nF

10nF

10nF

10nF

IF IN

–IN

+IN

LTC6412 DRIVER AMP

BASEBAND

PROCESSOR

10pF

2.2nF

10pF

33nH 33nH

3.3V

3.3V

3.3V

3.3V

3.3V

1k

DIGITAL
AGC
CONTROL

DATA
OUT

0.1µF

3.3V

ADC

8-BIT DAC
LTC2640-8

SPI BUS

1k

Figure 9. Digital control loop application circuit at IF = 240MHz. LTC6412 bypass capacitors to ground omitted for clarity.

At a typical operating intermediate
frequency (IF) of 240MHz, the part
delivers a constant OIP3 = 35dBm
and constant (IIP-NF) = 8dBm over
the –14dB to +17dB gain range. The
flat output noise (NF + Gain) and flat
OIP3 combination produces a uniform
spurious-free dynamic range (SRDR) >
120dB over the full gain control range
at 240MHz. The data sheet describes
the operating performance in more
detail, but a few excerpts are worth
noting here.

Figure 2 illustrates the gain vs fre­quency performance of the LTC6412.
Uniform gain slope and spacing are
maintained throughout the gain
control range and across the recom­mended operating frequency range.

Figure 3 illustrates the gain con­trol response to the ±VG inputs. The
linear-in-dB response is accurately
maintained throughout the gain con-

trol range with an RMS error ripple
of approximately 0.1dB as depicted
in Figure 4.

Figure 5 illustrates a typical gain
step response. The settling time of
400ns is smooth and roughly inde­pendent of the step size. The phase
change is also continuous through
any step and typically less than 5° for
signals of 240MHz or lower.

Typical Applications

Analog AGC

Automatic gain control (AGC) is usu­ally the first application that comes to
mind for an analog-controlled VGA.
The idea is to use the linear-in-dB
VGA together with a linear-in-dB
detector to form a servo control loop
that automatically adjusts the signal
amplitude to a set level. An example of
such a control loop is shown in Figure

6. The loop gain of 100 provides an AGC
accuracy of a few tenths of a dB, and
the dominant pole compensation from
CF = 1000pF provides a well-damped
response time of 15µs shown in Fig­ure 7. Adjusting CF over a 13:1 range
produces a similar proportional range
in settling time (see Figure 8).

The analog gain control loop is an
attractive solution for simple signals.
The linear -in-dB nature of both the
VGA and detector produces control
dynamics that are constant and linear
throughout the control range. The
detector shown in the example is a
peak detector, but an RMS detector
can also be used.

Digital AGC

The analog gain control loop is less
attractive for 3G and 4G communica­tion signals with a high crest factor

continued on page 30

Figure 10. Application circuit for static gain adjust and temperature gain slope compensation
using a PTAT temperature sensing IC. Adjust R1 and R2 as needed and route output to –VG
control terminal of the LTC6412.

Linear Technology Magazine • September 2009

Figure 11. Gain vs temperature performance
characteristics of the PTAT sensor based
circuit shown in Figure 10

21

L DESIGN FEATURES

5V

0.1µF

µC

0.1µF

LTC6930

V

+

GND

DIVA

DIVB

V

+

OUT

GND

IO2

IO1

IO3

CLK

DIVC

f

OSC

5V

0.1µF

µC

0.1µF

LTC6930

V

+

GND

DIVA

DIVB

V

+

OUT

GND

IO1

CLK

DIVC

f

OSC

Accurate Silicon Oscillator Reduces Overall System Power Consumption

Introduction

Choosing a clock used to be simple:
grab an off-the-shelf fixed-frequency
super-accurate, low jitter quartz
crystal, or cobble together a rather
noisy, inaccurate RC oscillator us­ing discrete components. Recently,
though, the number of clock choices
has expanded, making the decision
tougher, giving rise to a number of
important questions. Is crystal ac­curacy absolutely necessary? Are low
power consumption and reliability
important, suggesting an all silicon
solution? What about cheap ceramic
resonators—are they up to the task?

Each of thes e solut i o ns h as
strengths and weaknesses. Power
consumption, accuracy, noise and
durability must all come into consid­eration when choosing a clock. The
LTC6930 is a self-contained, fully
integrated all silicon oscillator that
occupies a unique space within the
world of clock solutions, providing
a combination of accuracy and low
power features that is hard to beat.

The LTC6930, which requires no
additional external components, can
accurately provide fixed frequencies
between 32.768kHz and 8.192MHz
over a wide supply range of 1.7V–5.5V
(Table 1). It typically dissipates be­tween 100µA and 500µA depending on
frequency and load, and is available
in both 8-lead 2mm × 3mm DFN and
standard MS8 packages.

Figure 1. The LTC6930 clock configured as a 2­speed clock, slow and fast clock speeds are set
via one I/O pin on a microprocessor

What is not immediately

obvious about the LTC6930

is that its low power

dissipation represents

only a small part of its

power-saving abilities. Its

accurate and fast start-up

and switching times save

substantially more system

power than the device

consumes by itself.

What is not immediately obvious
about the LTC6930 is that its low
power dissipation represents only a
small part of its power-saving abili-

by Albert Huntington

Figure 2. Fine control of the the LTC6930’s
frequency via three microprocessor I/O pins

ties. Its accurate and fast start-up
and switching times save substantially
more system power than the device
consumes by itself.

Smart Power Savings

Many electronic devices, especially
battery powered portable applications,
use low power sleep mode to conserve
power during times of low activity.
The depth and effectiveness of sleep
modes is limited by recovery require­ments—namely, how fast must the
system come back up to full power. A
standard crystal oscillator can be a ma­jor contributor to recovery delays.

Crystal oscillators can take tens
of milliseconds to produce an ac­curate output when recovering from

DIV Pin Settings

[DIVC][DIVB][DIVA]

LTC6930-4.19

LTC6930-5.00

LTC6930-7.37

LTC6930-8.00

LTC6930-8.19

22

Table 1. LTC6930 available frequencies and settings

÷1 ÷2

000 001 010 011 100 101 110 111

4.194304MHz 2.097152MHz 1.048576MHz 524.288kHz 262.144kHz 131.072kHz 65.536kHz 32.768kHz

5.000MHz 2.500MHz 1.250MHz 625.0kHz 312.5kHz 156.25kHz 78.125kHz 39.0625kHz

7.3728MHz 3.6864MHz 1.8432MHz 921.6kHz 460.8kHz 230.4kHz 115.2kHz 57.6kHZ

8.000MHz 4.000MHz 2.000MHz 1000kHz 500.0kHz 250.0kHz 125.0kHz 62.5kHz

8.192MHz 4.096MHZ 2.048MHz 1024kHz 512.0kHz 256.0kHz 128.0kHz 64.0kHz

÷4 ÷8

÷16 ÷32

Linear Technology Magazine • September 2009

÷64 ÷128

DESIGN FEATURES L

DIV SETTING (LOG)

1 10 100

SUPPLY CURRENT (µA)

600

8.192MHz, 3V

4.194MHz, 1.7V

500

400

300

200

100

0

6930 G04

4.194MHz, 3V

8.192MHz, 1.7V

TA = 25°C

V

OUT

500mV/DIV

200µs/DIV

OUT

500mV/DIV

200µs/DIV

a shutdown. The technique of using
two clocks, a fast clock for full power
operation and a slower sleep mode
clock, can degrade the accuracy and
recovery performance of the system—
where clock switching generates runt
pulses and slivers that can sabotage
sleep recovery times.

In contrast, the LTC6930 easily
and accurately transitions between
fast clock mode and a slower sleep
mode. The transition from one clock
frequency to another takes less than
a single clock cycle, and no runt
pulses or slivers are generated. The
LTC6930 also features a fast 100µs
start-up time and the first clock-out is
guaranteed to be clean. This makes it
possible for the designer to apply sleep
mode liberally, without worrying about
clock recovery, thus saving significant
overall system power.

Shifting the Clock Frequency

The output frequency of the LTC6930
is set by three DIV pins, which control
an internal clock divider. The factory
set master oscillator frequency may
be divided by a factor of up to 128,
and switching between these division
modes is accomplished within a single
clock period and without slivers or runt
pulses. All three pins may be tied to­gether to enable a simple digital signal
from a microcontroller to shift the clock
down by a factor of 128 as shown in
Figure 1. This is enough to bring an
8MHz clock down to 64kHz.

The DIV pins can be addressed
in various combinations for smaller
frequency shifts or independently for
complex power modulating systems
where a microcontroller has fine
control over its own clock speed, as
shown in Figure 2.

Although there are some power
savings within the LTC6930 when the
output frequency is lowered (Figure 3),
far greater savings are realized in the
overall system. Power consumption
in CMOS devices such as microcon­trollers is roughly proportional to their
operating clock speed. Slowing down
the clock by a factor of 128 during a
sleep condition can reduce the system
power by a factor of 100—very impor-

Linear Technology Magazine • September 2009

Figure 3. The LTC6930 supply current at
different divide ratios

tant in a system that spends significant
time in sleep mode.

Power Savings from
Fast Start-Up

Many systems are designed to sleep
most of the time and wake up briefly
on occasion to perform some task. If
a task requires particularly little time,
the total power dissipated for the task
may be dominated not by the awake
time, but by the time it takes for the
oscillator and associated sensory elec­tronics to power up. The guaranteed
fast start-up time of the LTC6930
allows system designers to budget
minimal recovery time and thus save
power in start-up settling time.

Crystal oscillators often specify
start-up times of up to 20ms, if they
specify them at all, and the first clocks
out may be of low amplitude and oth­erwise out of spec. The designers task
is further complicated by the fact that
start-up time may vary randomly. See
Figures 4 and 5 to see how a crystal
oscillator start-up time compares quite
unfavorably to the LTC6930 start-up.
A system that needs to wake up oc­casionally for a millisecond to take

Figure 4. Typical crystal oscillator start-up
transients

a single measurement may end up
spending 100ms waiting for its clock
to come up without a clean signal
and then settle in order to take that
single measurement. The fast and
clean 100µs start-up of the LTC6930
allows the designer of such a system
to reduce wake time, and therefore
power dissipation, again by a factor
of around 100.

A Word on Accuracy

The big question when moving from
a quartz crystal to a silicon oscilla­tor will always be one of accuracy. If
crystal oscillators do anything well,
it is provide a stable and accurate
frequency source, but accuracy is just
one concern out of many.

While each individual application is
different, Linear’s years of experience
with silicon oscillators allows us to
make some general recommendations
based on actual customer applications.
With an initial accuracy of better than

0.09% and a commercial grade accu­racy over temperature of better than

0.45%, the LTC6930 does not compete
with crystal oscillators in all areas, but
does provide a clock accurate enough
for the most applications.

Of course, there are applications
that require either accuracy or jitter
characteristics out of the reach of the
LTC6930, such as clocking high speed
analog-to-digital converters such as
the LTC2242 series, clocking jitter
sensitive high speed serial communi­cations systems such as Ethernet, and
long term timekeeping functions such
as a digital alarm clocks. Nevertheless,
silicon oscillators like the LTC6930
perform far better than crystal oscil­lators when power consumption is a

continued on page 35

Figure 5. Typical LTC6930 start-up

23

V

IN1

V

IN1

V

IN1

V

OUT1

1.8V
4A

V

OUT2

2.5V
4A

V

IN2

3.3V

V

OUT3

1V

1.5A

V

IN1

5V

V

IN3

1.5V

22µF

10µF

100µF

4.02k

2.37k

LTM4615

4.7µF

10µF

=2

82µF

3.32k

V

IN2

P

GOOD1

P

GOOD2

V

OUT2

FB2

COMP2

V

OUT1

FB1

COMP1
TRACK1
RUN/SS1

LDO_IN

EN3

TRACK2

RUN/SS2

LDO_OUT

FB3

P

GOOD3

+

100µF

22µF

82µF

10µF

+

GND

L DESIGN FEATURES

Easy Multivoltage Layout with Complete Dual and Triple Output Point-of-Load µModule Regulators in 15mm × 15mm Packages

by Eddie Beville and Alan Chern

Introduction

Imagine a multivoltage printed circuit
board so space-constrained that even
the most experienced layout engineer
would shiver at the thought of putting
together the puzzle of components for
DC/DC conversion. A typical multi­voltage solution either incorporates a
single multioutput DC/DC regulator
IC or several independent regulators.
Either solution requires a number of
discrete support components, such
as inductors, capacitors and resis­tors. Since there is a wide range of
available small, high performance ICs,
this type of system design is typical.
Unfortunately, even the best of these
regulators require careful placement
of support components to take into
account both electrical effects and
heat dissipation concerns.

Board-mounted point-of-load (POL)
DC/DC power supplies are becoming
increasingly popular as they simplify
board assembly and reduce external
components. The ideal setup would
have nearly everything packaged

Figure 2. Very few components are required for a triple independent input (5V, 3.3V, 1.5V) to
triple output (1.8V, 2.5V, 1V) µModule regulator design.

24

Figure 1. The LTM4614 dual output and LTM4615 triple output µModule regulators

into a single chip, with the following

The LTM4614 (dual output)
and LTM4615 (triple) cure the
headaches inherent in laying

out multivoltage systems

for space-constrained

applications. The MOSFETs,

inductor and other support

components are all built

into the package, so layout

involves little more than
finding a 15mm × 15mm

space on the board.

features in a board-mounted POL
power supply.

q

Minimal components—far fewer

than a discrete solution

q

Multiple voltage input and output

rails with available current
sharing

q

Independent input and output

regulation for application
flexibility

q

Worry-free thermal dissipation

q

Low noise output

q

High efficiency

Complete Dual and Triple
DC/DC Regulators in
IC Form Factors

The LTM4614 and LTM4615 cure
the headaches inherent in laying out
multivoltage systems for space-con­strained applications. Both devices
are point-of-load power supplies in a
15mm × 15mm × 2.8mm LGA surface
mount packages, each with two switch­ing 4A DC/DC regulators (see Figure

1). The LTM4615 adds a VLDO™ (very
low dropout) linear regulator, making
it a triple output voltage regulator.
The MOSFETs, inductor and other
support components are all built into
the package, so layout involves little

Linear Technology Magazine • September 2009

DESIGN FEATURES L

V

IN1

V

IN1

V

IN1

V

OUT1

1.8V
4A

V

OUT2

2.5V
4A

V

OUT3

1V

1.5A

V

IN1

5V

V

IN2

1.5V

22µF

10µF

100µF

4.02k

2.37k

LTM4615

GND

4.7µF

10µF

=2

82µF

3.32k

V

IN2

P

GOOD1

P

GOOD2

V

OUT2

FB2

COMP2

V

OUT1

FB1

COMP1
TRACK1
RUN/SS1

LDO_IN

EN3

TRACK2

RUN/SS2

LDO_OUT

FB3

P

GOOD3

+

100µF

22µF

V

IN1

V

IN

V

IN

V

OUT1

1.8V
4A

V

OUT2

2.5V
4A

V

OUT3

1V

1.5A

V

IN

5V

V

OUT1

1.8V

22µF

10µF

100µF

4.02k

2.37k

LTM4615

GND

4.7µF

10µF

=2

82µF

3.32k

V

IN2

P

GOOD1

P

GOOD2

V

OUT2

FB2

COMP2

V

OUT1

FB1

COMP1
TRACK1
RUN/SS1

LDO_IN

EN3

TRACK2

RUN/SS2

LDO_OUT

FB3

P

GOOD3

+

100µF

22µF

more than finding a 15mm × 15mm
space on the board.

The two switching regulators op-

erate from input voltages between

2.375V to 5.5V (6V peak) and each
delivers a resistor-set output voltage
of 0.8V to 5V at 4A of continuous
current (5A peak). They operate at a

1.25MHz switching frequency using
current mode architecture to enable
fast transient response to line and load
changes with no sacrifice in stability.
The output voltages can track each
other or another voltage. Other fea­tures include low output voltage ripple
and excellent thermal dissipation.

The LTM4615’s VLDO regulator
accepts input voltages from 1.14V
to 3.5V and is capable of up to 1.5A
of output current with an adjustable
output range of 0.4V to 2.6V, also
via a resistor. The VLDO regulator
has a low voltage dropout of 200mV
at maximum load. The regulator can
be used independently or used in
conjunction with either of the two
switching regulators to create a high
efficiency, low noise, large ratio step
down supply—simply tie one of the
switching regulator’s outputs to the
input of the VLDO regulator.

source input voltage, for example. If
the input source voltage is too high
for the VLDO regulator and a separate
source is not available, the VLDO input
of the LTM4615 can be tied to one of
the outputs as in Figure 4.

Single Input, Current-Shared
Outputs

For designs that require more than
the 4A-per -regu l ato r ma x imu m
output current, the two switching
regulators can be tied together to
form a paralleled, single-output 8A
design (see Figure 5). This design
also has efficiency advantages over a
higher-current rated, single switching
regulator design. In the case of the
LTM4615, the VLDO linear regulator
can still be used as an independent
supply.

Power Sharing Multiple Inputs,
Current-Shared Outputs

When a single input source cannot
provide enough current to support
a high power, single current-shared
output, another input, even at a
different voltage, can be used to pro­vide the additional current. Figure 6
shows two different input voltages to
power a single voltage current shar­ing output.

High Efficiency and Low
Noise Output Voltage Ripple

The LTM4615 is capable of operating
with all three regulators at full load
while maintaining optimum efficiency.
Figure 7 shows a typical LTM4615 de­sign for a 3.3V input to three outputs.
In Figure 7, the VLDO input is driven
by V

. The expected efficiency

OUT2

Flexible Input and Output
Combinations

The LTM4614 and LTM4615 power
supplies can be used in a wide range
of input and output combinations;
from entirely independent inputs and
outputs to single input, single output
designs where a parallel, current
sharing design enables high current
applications.

Independent Inputs and Outputs

The L TM4614’s and L TM4615’s
separate inputs and outputs make it
possible to run each internal regulator
from a different input. Figure 2 shows
an application converting 5V, 3.3V,
and 1.5V inputs to 1.8V, 2.5V, and 1V
output voltage rails, respectively.

Single Input, Independent Outputs

For designs that only have a single
source input voltage, tie the input
voltage rails together as in Figure 3,
where both inputs run off the 5V

Linear Technology Magazine • September 2009

Figure 3. Single switching regulator input (5V) and single linear regulator input (1.5V) to triple
output (1.8V, 2.5V, 1V) µModule regulator design

OUT1

(1.8V)

Figure 4. Single input (5V) to dual output (1.8V, 2.5V), with the linear regulator serving up a third
output (1V) from an input tied to V

25

L DESIGN FEATURES

R1

4.02k

LTM4614

V

IN2

V

IN1

V

OUT1

V

OUT2

FB1
FB2

COMP1
COMP2

TRACK2
RUN/SS2

TRACK1
RUN/SS1

V

OUT

1.8V
8A

V

IN

3.3V
82µF

10µF

+

GND

V

IN

100µF

=2

22µF

=2

V

OUT3

(1.0V)

2mV/DIV

V

OUT1

(1.8V)

2mV/DIV

V

OUT2

(1.2V)

2mV/DIV

500ns/DIV

EFFICIENCY (%)

LOAD CURRENT (A)

40

100

0

1 2 3

10

20

30

40

50

60

70

80

90

LDO

1.2V

IN

TO 1V

OUT

3.3V

IN

TO 1.2V

OUT

3.3VINTO 1.8V

OUT

V

IN1

V

IN

V

IN

V

OUT1

1.8V
4A

V

OUT2

1.2V
4A

V

OUT3

1V

1.5A

V

IN

3.3V

V

OUT2

1.2V

22µF

10µF

100µF

4.02k

10k

LTM4615

4.7µF

10µF

82µF

3.32k

V

IN2

P

GOOD1

P

GOOD2

V

OUT2

FB2

COMP2

V

OUT1

FB1

COMP1
TRACK1
RUN/SS1

LDO_IN

EN3

TRACK2

RUN/SS2

LDO_OUT

FB3

P

GOOD3

+

100µF

22µF

GND

R1

4.02k

LTM4614

V

IN2

V

IN1

V

OUT1

V

OUT2

FB1
FB2

COMP1
COMP2

TRACK2
RUN/SS2

TRACK1
RUN/SS1

V

OUT

1.8V
8A

V

IN1

3.3V

V

IN2

5V

82µF

10µF

+

82µF

10µF

+

GND

V

IN1

100µF

=2

22µF

=2

Figure 5. The two switching regulators share the load in a 1.8V/8A output system. (For an
alternative 8A µModule regulator, see the LTM4608A data sheet.)

Figure 6. The two switching regulators combine available power from two independent input
voltage rails (3.3V and 5V) and produce a single current-shared 1.8V/8A output.

of this design is shown in Figure 8.
Expect similar efficiency results with
the LTM4614 minus the additional
VLDO output.

To minimize the number of support
discrete components, both LTM4614
and LTM4615 include internal ceramic
capacitors. Layout problems associ­ated with placing external support
components are eliminated. Addi­tional output capacitors are needed
if load steps from 0A to the full 4A
are expected and if the input source
impedance is compromised by long
inductive leads or traces.

The benefit of combining a switch­ing regulator with a linear regulator is
the noise reduction benefits that can
be gained. By utilizing the switching
regulator’s high efficiency step-down
function and feeding its output to
the input of the VLDO regulator, an
exceptionally low ripple output is pro­duced—ideal for systems that require
a particularly clean signal. Figure 9
shows the low output voltage ripple for
all three outputs. The VLDO regulator
provides a very low noise 1V supply as
it is driven by the output of the 1.2V
switching regulator.

Figure 7. A single input, 3-output design using the LTM4615’s VLDO regulator

Figure 8. Efficiency of the circuit in Figure 7

26

Figure 9. Low voltage ripple on all three
outputs of Figure 7

Thermally Enhanced
Packaging

The LGA packaging allows heat sinking
from both the top and bottom. From the
bottom, the PCB copper layout draws
heat away from the part and into the
board. A heat sink can be placed on top
of the device, such as a metal chassis,

continued on page 32

Figure 10. Top view thermal imaging of the
unit at full load in an ambient temperature
environment with no airflow. Cursors 1 and 3
mark the temperature hot spots on the unit
for each of the two switching regulators. Both
temperatures are fairly similar indicating
balanced thermal conductivity.

Linear Technology Magazine • September 2009

DESIGN IDEAS L

GAIN (dB)

FREQUENCY (kHz)

10001

10

–80

10

100

–30

–40

–50

–60

–10

–20

0

LTC6602

GNDV

OCM

MUTE

DOUT

NC

R

BIAS

CLKIO

PAR/SER

CLKCNTL

SDO

CS

SCLK

SDI

LTC2630

8-BIT DAC

V

+

GND

CS

SCLK

SDI

V

DDINVDDAVDDD

0.1µF

0.1µF

0.1µF

0.1µF

5V 3V

3V

3V

V

OUT

SPI CONTROL OF LTC6602 SETS THE FILTER GAIN

AND THE LOWPASS AND HIGHPASS DIVISION RATIO

I
CHANNEL
OUTPUT

Q
CHANNEL
OUTPUT

I CHANNEL

INPUT

Q CHANNEL

INPUT

174k

FROM
ADC
V

REF

OUTPUT

TRANSMITTER

MUTE INPUT

SCK SDI CS2CS1

68.1k

Programmable Baseband Filter for
Software-Defined UHF RFID Readers

Introduction

Radio frequency identification (RFID)
is an auto-ID technology that identi­fies any object that contains a coded
tag. A UHF RFID system consists of a
reader (or interrogator) that transmits
information to a tag by modulating
an RF signal in the 860MHz–960MHz
frequency range. Typically, the tag is
passive—it receives all of its operating
energy from a reader that transmits
a continuous-wave (CW) RF signal. A
tag responds by modulating the reflec­tion coefficient of its antenna, thereby
backscattering an information signal
to the reader.

DESIGN IDEAS
Programmable Baseband Filter for

Software-Defined UHF RFID Readers

.........................................................27

Philip Karantzalis

µModule LED Driver Integrates All
Circuitry, Including the Inductor,

in a Surface Mount Package .............29

David Ng

Low Power Boost Regulator with
Dual Half-Bridge in 3mm × 2mm DFN
Drives MEMS and Piezo Actuators

.........................................................31

Jesus Rosales

Synchronous Boost Converter with
Fault Handling Generates 5V at

500mA in 1cm2 of Board Space .........33

Eddy Wells

Robust DC/DC Step-Down Converter
in 3mm × 3mm DFN Resists 60V

Input Surges .....................................34

Chuen Ming Tan

4W LED Driver Includes Power Switch,
Compensation Components and

Schottky in 16-Pin MSOP ...................36

Keith Szolusha

3mm × 3mm, 16-Bit ADC Brings
Accurate, Precise High Side

Current Sensing to Tight Spaces .......37

Leo Chen

Dual 8A DC/DC µModule Regulator Is

Easily Paralleled for 16A ..................38

Eddie Beville and Alan Chern

Self-Contained 3A µModule Buck
Regulator Produces 0.8V–24V

Output from 3.6V–36V Input ..............40

David Ng

Linear Technology Magazine • September 2009

Tag signal detection requires mea­suring the time interval between signal
transitions (a data “1” symbol has a
longer interval than a data “0” symbol).
The reader initiates a tag inventory by
sending a signal that instructs a tag
to set its backscatter data rate and
encoding. RFID readers can operate
in a noisy RF environment where
many readers are in close proximity.
The three operating modes, single­interrogator, multiple interrogator
and dense-interrogator, define the
spectral limits of reader and tag sig­nals. Software programmability of the
receiver provides an optimum balance
of reliable multitag detection and high
data throughput. The programmable
reader contains a high linearity direct
conversion I and Q demodulator, low
noise amplifiers, a dual baseband filter

Figure 2. An Adaptable RFID baseband filter with SPI control

by Philip Karantzalis

Figure 1. Filter response
for a 15kHz–150kHz passband

with variable gain and bandwidth and a
dual analog-to-digital converter (ADC).
The LTC6602 dual, matched, program­mable bandpass filter can optimize
high performance RFID readers.

2727

L DESIGN IDEAS

LTC6602 100kHz LOWPASS + HIGHPASS FILTER

10kHz HIGHPASS30kHz HIGHPASS

LTC6602 100kHz LOWPASS

TYPICAL TAG
SYMBOL SEQUENCE

10µs/DIV

The LTC6602
Dual Bandpass Filter

The LTC6602 features two identical fil­ter channels with matched gain control
and frequency-controlled lowpass and
highpass networks. The phase shift
through each channel is matched to
±1 degree. A clock frequency, either
internal or external, positions the
passband of the filter at the required
frequency spectrum.

frequencies, as well as, the filter band­width are set by division ratios of the
clock frequency. The lowpass division
ratio options are 100, 300, 600 and
the highpass division ratios are 1000,
2000, 6000. Figure 1 shows a typical
filter response with a 90MHz internal
clock and the division ratios set to 6000
and 600 for the highpass and lowpass,
respectively. A sharp 4th order ellipti­cal stopband response helps eliminate
out-of-band noise. Controlling the
baseband bandwidth permits software
definition of the operating mode of
the RFID receiver as it adapts to the
operating environment.

An Adaptable Baseband Filter
for an RFID Reader

Figure 2 shows a simple LTC6602­based filter circuit that uses SPI
serial control to vary the filter’s gain
and bandwidth to adapt to a complex
set of data rates and encoding. (The
backscatter link frequency range is
40kHz to 640kHz and the data rate
range is 5kbps to 640kbps.)

the filter, the internal clock frequency
is set by an 8-bit LTC2630 DAC. A
0V to 3V DAC output range positions
the clock frequency between 40MHz
and 100MHz (234.4kHz per bit). The
lowpass and highpass division ratios
are set by serial SPI control of the
LTC6602. The cutoff range for the
highpass filter is 6.7kHz to 100kHz
and 66.7kHz to 1MHz for the lowpass
filter. The optimum filter bandwidth
setting can be adjusted by a software
algorithm and is a function of the data
clock, data rate and encoding. The
filter bandwidth must be sufficiently
narrow to maximize the dynamic range
of the ADC input and wide enough

The lowpass and highpass corner

For fine resolution positioning of

Figure 3. Filter transient response to a tag symbol sequence

to preserve signal transitions and
pulse widths (the proper filter set­ting ensures reliable DSP tag signal
detection).

Figure 3 shows an example of the
filter’s time domain response to a
typical tag symbol sequence (a “short”
pulse interval followed by a “long”

The LTC6602 dual bandpass

filter is a programmable

baseband filter for high

performance UHF RFID

readers. Using the LTC6602

under software control

provides the ability to

operate at high data rates

with a single interrogator

or with optimum tag signal

detection in a multiple or

dense interrogator physical

setting. The LTC6602 is a

very compact IC in a 4mm

× 4mm QFN package and is

programmable with parallel

or serial control.

pulse interval). The lowpass cutoff
frequency is set equal to the recipro­cal of the shortest interval (f
1/10µs = 100kHz). If the lowpass cutoff
frequency is lower, the signal transi­tion and time interval will be distorted
beyond recognition. The setting of the
highpass cutoff frequency is more
qualitative than specific. The highpass
cutoff frequency must be lower than

CUTOFF

=

the reciprocal of the longest interval (for
the example shown, highpass f

CUTOFF

< 1/20µs) and as high as possible to
decrease the receiver’s low frequency
noise (of the baseband amplifier and
the down-converted phase and ampli­tude noise). The lower half of Figure
3 shows the filter’s overall response
(lowpass plus highpass filter). Com­paring the filter outputs with a 10kHz
and a 30kHz highpass setting, the
signal transitions and time intervals
of the 10kHz output are adequate for
detecting the symbol sequence (in an
RFID environment, noise will be su­perimposed on the output signal). In
general, increasing the lowpass f
and/or decreasing the highpass f

CUTOFF

CUTOFF

“enhances” signal transitions and
intervals at the expense of increased
filter output noise.

Conclusion

The LTC6602 dual bandpass filter is a
programmable baseband filter for high
performance UHF RFID readers. Using
the LTC6602 under software control
provides the ability to operate at high
data rates with a single interrogator
or with optimum tag signal detection
in a multiple or dense interrogator
physical setting. The LTC6602 is a
very compact IC in a 4mm × 4mm QFN
package and is programmable with
parallel or serial control.

References:

1

The RF in RFID, Daniel M. Dobkin, 9/07, Elsevier

Inc.

2

Class-1 Generation-2 UHF RFID Protocol for Com-

munications at 860 MHz to 960 MHz, Version 1.1.0,
www.epcglobalinc.org/standards/specs/

L

28

28

Linear Technology Magazine • September 2009

DESIGN IDEAS L

GND

LTM8040

V

IN

SHDN

V

IN

14V TO 36V

C1

2.2µF

LEDA

LPWR

ADJ
PWM
RT

BIAS

1A

LED

CURRENT

500mA/DIV

ADJ PIN

VOLTAGE

500mV/DIV

5ms/DIV

0 AMPS

0 VOLTS

GND

LTM8040

V

IN

SHDN

V

IN

4V TO 36V

C1

2.2µF

5.11k

LEDA

LPWR

ADJ
PWM

BIAS

GND

LTM8040

V

IN

SHDN

V

IN

4V TO 36V

C1

2.2µF

LEDA

LPWR

ADJ
PWM

BIAS

1A

LED

CURRENT

500mA/DIV

PWM

SIGNAL

5V/DIV

20ms/DIV

DN445 F05

µModule LED Driver Integrates All Circuitry, Including the Inductor, in a Surface Mount Package

Introduction

Once relegated to the hinterlands of
low cost indicator lights, the LED is
again in the spotlight of the lighting
world. LED lighting is now ubiquitous,
from car headlights to USB-powered
lava lamps. Car headlights exemplify
applications that capitalize on the
LED’s clear advantages—unwavering
high quality light output, tough­as-steel robustness, inherent high
efficiency—while a USB lava lamp
exemplifies applications where only
LEDs work. Despite these clear advan­tages, their requirement for regulated
voltage and current make LED driver
circuits more complex than the vener-

able light bulb, but some new devices
are closing the gap. For instance,
the LTM®8040 µModule LED driver
integrates all the driver circuitry into
a single package, allowing designers

Figure 1. Driving an LED string with the
LTM8040 is simple—just add the input
capacitor and connect the LED string

to refocus their time and effort on the
details of lighting design critical to a
product’s success.

A Superior LED Driver

The LTM8040 is a complete step-down
DC/DC switching converter system
that can drive up to 1A through a
string of LEDs. Its 4V to 36V input
voltage range makes it suitable for a
wide range of power sources, includ­ing 2-cell lithium-ion battery packs,
rectified 12VAC and industrial 24V.
The LTM8040 features both analog and
PWM dimming, allowing a 250:1 dim­ming range. The built-in 14V output
voltage clamp prevents damage in the
case of an accidental open LED string.
The default switching frequency of the
LTM8040 is 500kHz, but switching
frequencies to 2MHz can be set with a
resistor from the RT pin to GND.

by David Ng

Figure 2. Drive a 0V to 1.25V voltage into the ADJ pin to control the LED current amplitude

Figure 3. Control the LED current with
a single resistor from ADJ to ground

Figure 5. The LTM8040 can PWM LED current with minimal
distortion, even at frequencies as low as 10Hz.

Linear Technology Magazine • September 2009

Figure 4. The LTM8040 can PWM its
LED string with an external MOSFET.

Easy to Use

The high level of integration in the
LTM8040 minimizes external compo­nents and simplifies board layout. As
shown in Figure 1, all that is necessary
to drive an LED string up to 1A is the
LTM8040 and an input decoupling
capacitor. Even with all this built-in
functionality, the LTM8040 itself is
small, measuring only 15mm × 9mm
× 4.32mm.

Rich Feature Set

The LTM8040 features an ADJ pin
for precise LED current amplitude
control. The ADJ pin accepts a full­scale input voltage range of 0V to

1.25V, linearly adjusting the output
LED current from 0A to 1A. Figure
2 shows the ratiometric response of
the output LED current versus the
ADJ voltage. The ADJ pin is internally
pulled up through a 5.11k precision
resistor to an internal 1.25V refer­ence, so the output LED current can

2929

L DESIGN IDEAS

GAIN AT 70MHz (dB)

TEMPERATURE (°C)

100–60

20

–15

–40 –20 0 20 40 60 80

15

10

5

0

–5

–10

POT R1: SLOPE ADJUST

POT R2: GAIN ADJUST

0.080dB/°C

0.064dB/°C

0.048dB/°C

0.032dB/°C

0.016dB/°C

TO
LTC6412
–VGPIN

0.1µF

0.1µF

3.3V

3.3V

3.3V

R

1

SLOPE

100k

100k

100k

12k

20k
14k

20k

390k

MAXMIN

–

+

½LTC6078

R

2

GAIN

100k

MAX

MIN

20k

68k

NTC

also be adjusted by applying a single
resistor from ADJ to ground, as shown
in Figure 3.

The PWM control pin allows high
dimming ratios. With an external
MOSFET in series with the LED string

Figure 6. Only 9mm × 15mm × 4.32mm, the LTM8040
LED Driver is a complete system in an LGA package

LTC6412, continued from page 21

as shown in Figure 4, the LTM8040
can achieve dimming ratios in excess
of 250:1. As seen in Figure 5, there
is little distortion of the PWM LED
current, even at frequencies as low as
10Hz. The 10Hz performance is shown

because the control target is often
more complicated than a simple peak
or RMS amplitude, and the amplitude
noise introduced by the analog control
loop may be unacceptable. A common
solution for these systems is an analog
VGA driven by a DAC as depicted in
Figure 9.

The contradiction of a DAC control­ling an analog-controlled VGA may
appear at first as unusual and unec­essary, but the arrangement provides
key benefits. The gain step resolution is
not determined by the VGA, and 8–12
bit DAC’s are relatively inexpensive.
More importantly, the signal gain can
be adjusted with arbitrary smooth-

Figure 12. Thermistor-based application
circuit for static gain adjust and temperature
gain slope compensation. Adjust R1 and R2
as needed and route output to –VG control
terminal of the LTC6412.

ness, so the baseband processor can
continue its demodulation/decoding
operation without interruption. Most
digital VGAs produce unacceptable
signal discontinuities. The DAC does
have a glitch of its own, but it is a
baseband glitch that can be smoothed
with filters. The glitch in many digital
VGAs has no such remedy.

Gain and Temperature
Compensation

Many communication receivers re­quire frequent gain optimization, but
others are designed with over-perform­ing ADCs that can tolerate moderate
signal amplitude variation and avoid
much of the AGC hardware problem.
However, even these “fixed gain”
system blocks often require a gain

30

30

adjustment to compensate gain drift
overtemperature and any cumulative
gain tolerance of the other compo­nents. Several system components are
cascaded to form a chain that usually
includes a VGA to perform a one-time
adjustment of gain and temperature
slope to compensate the tolerances and
slopes of the other components. In this
scenario, the required temperature
and compensation information is not
known to the baseband processor or
it is impractical to send this data to a
suitably located VGA.

natural solution for this application
because it can easily interpret the out­put of most temperature transducers
without digitization. Figure 10 shows

An analog-controlled VGA is a

to illustrate the capabilities of the
LTM8040—this frequency is too low
for practical pulse width modulation,
being well within the discrimination
range of the human eye.

The LTM8040 also features a low
power shutdown state. When the
SHDN pin is active low, the input
quiescent current is less than 1µA.

Conclusion

The LTM8040 µModule LED driver
makes it easy to drive LEDs. Its high
level of integration and rich feature set,
including open LED protection, analog
and PWM dimming, save significant
design time and board space.

Figure 13. Gain vs temperature performance
characteristics of the thermistor-based circuit
shown in Figure 12

L

a simple application circuit using a
common PTAT temperature sensor
and an op amp to create the required
–VG signal to adjust room temperature
gain and temperature slope as shown
in Figure 11. If temperature slope
accuracy is only important for T >
0°C, then the same function can be
performed with an inexpensive NTC
thermistor as shown in Figures 12 and

13. Trying doing that with a digitally
controlled VGA!

Conclusion

By combining the advanced SiGe
process with an innovative design, the
LTC6412 offers unparalleled analog
VGA performance at 3.3V. The tiny
16mm² leadless package and minimal

external components produce a cost
effective, fully differential VGA solution
in less than 1cm² of PCB area.

Linear Technology Magazine • September 2009

L

DESIGN IDEAS L

SW CAP

GND

FBP

OUT 2

OUT 1

IN 2

IN 1

LT8415

C1

2.2µF

LOGIC
LEVEL

L1

100µH

22nF

0.1µF

1nF

200pF

V

IN

2.6V to 5V

VCCV

OUT

V

REF

SHDN

137K

887K

34V/0V

34V/0V

L1: COILCRAFT DO2010-104ML

IN1/IN2

1V/DIV

OUT1

10V/DIV

OUT2

10V/DIV

5µs/DIV

IN1/IN2

1V/DIV

OUT1

10V/DIV

OUT2

10V/DIV

5µs/DIV

V

OUT

IN2 OUT2

V

OUT

V

IN

IN1 OUT1

ENABLE

V

OUT

40V MAX

BOOST

CONVERTER

LT8415

Low Power Boost Regulator with Dual Half-Bridge in 3mm × 2mm DFN Drives MEMS and Piezo Actuators

by Jesus Rosales

Introduction

Advances in manufacturing technol­ogy have made it possible for actuators,
sensors, RF relays, and other moveable
parts to be manufactured at a very
small scale. These devices, referred
to as MEMS (micro-electro-mechani­cal systems) or micro-machines, are
finding their way into daily life in
applications unheard of just a few
years ago. MEMS are used in automo­tive, military, medical and consumer
product applications.

Many types of MEMS devices con­sume very little power to operate and
generally require the use of two sup­port circuits, a step-up converter and a
dual half-bridge driver. These support
circuits must be very small and highly
efficient to keep pace with ever-shrink­ing MEMS applications. To this end,
the LT8415 integrates the step-up
converter power switch and diode and
the dual half-bridge driver in a 12-pin,
3mm × 2mm DFN package. Its novel
switching architecture consumes very
little power throughout the load range,

making it an ideal match for driving
low current MEMS.

The LT8415 generates output volt­ages up to 40V from sources ranging
from 2.5V to 16V. The output is then
available for the integrated comple­mentary half-bridge drivers and is
available via OUT1 and OUT2 (see Fig­ure 1). Each half-bridge is made up of
an N-channel MOSFET and a P-chan­nel MOSFET, which are synchronously
controlled by a single pin and never
turn on at the same time. OUT1 and
OUT2 are of the same polarity as IN1
and IN2, respectively. When the part is
turned off, all MOSFETs are turned off,
and the OUT1 and OUT2 nodes revert
to a high impedance state with 20MΩ
pull-down resistors to ground.

2.6V–5V Input to 34V Output
MEMS Driver

Figure 2 shows a MEMS driver that
takes a 2.6V–5V input and produces
a 34V output. This circuit draws very
little source current when the dual
half-bridge is disabled. The input
current is only 320µA at 2.6VIN and
128µA at 5VIN. A logic level signal at
IN1 and IN2 activates the dual half­bridge switches. Figure 3 shows the
turn-on delay and rise time for OUT1
and OUT2 with both half-bridges ac­tivated. Figure 4 shows the turn-off
delay and fall time with the 200pF
and 1nF capacitive loads shown in
Figure 2. See the data sheet details
for measuring delay time.

Figure 1. Simplified block diagram of the
LT8415

Linear Technology Magazine • September 2009

Figure 2. 2.6V–5V input to 34V dual half-bridge boost converter

Figure 3. Turn-on delay and rise time for OUT1
and OUT2

Figure 4. Turn-off delay and fall time for OUT1
and OUT2

3131

L DESIGN FEATURES

SW CAP

GND

FBP

OUT 2

OUT 1

IN 2

IN 1

LT8415

2.2µF

LOGIC
LEVEL

L1

100µH

0.1µF

1µF

16V

1.6mA AT 3V

IN

10mA AT 10V

IN

1nF

200pF

V

IN

3V to 10V

V

CCVOUT

V

REF

SHDN

604K

412K

16V/0V

16V/0V

L1: COILCRAFT DO2010-104ML

V

OUT1

(1.8V)

500mV/DIV

V

OUT2

(1.2V)

500mV/DIV

200µs/DIV

V

IN1

V

IN1

V

OUT1

1.8V
4A

V

OUT2

1.2V
4A

V

OUT1

V

OUT3

1V

1.5A

V

IN

5V

V

IN

V

OUT2

22µF

10µF

100µF

4.02k

10k

LTM4615

4.7µF

3.32k

V

IN2

P

GOOD1

P

GOOD2

V

OUT2

FB2

COMP2

V

OUT1

FB1

COMP1
TRACK1
RUN/SS1

LDO_IN

EN3

TRACK2

RUN/SS2

LDO_OUT

FB3

P

GOOD3

100µF

22µF

GND

10k

4.99k

10µF

=2

82µF

+

Figure 5. 3V–10V input to 16V dual half-bridge plus 16V output boost converter

3V–10V Input to 16V Output
MEMS Driver and Bias Supply

Figure 5 shows a 3V–10V input to
16V output converter, where the out­put drives the dual half-bridge and
also provides bias current for other

circuitry. The converter in Figure 2
can be used in a similar fashion, but
the current available at the output is
reduced as the output voltage is in­creased. See the data sheet for details
about maximum output current.

Integrated Resistor Divider

The LT8415 contains an integrated
resistor divider such that if the FBP
pin is at 1.235V or higher, the output
is clamped at 40V. For lower output
voltage levels use R1 and R2, calculat­ing their values as instructed by the
data sheet. This method of setting the
output voltage ensures the voltage di­vider draws minimal current from the
input when the part is turned off.

Conclusion

The LT8415 is an ideal match for
driving low power MEMS. It integrates
a step-up converter power switch
and diode, a complementary dual
half-bridge, and a novel switching
architecture that minimizes power
dissipation.

L

LTM4614/15, continued from page 26

to promote good thermal conductivity.
Figure 10 shows that thermal dissipa­tion is well-balanced between the two
switching regulators.

Output Voltage Tracking

Tracking can be programmed using
the TRACK1 and TRACK2 pins. To
implement coincident tracking, at the
slave’s TRACK pin, divide the master
regulator’s output with a resistor
divider that is the same as the slave
regulator’s feedback divider. Figure 11

32

shows a tracking design and Figure 12
shows the output. V
in master-slave design with both
outputs ramping up coincidently. The
smooth start-up time is attributed to
the soft-start capacitor.

Conclusion

The cumbersome designs typical of
multivoltage regulation are a thing of
the past. The LTM4614 and LTM4615
µModule multiple-output regulators
can be easily fit into space-constrained

Figure 11. Output voltage tracking design example

OUT2

tracks V

OUT1

system boards with far fewer compo­nents than discrete solutions. The
dual-output LTM4614 µModule regu­lator and triple-output LTM4615 are
small in size, have excellent thermal
dissipation and have high efficiency.
Independent input and output voltage
rails give these µModule regulators
unmatched flexibility. They can be
used in a variety of input-output com­binations, including input and output
current sharing, output voltage track­ing, and low noise output.

Figure 12. Start-up waveforms for the circuit
in Figure 11

Linear Technology Magazine • September 2009

L

DESIGN IDEAS L

LTC3529

V

IN

SW

3.3µF

370Ω

Li-Ion

2.5V
TO

4.2V

V

OUT

5V
500mA

4.7µH

FAULT SNSGND

RST V

OUT

SHDN PGND

C

OUT

10µF

ONOFFAUTO-RESTART

ONOFF

+

1 100010010

POWER LOSS (W)

0.9

0.8

0.7

0.6

0

0.5

0.4

0.3

0.2

0.1

POWER LOSS

EFFICIENCY

VIN = 3.6V
INDUCTOR = 4.7µH,
COOPER BUSSMANN SD25-4R7

LOAD CURRENT (mA)

EFFICIENCY (%)

100

90

80

70

0

10

20

30

60

50

40

Synchronous Boost Converter with Fault Handling Generates 5V at 500mA in 1cm2 of Board Space

by Eddy Wells

Introduction

Today’s power supply designs must
meet a number of stringent and
sometimes competing requirements.
In many cases the requirement for a
small solution is at odds with the need
for high conversion efficiency and the
need to safely deal with fault condi­tions. The LTC3529 step-up DC/DC
converter is designed to provide a
“no compromises” design, offering
high efficiency to minimize dissipated
heat and maximize battery life while
still maintaining a small footprint for
size-constrained power applications
requiring a 5V supply.

The LTC3529 can detect a shorted
output condition, disable the IC, and
report the event to a host micropro­cessor. This feature is important for
portable applications where devices
communicate with each other directly,
or system power applications where
voltages on multiple boards must
be monitored and maintained. As
shown in Figure 1, the LTC3529 of­fers a compact and efficient solution
consisting of only three tiny external
components.

Lithium-Ion to 5V, 2.5W
Converter

Figure 2 shows an LTC3529-based
solution for converting from a single
lithium-ion battery or 3.3V board sup­ply to 5V with up to 500mA of load

Figure 1. A tiny (1cm2) yet complete solution
drives USB On-The-Go bus power.

current. Requiring only an inductor
and input/output filter capacitors,
the entire converter occupies only
about 1cm2 of board space. The IC
includes internal compensation, the
output divider, and soft-start circuitry
to minimize external components. In
shutdown, the LTC3529 disconnects
the output from the input and draws
less than 1µA from the source.

In fixed frequency PWM mode, the
efficiency for a typical Li-Ion source to
5V peaks at 92%, as shown in Figure
4, and remains above 80% for load cur­rents greater than 30mA. The LTC3529
delivers up to 500mA of current at a
5V output and is therefore suitable for
both low and high power USB applica­tions. As with any DC/DC converter,
a tradeoff exists between switching

frequency, inductor value, output
capacitance and output ripple.

To allow the use of tiny external
components, the LTC3529 operates
at 1.5MHz and is stable with a 4.7µH
inductor and output capacitances
of 4.7µF (compatible with USB On­The-Go specifications) or greater.
The Li-Ion-to-5V converter in Figure
3 utilizes a 10µF output capacitor,
and exhibits a peak-to-peak output
ripple of only 10mV. Low ESR and ESL
ceramic capacitors (such as X5R) are
recommended for both VIN and V
bypassing.

Fault Detection

The LTC3529 is robust to output short
circuits, a problem that arises as the
terminals of the IC are exposed to the
outside world to facilitate connection
between portable devices or system
board edge connectors. To defend
against output shorts, the LTC3529
shuts down when an excessive current
draw is detected through the internal
MOSFET switches continuously for
15ms.

Figure 4 illustrates the fault han­dling protocol of the LTC3529. Based

continued on page 35

OUT

Linear Technology Magazine • September 2009

Figure 2. Li-Ion to 5V synchronous boost converter

Figure 3. Efficiency for the circuit in Figure 2

3333

L DESIGN IDEAS

POWER LOSS (mW)

EFFICIENCY (%)

LOAD CURRENT (mA)

1001

100

50

50

0

10

60

70

80

90

10

20

30

40

EFFICIENCY

POWER LOSS

V

OUT

= 5V

VIN = 24V

VIN = 10V

VIN = 10V

V

OUT

1V/DIV

V

IN

10V/DIV

50µs/DIV

V

IN

LTC3642-5
RUN
HYST

SW

V

IN

5V TO 45V

C

IN

1µF

C

OUT

10µF

V

OUT

5V
50mA

V

OUT

SS

I

SET

GND

150µH

C

IN

: TDK C3225X7R1H105KT

C

OUT

: MURATA GRM32DR71C106KA01

L1: COILCRAFT LPS6225-154ML

Robust DC/DC Step-Down Converter in 3mm × 3mm DFN Resists 60V Input Surges

Introduction

Industrial and test equipment must
often run on relatively unregulated
9V-to-24V rails that also support high
current and inductive load switching of
electromechanical devices. When such
devices switch on and off, momentary
power surges disrupt power flow,
causing voltage fluctuations and large
overvoltage spikes on the rail.

The L TC3631 , LTC3 63 2 an d
LTC3642 are robust, monolithic DC/
DC step-down solutions that produce
a well-regulated supply even in volatile
voltage environments. All can operate
from a wide input voltage ranges and
sustain repetitive 60V surges (see
Table 1). The output voltage is immune
to large voltage swings in the input
(see Figure 1).

Compact and Easy to Use

The LTC3642 comes in compact 3mm
× 3mm DFN and MS8E packages with
integrated MOSFETs, as shown in
Figure 2. It is extremely easy to use,
requiring no loop compensation. The

3.3V and 5V fixed output versions only
need two capacitors and an inductor
for operation (see Figure 3).

The constant peak switch current
thresholds of these devices inherently
protect them from output short cir­cuits. Moreover, each of these devices
can reduce its peak switch current
threshold such that smaller input and
output capacitors can be used.

When operating with a high input
voltage source, the LTC3642’s RUN
pin can be optionally configured to

Figure 1. The LTC3642 continues to regulate
the output despite a >45V spike on the input.

Figure 3. With the LTC3642EDD-3.3/5 only two capacitors
and an inductor are required for operation

increase its undervoltage lockout
(UVLO). Until the input voltage ex­ceeds the UVLO, the input remains
disconnected from the load. The RUN
pin can be tied directly to the input
voltage and can be used together with
the hysteresis pin to prevent unwanted
UVLO triggering due to noisy input
supplies and high voltage coupling
in harsh environments. When above
the UVLO, the LTC3642 soft starts
its output with an internal 0.75ms
timer. The duration of the soft-start
timer can be increased by adding an

Figure 2. The solution size of LTC3642-3.3/5
in a 3mm × 3mm DFN package

regulator which does not suffer sig­nificant power loss as a result of IR
drop between the input and output.
High efficiency is also achieved with
Burst Mode® operation, which reduces
switching activity at light loads to
minimize switching losses. Figure
4 shows a fairly constant efficiency
curve from light load all the way to
full load. During shutdown, this device
only draws 3µA even at a maximum
input voltage of 45V. With such high
efficiency, the LTC3642 is a good fit
in battery-operated motorized vehicles,

external capacitor in the SS pin.

High Efficiency

Unlike a linear regulator, the LTC3642
is a monolithic synchronous buck

by Chuen Ming Tan

Table 1. Comparison of monolithic wide input range buck regulators

Maximum Output Current 100mA 20mA 50mA

Input Voltage Operating Range 4.5V–45V 4.5V–50V 4.5V–45V

34

34

Input Voltage Abs Max 60V 60V 60V

LTC3631 LTC3632 LTC3642

Figure 4. Efficiency for circuit in Figure 3

Linear Technology Magazine • September 2009

DESIGN IDEAS L

V

IN

5V/DIV

V

OUT

10V/DIV

V

SW

20V/DIV

10ms/DIV

V

IN

LTC3642

RUN

C

IN

1µF

HYST

SW

V

IN

12V

V

OUT

–24V
18mA

V

FB

SS

I

SET

GND

L1

100µH

R1

1.47M

R2

49.9k
CIN: TDK C3225X7R1H105KT
C

OUT

: MURATA GRM32DR71C106KA01

L1: TYCO/COEV DQ6530-101M

C

OUT

10µF

I

OUT

FAULT

V

OUT

10ms/DIV

I

OUT

FAULT

V

OUT

10ms/DIV

Figure 5. Generating a negative 24V output
voltage from a positive 12V input voltage

portable medical instruments and
certain automotive applications.

Positive-to-Negative Converter

The LTC3642 can produce a negative
output voltage from a positive input
voltage without the use of transformers
(see Figure 5). In this configuration,
the LTC3642 actually operates in an
inverting buck-boost mode. Its wide in-

LTC6930, continued from page 23

concern, and extreme accuracy is not
paramount. Such applications include
clocking microprocessors and micro­controllers, acting as a time base for
low speed serial communication pro­tocols such as USB and RS232, digital
audio applications, clocking switching
power supplies and anywhere a general
purpose clock is needed.

Figure 6. The LTC3642’s wide input voltage swing makes it suitable
for generating a negative output from positive input voltage.

put voltage range, up to 45V, provides
sufficient headroom to generate any
negative voltage between –0.8V and
–40.5V. Figure 6 shows LTC3642 pro­ducing a –24V output from a 12V input
supply from start-up. The LTC3642
is inherently stable in this configura­tion with no external compensation
components required.

Conclusion

When comparing clock power dissipa­tion it is important to consider not just
the dissipation of the oscillator itself,
but also how the oscillator’s features
and start-up times effect the dissi­pation of the entire system. Crystal
oscillators not only dissipate more cur­rent than other solutions, but can have

Conclusion

The LTC3642, LTC3631 and LTC3632
are a rugged DC/DC converters for use
in applications where a stable voltage
output must be produced from poorly
regulated high voltage rails. Their
compact size and high efficiency make
them easy to use in a wide variety of low
power applications, including mobile
and battery powered devices.

other start-up and control character­istics that lead to power waste. When
the LTC6930’s on-the-fly frequency
programmability and one-clock-cycle
settling time are considered, it is clear
that it conserves much more system
power than its dissipation specification
would indicate

L

L

LTC3529, continued from page 33

on a pin-selectable setting, the IC can
be configured to either periodically
attempt to power up (RST pin high,
Figure 4a), or remain shut down un­til power is cycled to the device (RST
pin low, Figure 4b). The waveform
indicating the fault condition is seen
at the Fault pin and is produced by
an internal open-drain device whose
input is pulled high in the event of
a fault. The Fault pin can either be
connected to a microprocessor or
drive an LED.

Conclusion

High conversion efficiency and the
ability to detect and handle output
shorts make the LTC3529 an ideal so-

Linear Technology Magazine • September 2009

4a. RST high: converter attempts power-up
every 15ms.

Figure 4. A fault detection mechanism powers down
the converter, providing robustness to output shorts

lution for either peer-to-peer portable
applications or point-of-load board
power with robust fault handling.
The 1.5MHz switching frequency

4b. RST low: converter remains shut down
until power is cycled.

and highly integrated design of the
LTC3529 yield compact solutions with
minimal design effort.

L

3535

L DESIGN IDEAS

SW
V

IN

V

REF

CTRL

SHDN/UVLO

ISP

ISN

FB

OPENLED

ANODE

LT3519

GND

68µH

1M

V

IN

6V TO 30V

243k

137k

100k

1M

1M

29.4k

M1

2.49Ω

4.7µF

1µF

38V LED
100mA

CATHODE

PWM

PWM

V

OUT

SW
V

IN

V

REF

CTRL

SHDN/UVLO

ISP

ISN

FB

OPENLED

ANODE

LT3519

GND

L1A

68µH

C3

2.2µF

1M

V

IN

4V TO 24V

432k

158k

100k

1M

1M

69.8k

M1

R

SENSE

1.68Ω

C2

4.7µF

C1
1µF

16V LED
150mA

CATHODE

PWM

PWM

t

L1B

68µH

t

C1: TDK C3216X7R1H105K
C2: TDK C3216X7R1E475K
C3: TDK C3216X7R1E225K
L1: COILTRONICS DRQ74-680-R (COUPLED INDUCTOR)
M1: VISHAY SILICONIX Si2318DS

NOTE:
V

IN

= 6V RISING TURN ON

V

IN

= 4V FALLING UVLO

V

IN

> 9V FULL LED CURRENT AND FOLDBACK BELOW

V

OUT

18.5V OVERVOLTAGE PROTECTION

V

OUT

5V

PWM

5V/DIV

I

LED

0.1A/DIV

I

L

0.3A/DIV

2µs/DIVV

IN

= 12V

4W LED Driver Includes Power Switch, Compensation Components and Schottky in 16-Pin MSOP

Introduction

As the number of applications for
medium power (1W–4W) LED strings
grows, so does the need for compact, ef­ficient, high performance LED drivers.
The LT3519 LED driver satisfies the
needs of a wide variety of applications,
including LCD displays, automotive
and avionic applications, architec­tural and industrial lighting, portable
projection and scanners. It’s 16-pin
MSOP package includes accurate LED
current regulation, small size, high
efficiency, PWM and analog dimming
for brightness control and open circuit
protection with fault detect.

Easy Layout: Integrated
Power Switch, Compensation
Components and Schottky

The 400kHz LT3519 LED driver fea­tures an integrated 750mA 45V peak
power switch, integrated compensa­tion components and an integrated
low leakage Schottky diode, making
designs simple and small. Despite
this high level of integration, it can be
used in a wide variety of topologies,
including boost, SEPIC, buck mode
or buck-boost mode. For maximum
versatility, the Schottky diode anode
(ANODE) and internal power switch
emitter (SW) pins are separately pinned
out, so a SEPIC coupling capacitor can
be inserted between these two.

The internal compensation compo-

nents are chosen to match the 2.2µF

Figure 2. Integrated PWM dimming yields
1000:1 dimming at 120Hz

36

36

by Keith Szolusha

Figure 1. A 4W boost LED driver with 89% efficiency at 12V

to 4.7µF output capacitors in all of
the topologies mentioned above. The
integrated compensation network

all that is needed to assert the open
collector OPENLED output pin during

a fault.
combined with current mode con­trol yields fast and stable transient
response.

OPENLED detection and fault re­porting are included. A simple resistor
divider sets the overvoltage protection
output voltage in case of an open LED
string and a small pull-up resistor is

Figure 3. A SEPIC LED driver with short-circuit protection

4W Boost LED Driver

The simple boost LED driver in Figure
1 drives up to 38V of LEDs at 100mA
from an automotive input voltage
range. The 400kHz switching frequen­cy is common for automotive, avionic
and industrial solutions; it combines

Linear Technology Magazine • September 2009

IN

continued on page 39

CS, SCK,
SDI, SDO
TO CONTROLLER

R

OUT

1k

V

OUT

0.1µF

5V
L
O
A
D

V

OUT

t7

SENSE

R

OUT

R

IN

LTC6102

R

IN

150Ω

V

SENSE

6mΩ

4V TO 60V

V

+

V

–

OUT

+IN

+

–

–INF

–INS

V

REG

0.1µF

–

+

LTC2460

16-BIT ADC

IN

GND

V

CC

DESIGN IDEAS L

3mm × 3mm, 16-Bit ADC Brings Accurate, Precise High Side Current Sensing to Tight Spaces

Introduction

Power monitoring circuits are increas­ingly used throughout automotive,
industrial, communications and
computing applications as electronics
designers strive to continually im­prove thermal performance, increase
efficiency and generally make their
products more “green.”

The problem is that power monitor­ing always looks like the perfect feature
until space and cost constraints come
into play. Power monitoring is usually
considered an ancillary function, so
its footprint should be as small as
possible to maximize space available
to the main application. The LTC2460
16-bit Delta-Sigma ADC solves the
space and design cost problem when
paired with one of Linear Technology’s
current sense amplifiers, such as the
LTC6102.

The LTC2460 proves that big-fea­ture ADCs can come in tiny packages.
It is available in a 3mm × 3mm DFN
(or a 12-pin MSOP), and integrates a
10ppm/°C precision reference. The
integrated reference paired together
with an extremely easy to drive input
stage (50nA average input current)
makes it possible to use the LTC2460
with little to no support circuitry.

Measuring Power Means
Measuring Current

Measuring power supply input and
output voltages is fairly straightfor­ward, as any voltage can be scaled
with a simple divider or amplifier and
compared to a voltage reference. Cur­rent measurement is generally more
complicated, especially at commonly
used high voltages such as –12V, 24V
and 48V.

To measure current, a small sense
resistor is placed directly in series with
the supply. A current sense amplifier
takes the small voltage drop across this

Linear Technology Magazine • September 2009

Figure 1. A simple and compact high side current sensing solution that combines a high
resolution ∆Σ ADC (LTC2460) with a high precision current sensing amplifier (LTC6102).

sense resistor and sources a propor­tional signal current. If the current is
monitored on a high voltage power sup­ply, an accurate and precise current
sense amplifier, such as the LTC6102,
is required to accurately resolve the
small voltage drop riding on the high
common mode voltage.

Typically the signal current pro­duced by the current sense amp is
converted via a grounded resistor to a
properly scaled voltage, which, in this
case, can be measured directly using
the LTC2460’s easy to drive input.
The 16-bit output data can then be
used to compute power consumption
and efficiency.

mum current out of its OUT pin. This
current is converted into a voltage
across the 1kΩ resistor to ground,
which allows the connected LTC2460
to measure a 0V to 1V input. This input
range spans 80% of the ADC’s input
resolution. Of course, the output of the
current sense amplifier can be scaled
to use as much of the LTC2460’s input
range as needed, while providing for
overrange conditions.

Another advantage to the LTC2460
is the narrow input bandwidth of
approximately 30Hz. This provides
excellent rejection of power supply
ripple noise, and allows accurate
measurement of the DC component
of the current.

Accurate, Precise and Very
Compact High Side Current
Sense Design

Figure 1 shows a 48V, 8A current mea­surement application. The LTC6102
is a precision current sense amplifier
that offers 10µV maximum input off­set voltage, 50nV/°C input offset drift
(maximum), and low 3nA (maximum)
input bias current.

This current sense amplifier has

zero-drift and sources a 1mA maxi-

Conclusion

The LTC2460 and the LTC6102 facili­tate a compact, high resolution, high
accuracy current sense solution. The
LTC2460 is a 16-bit ADC in a tiny
package that includes an integrated
precision reference, while the LTC6102
provides high precision, current mea­surements that in turn can be easily
digitized by the ADC.

by Leo Chen

L

3737

L DESIGN IDEAS

SW2 SGND1 GND1 SGND2 GND2

SW1 CLKIN1 CLKOUT1 CLKIN2 CLKOUT2

V

IN1

SV

IN1

RUN1

PLLLPF1

MODE1

PHMODE1

TRACK1

V

IN2

SV

IN2

RUN2

PLLLPF2

MODE2

PHMODE2

TRACK2

V

OUT1

FB1

ITH1

ITHM1

PGOOD1

BSEL1

MGN1

V

OUT2

FB2

ITH2

ITHM2

PGOOD2

BSEL2

MGN2

R

SET1

10k

V

OUT1

1.2V
8A

R

SET2

4.99k

V

OUT2

1.8V
8A

C3
22µF

C5
100µF

C4
100µF

V

IN

3.3V to 5V

C1
22µF

LTM4616

(15mm s 15mm s 2.8mm)

C2
150µF

C6
22µF

C8
100µF

C7
100µF

LOAD CURRENT (A)

0

EFFICIENCY (%)

50

40

30

80

70

60

9

20

10

0

1 2 3 4 5 6 7 8

100

90

A: 5VIN = 1.2V

OUT

B: 3.3VIN = 1.2V

OUT

C: 5VIN = 1.8V

OUT

D: 3.3VIN = 1.8V

OUT

A

C

D

B

Dual 8A DC/DC µModule Regulator Is Easily Paralleled for 16A

by Eddie Beville and Alan Chern

Two Independent 8A
Regulator Systems in a
Single Package

The LTM4616 is a dual input, dual
output DC/DC µModule regulator in a
15mm × 15mm × 2.8mm LGA surface
mount package. Only a few external
components are needed since the
switching controller, MOSFETs, induc­tor and other support components are
integrated within the tiny package.

Both regulators feature an input

supply voltage range of 2.375V to

5.5V and an adjustable output voltage
range of 0.6V to 5V with up to 8A of
continuous output current (10A peak).
For higher output current designs, the
LTM4616 can operate in a 2-phase
parallel mode allowing the part to
deliver a total output current of 16A.
The default switching frequency is
set to 1.5MHz, but can be adjusted
to either 1MHz or 2MHz via the
PLLLPF pins. Moreover, CLKIN can be
externally synchronized from 750kHz
to 2.25MHz. The device supports
output voltage tracking for supply rail
sequencing. Safety features include
protection against short circuit,
overvoltage and thermal shutdown
conditions.

Simple and Efficient

The LTM4616 can be used as com­pletely independent dual switching
regulators with different inputs and
outputs or paralleled to provide a
single output. Figure 1 shows a typical
design for a 5V common input and two
independent outputs, 1.8V and 1.2V.
Figure 2 shows the efficiency of the
circuit at both 5V and 3.3V inputs.

Few external components are needed
since the integrated output capacitors
can accommodate load steps to the full
8A. Each output voltage is set by a single
set resistor from FB1 (or FB2) to GND.
In parallel operation, the FB pins can
be tied together with a single resistor
for adjustable output voltage.

one another, making the top value
5k.

to CLKIN2 when operating from a
single input voltage. This minimizes
the input voltage ripple by running
the two regulators out of phase with
each other. If more than 16A output
current is required, then multiple
LTM4616 regulators can be configured
for multiphase operation with up to 12
phases via the PHMODE pin. Figure
4 shows the expected efficiency of
the parallel system at 5V and 3.3V
inputs to 1.8V output. Note that the

Parallel Operation for
Increased Output Current

You can double the maximum output
current to 16A by running the two
outputs in parallel as shown in Figure

3. Note that the FB pins share a single
voltage-set feedback resistor that is
half the value of the feedback resistor
in the usual two output configuration.
This is because the internal 10k top
feedback resistors are in parallel with

It is preferred to connect CLKOUT1

Figure 2. LTM4616 efficiency: dual output

38

38

Figure 1. Dual output LTM4616 for a single 3.3V to 5V input, independent 1.8V and 1.2V outputs at 8A each

Linear Technology Magazine • September 2009

two regulators drive equal output cur-

GND

CLKOUT1 CLKIN2

V

IN1

SV

IN1

RUN1

PLLLPF1

MODE1

PHMODE1

TRACK1

V

IN2

SV

IN2

RUN2

MODE2

PHMODE2

TRACK2

V

OUT1

FB1

ITH1

ITHM1

MGN1

V

OUT2

FB2

ITH2

ITHM2

MGN2

V

OUT

1.8V
16A MAX

C3
22µF
x2

C4
100µF
x4

V

IN

3.3V to 5V

C1
22µF
x2

C2
150µF

R

SET

2.49k

LTM4616

(15mm s 15mm s 2.8mm)

LOAD CURRENT (A)

0

EFFICIENCY (%)

50

40

30

80

70

60

18

20

10

0

2 4 6 8 10 12 14 16

100

90

3.3V

IN

5V

IN

0A

TIME (2ms/DIV)

OUTPUT CURRENT (2A/DIV)

0A

I

OUT1

I

OUT2

rent even during soft-start, as shown
in Figure 5.

Conclusion

Whether you require a single 16A high
current output or dual 8A outputs with
sequencing, the LTM4616 provides a
simple and efficient solution.

L

Figure 3. LTM4616 with 16A parallel operation

Figure 5. Balanced current sharing for even
heat dissipation [5VIN to 1.8V

DESIGN IDEAS L

at 16A]

OUT

LT3519, continued from page 36

high efficiency, small inductor and
capacitor size, and high PWM dimming
capability while avoiding frequencies
in the AM broadcast band. A small
inductor with about 750mA satura­tion current rating, a few ceramic
capacitors and several tiny resistors
are all that are needed to complete the
design. As shown in Figure 2, the tiny
PWM dimming MOSFET can be used
to provide over 1000:1 pwm dimming
at 120Hz using the integrated LT3519
PWM dimming architecture and an
extremely low leakage integrated
Schottky diode.

A 1000:1 dimming ratio at 120Hz
is exceptionally high for a 400kHz
switching regulator. It can be tempt­ing to bump up the dimming ratio by
choosing a higher frequency driver,
since in general, higher switching
frequency corresponds to higher
PWM dimming ratios. In this case,
avoiding the AM band means jumping

Linear Technology Magazine • September 2009

Figure 4. Efficiency: single 1.8V output

to 2MHz, which in the end reduces
the maximum duty cycle and the
efficiency. The 400kHz switching
frequency of the LT3519 does what
2MHz converters cannot do: it provides
high duty cycle for operation down to
6VIN with 38V

and as high as 89%

LED

efficiency at 12VIN. If PWM dimming
is not needed, the MOSFET M1 can
be removed and the analog dimming
(CTRL) pin can be used to adjust the
regulated LED current below 100mA
for simple brightness control.

2.4W SEPIC LED Driver

When the LED string voltage is within
the input rail voltage range, a SEPIC
topology is called for. The SEPIC pro­duces a high PWM dimming ratio and
also gives short-circuit protection. The
SEPIC in Figure 3 drives 16V LEDs at
150mA from a 4V to 24V input range.
Since the anode of the integrated catch
diode (ANODE) is made available at

a pin independent of the npn power
switch emitter (SW), the coupling
capacitor is easily inserted between
the two. The maximum voltage that
the SW pin sees is a little above the
input voltage plus the output voltage,
so the 45V 750mA integrated power
switch is a perfect match for these
specifications.

Conclusion

The 400kHz LT3519 is a 4W LED
driver that integrates a number of
required components, including a 45V,
750mA power switch, a low leakage
Schottky diode and compensation
components. It also features PWM
dimming, overvoltage protection and
OPENLED fault detection, making it a
small, simple, and efficient choice for
automotive, avionic, industrial and
other LED driver applications.

L

3939

L DESIGN IDEAS

V

IN

RUN/SS

SHARE

RT ADJ

V

OUT

GND

LTM8025

VIN*

22V TO 36V

*RUNNING VOLTAGE RANGE. PLEASE REFER TO
APPLICATIONS INFORMATION SECTION FOR START-UP DETAILS

V

OUT

12V AT 3A

47.5k

34.8k

4.7µF

22µF

PGOOD

SYNC

AUX

BIAS

OUTPUT CURRENT (mA)

40

80

60

70

100

90

50

EFFICIENCY(%)

0 500 1000 1500 2000 2500 3000

VIN = 24V

V

IN

RUN/SS

SHARE

RT ADJ

V

OUT

GND

LTM8025

V

IN

3.6V TO 24V

V

OUT

1.8V AT 3A

147k

383k

10µF 300µF

PGOOD

SYNC

AUX

BIAS

BIAS

AUX

V

OUT

V

IN

GND GND

GND

THERMAL VIAS TO GND

R

T

R

ADJ

PGOOD

C

IN

C

OUT

SYNC
SHDN

Self-Contained 3A µModule Buck Regulator Produces 0.8V–24V Output from 3.6V–36V Input

Introduction

Ask any group of engineers, “What
would you do with a 3A DC/DC con­verter?” and you will probably get a
wide range of answers—from powering
a DSP rail at 1.8V to running a bank
of 24V switching I/O. Typically, these
two particular applications would
require completely different DC/DC
controller ICs and topologies. How­ever, the LTM8025 µModule DC/DC
converter can satisfy the requirements
of these and just about any other 3A
applications.

The LTM8025 3A µModule DC/DC
buck converter operates from 3.6V to
36V inputs to produce output voltages
as low as 0.8V and as high as 24V.
Furthermore, the LTM8025 features
single cycle Burst Mode® operation,
so it is able to handle a wide range
of load currents, from no load to 3A,
with minimum ripple.

Figure 1. Layout is easy with the LTM8025.

The LTM8025 integrates the con­troller, control circuitry, inductor,
power switch and rectifier all into a
single IC form factor 15mm × 9mm ×

4.32mm package. This LGA package is
RoHS (e3) compliant, and features gold
pads for easy assembly in both leaded
and unleaded solder processes.

by David Ng

Easy Layout

The LTM8025’s high level of integration
simplifies the design of just about any
3A power supply. Just add two resis­tors, input and output capacitance to
make a complete power supply. Layout
is easy, as shown in Figure 1. Figures
2 and 3 show the schematic and ef­ficiency of the LTM8025 producing
12V bus power from a 24V source,
while Figure 4 shows the LTM8025
producing 1.8V from an input range
of 3.6V to 36V.

Versatile Feature Set

The LTM8025 may be operated over
a wide frequency, from 200kHz to

2.4MHz, and may be synchronized to
an external clock source through the
SYNC pin. The LTM8025 start-up is
controlled through its RUN/SS pin,
which also serves to put the part into

continued on page 42

Figure 2. A complete 12V at 3A power supply requires
only the LTM8025, two capacitors and two resistors.

40

40

Figure 3. The LTM8025 boasts high efficiency.

Figure 4. The LTM8025 can produce low voltages from a wide input range.

Linear Technology Magazine • September 2009

New Device Cameos

NEW DEVICE CAMEOS L

16-Bit Quad SPI DAC
Achieves ±1LSB INL & DNL
with Software-Programmable
Unipolar & Bipolar Outputs

The LTC2754-16 is a quad 16-bit cur ­rent output digital-to-analog converter
(DAC) that achieves ±1LSB integral
nonlinearity (INL) and differential non­linearity (DNL). All four DACs can be
software programmed or pin-strapped
for one of six unipolar or bipolar out­put ranges via a simple 4-wire serial
interface. Software programmability
eliminates the need for expensive
precision resistors, gain stages and
manual jumpers. The LTC2754-16’s
precision DC specifications and flex­ible SoftSpan™ output configurability
make it ideal for multichannel data
acquisition modules and automated
test equipment. A pin- and software­compatible 12-bit option is also
available, making it easy to transition
between different resolutions in the
end-product.

The LTC2754-16 is capable of
producing six unique software-pro­grammable unipolar and bipolar
output ranges up to ±10V. The six
SoftSpan output voltage ranges in­clude two unipolar ranges (0V to 5V, 0V
to 10V) and four bipolar ranges (±10V,
±5V, ±2.5V, –2.5V to +7.5V). Voltage
controlled offset and gain adjustment
pins are also included for each DAC,
making it possible to fine-tune each
DAC output. The LTC2754-16 outputs
any of the six SoftSpan ranges while
operating from a single 2.7V to 5.5V
supply and drawing only 1µA maxi­mum supply current.

The LTC2754-16 also offers very
good AC specifications, including
full-scale settling time of only 2µs
and low glitch impulse of 0.26nV•s
with a 3V supply or 1.25nV•s with a
5V supply .

The LTC2754-16’s 2MHz mul­tiplying bandwidth and good AC
specifications are key for applications
such as waveform generation. Fast
settling and low glitch reduce the har­monic distortion, making it possible to
produce higher frequency, lower noise

output waveforms. The LTC2754-16’s
serial interface operates at clock rates
up to 40MHz and allows readback of
any internal register, as well as the
DAC output span setting.

The LTC2754-12 is a pin-compat­ible 12-bit device, with both 16-bit
and 12-bit versions available in 7mm
× 8mm QFN-52 packages. The serial
LTC2754 joins a family of quad, dual
and single DACs (LTC2755/LTC2753/
LTC2751) that communicate via paral­lel I/O. The entire family is available
in commercial and industrial tempera­ture ranges.

Ultralow Power Supervisor

The LTC2935 is an ultralow power
voltage supervisor that features sys­tem initialization, power-fail warning
and reset generation functions. Low
quiescent current (500nA) makes the
LTC2935 an ideal choice for battery
operated applications.

The reset output (RST) holds a
system in reset during low battery
conditions. The reset output pulls
high once the supervised voltage has
been in compliance for 200ms. The
power fail output (PFO) provides an
early warning of impending battery
failure. Supervisor accuracy is ±1.5%
over the full temperature range (–40°C
to 85°C).

Three binary threshold-select inputs
configure one of eight integrated reset
thresholds ranging from 1.6V to 3.45V
in fixed increments. The LTC2935-1
and LTC2935-2 incorporate thresh­olds tailored for lithium-ion battery
applications while the LTC2935-3 and
LTC2935-4 incorporate lower voltage
thresholds tailored for alkaline battery
applications.

The LTC2935 voltage comparators
apply hysteresis during power-up to
prevent load step oscillations. Load
steps can cause battery voltage to drop
due to internal battery resistance. The
LTC2935 requires the input voltage to
exceed the configured reset threshold
by 5% and the power-fail threshold
by 2.5% during power-up. The volt­age comparators reduce the monitor

thresholds back to the configured
settings after passing the turn-on
thresholds.

A manual reset can be invoked
at any time through a pushbutton
switch connected to the manual
reset input (MR). Outputs RST and
PFO are available with open-drain
(LTC2935-1, LTC2935-3) or active
pull-up (LTC2935-2, LTC2935-4)
circuits. The supervisor is available in
a compact 8-lead 2mm × 2mm DFN
and TSOT-23.

Surge Stopper with
Fault Latchoff

The LT4356-2 and LT4356-3 are surge
stoppers that protect loads from high
voltage transients. They regulate the
output during an over voltage event
by controlling the gate of an external
N-channel MOSFET. The output is
limited to a safe value, thereby allowing
the loads to continue functioning. The
current is also monitored through a
sense resistor and limited to 50mV.

A fault timer is started in the event
of either type of fault, voltage or cur­rent, and the pass transistor is turned
off if the condition persists. After a
cool down period set by the timer ca­pacitor, the MOSFET turns back on
for LT4356-2. For the LT4356-3, the
pass transistor is latched off after the
fault timer has expired. Toggling the
SHDN pin resets the part and allows
the MOSFET to turn back on.

The LT4356 operates over a wide
supply range of 4V to 80V and wide
temperature range of –45°C to 125°C,
making it suitable for automotive
applications. In a reverse battery
condition, the VCC, SNS, and SHDN
pins can be pulled to 60V below the
GND potential without damage. The
LT4356-2 also has an auxiliary ampli­fier that is active during shutdown,
allowing it to continue monitoring the
input supply or keep the output alive
to the load. On the other hand, the
LT4356-3 shuts down to a low current
mode, 7µA, with all the functional
circuitry turned off.

Linear Technology Magazine • September 2009

4141

L NEW DEVICE CAMEOS

V

IN

RUN/SS

SHARE

RT ADJ

V

OUT

GND

LTM8025

VIN*

4.1V TO 36V

V

OUT

2.5V AT 5.6A

3V

137k

113k

PGOOD

SYNC

AUX

BIAS

V

IN

RUN/SS

SHARE

RT ADJ

V

OUT

GND

LTM8025

OPTIONAL

SYNC

137k

2.2µF

2.2µF

300µF

PGOOD

SYNC

AUX

BIAS

*RUNNING VOLTAGE RANGE. PLEASE REFER TO APPLICATIONS INFORMATION
SECTION FOR START-UP DETAILS
NOTE: SYNCHRONIZE THE TWO MODULES TO AVOID BEAT FREQUENCIES,
IF NECESSARY. OTHERWISE, TIE EACH SYNC TO GND

The LT4356-2 is offered in 12-pin
DFN and 16-pin SO packages, while
the LT4356-3 is available in 12-pin
DFN, 16-pin SO, and 10-pin MSOP
packages.

Dual Output Synchronous
DC/DC Controller Draws Only
170µA in Battery-Powered
Systems

The LTC3868/-1 is a low quiescent
current, 2-phase dual output synchro­nous step-down DC/DC controller.
The LTC3868/-1 draws only 170µA
with one output active and only 300µA
when both outputs are active, making
it ideal for battery-powered applica­tions. With both outputs shut down,
the LTC3868/-1 draws only 8µA. The
LTC3868/-1 has an input supply
range of 4V to 24V and each output can
be set from 0.8V to 14V at output cur­rents up to 20A. With efficiency as high
as 95%, a LTC3868/-1 based DC/DC
converter is well suited for powering
industrial and medical devices, along
with portable instruments, notebook
and netbook computers.

The LTC3868/-1 operates with
a user -adjustable, fixed frequency
between 50kHz and 900kHz, and

can be synchronized to an external
clock from 75kHz to 850kHz using
its phase-locked-loop (PLL). The user
can select from continuous operation,
pulse-skipping and low ripple Burst
Mode operation during light loads.
These parts also safely start up with
a prebiased load by powering up and
down in pulse-skipping mode.

The LTC3868/-1’s 2-phase op­eration reduces input capacitance
requirements and its current mode
architecture provides easy loop com­pensation and fast transient response.
Both outputs have adjustable soft­start to control the turn-on time, and
the output overload protection feature
latches off the converter until the input
voltage is recycled. The LTC3868/
LTC3868-1 also features a tight ±1.5%
reference voltage accuracy over a
–40°C to 85°C operating temperature
range. The LTC3868 is the fully fea­tured part with additional functions
beyond the LTC3868-1 including a
clock out, phase modulation, two
power good outputs and adjustable
current limit.

The LTC3868 is offered in a 32-lead
5mm × 5mm QFN package and the

LTC3868-1 in a 28-pin SSOP or 4mm
× 5mm QFN-28 packages.

Boost & Inverting DC/DC
Converter for Active Matrix
OLED & CCD Bias

The LT3582, LT3582-5 and LT3582­12 dual channel DC/DC converters
that deliver both positive and negative
outputs required in many biasing ap­plications such as active matrix OLED
(organic light-emitting diode) displays
as well as CCD (charge coupled device)
applications. The LT3582/-5/-12 offer
an I2C interface that can dynamically
program output voltages, power se­quencing and output voltage ramps as
the application requires. Alternatively,
these parameters can be set in manu­facturing and made permanent via
the built in nonvolatile OTP (one time
programmable) memory. The LT3582’s
positive output voltage can be set be­tween 3.2V and 12.775 in 25mV steps,
whereas the negative output can be set
between –1.2V and –13.95V in 50mV
steps. The LT3582-5 and LT3582-12
are pre-configured with ±5V and ±12V
outputs respectively, useful in many
signal conditioning applications.

L

LTM8025, continued from page 40

a low power off state in which the part
draws less than 1µA. Furthermore,
the part comes with a PGOOD pin to
indicate that the output is within 90%
of its target voltage.

Parallel Multiple LTM8025s
for High Current Capability

The LTM8025 is equipped with a
SHARE pin to allow parallel operation
for applications requiring more than
3A load current. Figure 5 shows two
synchronized LTM8025’s providing

2.5V

at 5.6A.

OUT

Conclusion

The highly versatile LTM8025 3A
µModule DC/DC buck converter is
easy to use and fits just about any
step-down regulator need. Its wide
input and output ranges and high level
of integration reduce design effort and
associated costs.

L

Figure 5. Two LTM8025s can be operated in parallel for more than 3A load current.

42

42

Linear Technology Magazine • September 2009

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LTspice® IV (www.linear.com/ltspice) — LTspice is a
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• Powerful general purpose Spice simulator with
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• Complete and easy to use schematic capture and
waveform viewer.

• Macromodels for most of Linear Technology’s
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• Ready to use demonstration circuits for over one hun­dred of Linear Technology’s most popular products.

FilterCAD® — FilterCAD 3.0 is a computer-aided design
program for creating filters with Linear Technology’s
filter ICs.

Noise Program — This program allows the user to cal­culate circuit noise using Linear Technology op amps to
determine the best op amp for low noise applications.

SPICE Macromodel Library — The Library includes
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SPICE simulation package.

Linear Technology Magazine • September 2009

4343

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Linear Technology Magazine • September 2009