LT1934/LT1934-1
BOOST
V
IN
LT1934
SHDN
1934 TA01
C2
2.2μF
0.22μF
10pF
C1: SANYO 4TPB100M
C2: TAIYO YUDEN GMK325BJ225MN
D1: ON SEMICONDUCTOR MBR0540
D2: CENTRAL CMDSH-3
L1: SUMIDA CDRH4D28-470
V
OUT
3.3V
250mA
D2
604k
1M
L1
47μH
D1
V
IN
4.5V TO 34V
ON OFF
SW
FB
GND
C1
100μF
+
LOAD CURRENT (mA)
60
EFFICIENCY (%)
70
80
90
100
0.1 10 100
1934 TA02
50
1
LT1934
V
IN
= 12V
V
OUT
= 5V
V
OUT
= 3.3V
Micropower Step-Down
Switching Regulators
in ThinSOT and DFN
FEATURES
n
Wide Input Voltage Range: 3.2V to 34V
n
Micropower Operation: IQ = 12μA
n
5V at 250mA from 6.5V to 34V Input (LT1934)
n
5V at 60mA from 6.5V to 34V Input (LT1934-1)
n
3.3V at 250mA from 4.5V to 34V Input (LT1934)
n
3.3V at 60mA from 4.5V to 34V Input (LT1934-1)
n
Low Shutdown Current: < 1μA
n
Low V
n
Low Profi le (1mm) SOT-23 (ThinSOTTM) and
Switch: 200mV at 300mA
CESAT
(2mm × 3mm × 0.8mm) 6-Pin DFN Package
APPLICATIONS
n
Wall Transformer Regulation
n
Automotive Battery Regulation
n
Standby Power for Portable Products
n
Distributed Supply Regulation
n
Industrial Control Supplies
DESCRIPTION
The LT®1934 is a micropower step-down DC/DC converter
with internal 400mA power switch, packaged in a low
profi le (1mm) ThinSOT. With its wide input range of 3.2V
to 34V, the LT1934 can regulate a wide variety of power
sources, from 4-cell alkaline batteries and 5V logic rails
to unregulated wall transformers and lead-acid batteries.
Quiescent current is just 12μA and a zero current shutdown mode disconnects the load from the input source,
simplifying power management in battery-powered systems. Burst Mode
power switch result in high effi ciency over a broad range
of load current.
The LT1934 provides up to 300mA of output current. The
LT1934-1 has a lower current limit, allowing optimum
choice of external components when the required output
current is less than 60mA. Fast current limiting protects
the LT1934 and external components against shorted
outputs, even at 34V input.
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
Burst Mode is a registered trademark of Linear Technology Corporation.
ThinSOT is a trademark of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
®
operation and the low drop internal
TYPICAL APPLICATION
3.3V Step-Down Converter
Effi ciency
1934fd
1
LT1934/LT1934-1
ABSOLUTE MAXIMUM RATINGS
Input Voltage (VIN) ................................................... 34V
BOOST Pin Voltage ................................................. 40V
BOOST Pin Above SW Pin ........................................ 20V
SHDN Pin ................................................................. 34V
FB Voltage .................................................................. 6V
SW Voltage ............................................................... V
PIN CONFIGURATION
TOP VIEW
BOOST 1
GND 2
FB 3
S6 PACKAGE
6-LEAD PLASTIC TSOT-23
T
= 125°C, θJA = 250°C/ W, θJC = 102°C/W
JMAX
6 SW
5 V
IN
4 SHDN
(Note 1)
Operating Temperature Range (Note 2)
LT1934E/LT1934E-1 .............................– 40°C to 85°C
LT1934I/LT1934I-1 .............................– 40°C to 125°C
Maximum Junction Temperature .......................... 125°C
Storage Temperature Range ...................– 65°C to 150°C
Lead Temperature (Soldering, 10 sec)
IN
TSOT-23 ............................................................ 300°C
TOP VIEW
6
1
BOOST
SW
2
V
3
IN
6-LEAD (2mm s 3mm) PLASTIC DFN
EXPOSED PAD (PIN 7) IS GND, MUST BE SOLDEDED TO PCB
θJA = 73.5°C/ W, θJC = 12°C/W
7
DCB PACKAGE
FB
5
GND
SHDN
4
ORDER INFORMATION
LEAD FREE FINISH TAPE AND REEL S6 PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT1934ES6#PBF LT1934ES6#TRPBF LTXP 6-Lead Plastic TSOT-23 –40°C to 85°C
LT1934ES6-1#PBF LT1934ES6-1#TRPBF LTF8 6-Lead Plastic TSOT-23 –40°C to 85°C
LT1934IS6#PBF LT1934IS6#TRPBF LTAJB 6-Lead Plastic TSOT-23 –40°C to 85°C
LT1934IS6-1#PBF LT1934IS6-1#TRPBF LTAJC 6-Lead Plastic TSOT-23 –40°C to 85°C
LT1934IDCB#PBF LT1934IDCB#TRPBF LCFZ
LT1934EDCB#PBF LT1934EDCB#TRPBF LCFZ
LT1934IDCB-1#PBF LT1934IDCB-1#TRPBF LDHC
LT1934EDCB-1#PBF LT1934EDCB-1#TRPBF LDHC
LEAD BASED FINISH TAPE AND REEL S6 PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT1934ES6 LT1934ES6#TR LTXP 6-Lead Plastic TSOT-23 –40°C to 85°C
LT1934ES6-1 LT1934ES6-1#TR LTF8 6-Lead Plastic TSOT-23 –40°C to 85°C
LT1934IS6 LT1934IS6#TR LTAJB 6-Lead Plastic TSOT-23 –40°C to 85°C
LT1934IS6-1 LT1934IS6-1#TR LTAJC 6-Lead Plastic TSOT-23 –40°C to 85°C
LT1934IDCB LT1934IDCB#TR LCFZ
LT1934EDCB LT1934EDCB#TR LCFZ
LT1934IDCB-1 LT1934IDCB-1#TR LDHC
LT1934EDCB-1 LT1934EDCB-1#TR LDHC
Consult LTC Marketing for parts specifi ed with wider operating temperature ranges. *The temperature grade is identifi ed by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
This product is only offered in trays. For more information go to: http://www.linear.com/packaging/
6-Lead (2mm × 3mm) Plastic DFN
6-Lead (2mm × 3mm) Plastic DFN
6-Lead (2mm × 3mm) Plastic DFN
6-Lead (2mm × 3mm) Plastic DFN
6-Lead (2mm × 3mm) Plastic DFN
6-Lead (2mm × 3mm) Plastic DFN
6-Lead (2mm × 3mm) Plastic DFN
6-Lead (2mm × 3mm) Plastic DFN
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
–40°C to 125°C
1934fd
2
LT1934/LT1934-1
ELECTRICAL CHARACTERISTICS
The ● denotes specifi cations which apply over the full operating
temperature range, otherwise specifi cations are at T
= 25°C. VIN = 10V, V
A
SYMBOL CONDITIONS MIN TYP MAX UNITS
Undervoltage Lockout
–40°C ≤ T
–40°C ≤ T
Quiescent Current V
= 1.3V
FB
–40°C ≤ T
–40°C ≤ T
V
= 0V 0.01 2 μA
SHDN
FB Comparator Trip Voltage V
Falling –40°C ≤ TA ≤ 85°C
FB
–40°C ≤ T
FB Comparator Hysteresis 10 mV
FB Pin Bias Current VFB = 1.25V –40°C ≤ TA ≤ 85°C
–40°C ≤ T
FB Voltage Line Regulation 4V < V
Switch Off Time V
Maximum Duty Cycle V
< 34V 0.007 %/V
IN
> 1V
FB
V
= 0V
FB
= 1V –40°C ≤ TA ≤ 85°C
FB
–40°C ≤ T
Switch V
CESAT
ISW = 300mA (LT1934, S6 Package)
I
= 300mA (LT1934, DCB Package)
SW
I
= 75mA (LT1934-1, S6 Package)
SW
I
= 75mA (LT1934-1, DCB Package)
SW
Switch Current Limit LT1934
LT1934-1
BOOST Pin Current ISW = 300mA (LT1934)
I
= 75mA (LT1934-1)
SW
Minimum Boost Voltage (Note 3) I
= 300mA (LT1934)
SW
I
= 75mA (LT1934-1)
SW
Switch Leakage Current 2μA
SHDN Pin Current V
SHDN
V
SHDN
= 2.3V
= 34V
SHDN Input Voltage High 2.3 V
SHDN Input Voltage Low 0.25 V
= 15V, unless otherwise noted.
BOOST
≤ 85°C
A
≤ 125°C
A
≤ 85°C
A
≤ 125°C
A
l
l
l
l
l
≤ 125°C
A
l
l
≤ 125°C
A
l
l
≤ 125°C
A
l
1.22
1.21
1.25
1.25
1.4 1.8
85
83
350
90
3
3
3
12
12
12
2
2
12
88
88
200
225
65
70
400
120
8.5
6.0
1.8
1.7
0.5
1.5 5
3.2
3.6
3.6
22
26
26
1.27
1.27
±15
±60
2.3 μs
300
120
490
160
12
10
2.5
2.5
V
V
V
μA
μA
μA
V
V
nA
nA
μs
%
%
mV
mV
mV
mV
mA
mA
mA
mA
V
V
μA
μA
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT1934E and LT1934E-1 are guaranteed to meet performance
specifi cations from 0°C to 85°C. Specifi cations over the –40°C to 85°C
operating temperature range are assured by design, characterization and
correlation with statistical process controls. The LT1934I and LT1934I-1
specifi cations are guaranteed over the –40°C to 125°C temperature range.
Note 3: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the internal power switch.
1934fd
3
LT1934/LT1934-1
TYPICAL PERFORMANCE CHARACTERISTICS
LT1934 Effi ciency, V
100
LT1934
= 5V
V
OUT
L = 47μH
90
= 25°C
T
A
80
70
EFFICIENCY (%)
60
50
0.1 10 100
VIN = 12V
VIN = 24V
1
LOAD CURRENT (mA)
LT1934-1 Effi ciency, V
100
LT1934-1
= 3.3V
V
OUT
L = 100μH
90
= 25°C
T
A
80
70
EFFICIENCY (%)
60
VIN = 12V
= 5V LT1934 Effi ciency, V
OUT
1934 G01
= 3.3V Current Limit vs Temperature Off Time vs Temperature
OUT
VIN = 24V
100
LT1934
= 3.3V
V
OUT
L = 47μH
90
= 25°C
T
A
80
70
EFFICIENCY (%)
60
50
0.1 10 100
500
400
300
200
100
SWITCH CURRENT LIMIT (mA)
VIN = 12V
1
LOAD CURRENT (mA)
LT1934
LT1934-1
= 3.3V LT1934-1 Effi ciency, V
OUT
VIN = 5V
VIN = 24V
1934 G02
100
LT1934-1
V
OUT
L = 150μH
90
= 25°C
T
A
80
70
EFFICIENCY (%)
60
50
0.1
3.0
2.5
2.0
1.5
OFF TIME (μs)
1.0
0.5
= 5V
1 10 100
LOAD CURRENT (mA)
OUT
VIN = 12V
VIN = 24V
= 5V
1934 G03
50
0.1
1 10 100
LOAD CURRENT (mA)
1934 G04
0
–50 –25
Frequency Foldback VFB vs Temperature
16
TA = 25oC
14
12
10
8
6
SWITCH OFF TIME (μs)
4
2
0
0.2 0.4 0.8
0
FEEDBACK PIN VOLTAGE (V)
0.6
1.0
1.2
1934 G07
1.27
1.26
1.25
1.24
FEEDBACK VOLTAGE (V)
1.23
1.22
–50 –25
4
50
25
0
TEMPERATURE (°C)
50
25
0
TEMPERATURE (°C)
0
–50
100
125
1934 G05
75
–25 0
TEMPERATURE (°C)
50 100 125
25 75
1934 G06
SHDN Bias Current
vs SHDN Voltage
2.0
TA = 25°C
1.5
1.0
0.5
SHDN PIN CURRENT (μA)
0
0
100
125
1934 G08
75
2468
SHDN PIN VOLTAGE (V)
10 12
1934 G09
1934fd
TYPICAL PERFORMANCE CHARACTERISTICS
LT1934/LT1934-1
Quiescent Current
vs Temperature
20
15
10
5
QUIESCENT CURRENT (μA)
0
–50
–25 0 25 50
TEMPERATURE (°C)
75 100 125
1934 G10
Undervoltage Lockout
vs Temperature
4.0
3.5
3.0
UVLO (V)
2.5
2.0
–50
–25 0 25 50
TEMPERATURE (°C)
Minimum Input Voltage
V
= 5V
OUT
8
LT1934
= 5V
V
OUT
= 25°C
T
A
BOOST DIODE TIED TO OUTPUT
7
VIN TO START
75 100 125
1934 G11
Minimum Input Voltage
V
= 3.3V
OUT
6.0
LT1934
= 3.3V
V
OUT
= 25°C
T
5.5
A
BOOST DIODE TIED TO OUTPUT
5.0
4.5
4.0
INPUT VOLTAGE (V)
3.5
3.0
VIN TO START
VIN TO RUN
0.1 10 100
1
LOAD CURRENT (mA)
1934 G12
6
VIN TO RUN
0.1 10 100
1
LOAD CURRENT (mA)
PIN FUNCTIONS
(TSOT-23/DFN)
INPUT VOLTAGE (V)
5
4
BOOST (Pin 1/Pin 1): The BOOST pin is used to provide a
drive voltage, higher than the input voltage, to the internal
bipolar NPN power switch.
GND (Pin 2/Pin 5): Tie the GND pin to a local ground plane
below the LT1934 and the circuit components. Return the
feedback divider to this pin.
FB (Pin 3/Pin 6): The LT1934 regulates its feedback pin
to 1.25V. Connect the feedback resistor divider tap to this
pin. Set the output voltage according to V
+ R1/R2) or R1 = R2 (V
/1.25 – 1).
OUT
= 1.25V (1
OUT
SHDN (Pin 4/Pin 4): The SHDN pin is used to put the LT1934
1934 G13
in shutdown mode. Tie to ground to shut down the LT1934.
Apply 2.3V or more for normal operation. If the shutdown
feature is not used, tie this pin to the V
VIN (Pin 5/Pin 3): The V
pin supplies current to the
IN
IN
pin.
LT1934’s internal regulator and to the internal power
switch. This pin must be locally bypassed.
SW (Pin 6/Pin 2): The SW pin is the output of the internal
power switch. Connect this pin to the inductor, catch diode
and boost capacitor.
Exposed Pad (Pin 7, DFN Package): This pin must be
soldered to ground plane.
1934fd
5
LT1934/LT1934-1
BLOCK DIAGRAM
V
V
IN
IN
C2
+
+
–
BOOST
D2
ON OFF
SHDN
V
GND
REF
12μs DELAY
1.8μs DELAY
1.25V
ON TIME
OFF TIME
R2 R1
RSQa
Q
SW
C3
L1
D1
V
OUT
C1
+
ENABLE
–
FEEDBACK
COMPARATOR
FB
1934 BD
6
1934fd
LT1934/LT1934-1
OPERATION
(Refer to Block Diagram)
The LT1934 uses Burst Mode control, combining both low
quiescent current operation and high switching frequency,
which result in high effi ciency across a wide range of load
currents and a small total circuit size.
A comparator monitors the voltage at the FB pin of the
LT1934. If this voltage is higher than the internal 1.25V
reference, the comparator disables the oscillator and power
switch. In this state, only the comparator, reference and
undervoltage lockout circuits are active, and the current into
pin is just 12μA. As the load current discharges the
the V
IN
output capacitor, the voltage at the FB pin falls below 1.25V
and the comparator enables the oscillator. The LT1934
begins to switch, delivering current to the output capacitor. The output voltage rises, and when it overcomes the
feedback comparator’s hysteresis, the oscillator is disabled
and the LT1934 returns to its micropower state.
The oscillator consists of two one-shots and a fl ip-fl op.
A rising edge from the off-time one-shot sets the fl ip-fl op,
which turns on the internal NPN power switch. The switch
remains on until either the on-time one-shot trips or the
current limit is reached. A sense resistor and amplifi er
monitor the current through the switch and resets the
fl ip-fl op when this current reaches 400mA (120mA for
the LT1934-1). After the 1.8μs delay of the off-time oneshot, the cycle repeats. Generally, the LT1934 will reach
current limit on every cycle—the off time is fi xed and
the on time is regulated so that the LT1934 operates at
the correct duty cycle. The 1.8μs off time is lengthened
when the FB pin voltage falls below 0.8V; this foldback
behavior helps control the output current during start-up
and overload. Figure 1 shows several waveforms of an
LT1934 producing 3.3V from a 10V input. When the switch
is on, the SW pin voltage is at 10V. When the switch is
off, the inductor current pulls the SW pin down until it is
clamped near ground by the external catch diode.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate
the bipolar switch for effi cient operation.
If the SHDN pin is grounded, all internal circuits are turned
off and V
current reduces to the device leakage current,
IN
typically a few nA.
V
OUT
50mV/DIV
V
SW
10V/DIV
I
SW
0.5A/DIV
I
LI
0.5A/DIV
5μs/DIV
Figure 1. Operating Waveforms of the LT1934 Converting
10V to 3.3V at 180mA (Front Page Schematic)
1934 F01a
1934fd
7
LT1934/LT1934-1
APPLICATIONS INFORMATION
Which One to Use: LT1934 or LT1934-1?
The only difference between the LT1934 and LT1934-1
is the peak current through the internal switch and the
inductor. If your maximum load current is less than 60mA,
use the LT1934-1. If your maximum load is higher, use
the LT1934; it can supply up to ~300mA.
While the LT1934-1 can’t deliver as much output current,
it has other advantages. The lower peak switch current
allows the use of smaller components (input capacitor,
inductor and output capacitor). The ripple current at the
input of the LT1934-1 circuit will be smaller and may be
an important consideration if the input supply is current
limited or has high impedance. The LT1934-1’s current
draw during faults (output overload or short) and startup is lower.
The maximum load current that the LT1934 or LT1934-1
can deliver depends on the value of the inductor used.
Table 1 lists inductor value, minimum output capacitor
and maximum load for 3.3V and 5V circuits. Increasing
the value of the capacitor will lower the output voltage
ripple. Component selection is covered in more detail in
the following sections.
Minimum Input Voltage
The minimum input voltage required to generate a particular output voltage is determined by either the LT1934’s
undervoltage lockout of ~3V or by its maximum duty cycle.
Table 1
MAXIMUM
LOAD
300mA
250mA
200mA
300mA
250mA
200mA
60mA
45mA
20mA
60mA
45mA
20mA
PART V
LT1934 3.3V 100μH
LT1934-1 3.3V 150μH
OUT
5V 150μH
5V 220μH
L
47μH
33μH
68μH
47μH
100μH
68μH
150μH
100μH
MINIMUM
C
OUT
100μH
47μH
33μH
47μH
33μH
22μH
15μH
10μH
10μH
10μH
4.7μH
4.7μH
The duty cycle is the fraction of time that the internal
switch is on and is determined by the input and output
voltages:
DC = (V
where V
(~0.4V) and V
+ VD)/(VIN – VSW + VD)
OUT
is the forward voltage drop of the catch diode
D
is the voltage drop of the internal switch
SW
(~0.3V at maximum load for the LT1934, ~0.1V for the
LT1934-1). This leads to a minimum input voltage of:
V
IN(MIN)
with DC
= (V
MAX
= 0.85.
+ VD)/DC
OUT
– VD + V
MAX
SW
Inductor Selection
A good fi rst choice for the inductor value is:
L = 2.5 • (V
where I
LIM
+ VD) • 1.8μs/I
OUT
LIM
is the switch current limit (400mA for the
LT1934 and 120mA for the LT1934-1). This choice provides
a worst-case maximum load current of 250mA (60mA for
the LT1934-1). The inductor’s RMS current rating must
be greater than the load current and its saturation current
should be greater than I
. To keep effi ciency high, the
LIM
series resistance (DCR) should be less than 0.3Ω (1Ω
for the LT1934-1). Table 2 lists several vendors and types
that are suitable.
This simple rule may not provide the optimum value for
your application. If the load current is less, then you can
relax the value of the inductor and operate with higher
ripple current. This allows you to use a physically smaller
inductor, or one with a lower DCR resulting in higher
effi ciency. The following provides more details to guide
inductor selection. First, the value must be chosen so that
the LT1934 can supply the maximum load current drawn
from the output. Second, the inductor must be rated appropriately so that the LT1934 will function reliably and
the inductor itself will not be overly stressed.
Detailed Inductor Selection and
Maximum Load Current
The square wave that the LT1934 produces at its switch
pin results in a triangle wave of current in the inductor. The
LT1934 limits the peak inductor current to I
. Because
LIM
8
1934fd
APPLICATIONS INFORMATION
Table 2. Inductor Vendors
VENDOR PHONE URL PART SERIES COMMENTS
Murata (404) 426-1300 www.murata.com LQH3C Small, Low Cost, 2mm Height
Sumida (847) 956-0666 www.sumida.com CR43
Coilcraft (847) 639-6400 www.coilcraft.com DO1607C
Würth
Electronics
(866) 362-6673 www.we-online.com WE-PD1, 2, 3, 4
LT1934/LT1934-1
CDRH4D28
CDRH5D28
DO1608C
DT1608C
the average inductor current equals the load current, the
maximum load current is:
I
OUT(MAX)
where I
= IPK – ΔIL/2
is the peak inductor current and ΔIL is the
PK
peak-to-peak ripple current in the inductor. The ripple
current is determined by the off time, t
= 1.8μs, and
OFF
the inductor value:
= (V
ΔI
L
is nominally equal to I
I
PK
+ VD) • t
OUT
/L
OFF
. However, there is a slight
LIM
delay in the control circuitry that results in a higher peak
current and a more accurate value is:
= I
I
PK
+ 150ns • (VIN – V
LIM
OUT
)/L
These expressions are combined to give the maximum
load current that the LT1934 will deliver:
I
OUT(MAX)
• (V
I
OUT(MAX)
• (V
= 350mA + 150ns • (VIN – V
+ VD)/2L (LT1934)
OUT
= 90mA + 150ns • (VIN – V
+ VD)/2L (LT1934-1)
OUT
)/L – 1.8μs
OUT
)/L – 1.8μs
OUT
The minimum current limit is used here to be conservative.
The third term is generally larger than the second term,
so that increasing the inductor value results in a higher
output current. This equation can be used to evaluate
a chosen inductor or it can be used to choose L for a
given maximum load current. The simple, single equation
rule given above for choosing L was found by setting
= I
ΔI
L
/2.5. This results in I
LIM
OUT(MAX)
~ 0.8I
(ignoring
LIM
the delay term). Note that this analysis assumes that the
inductor current is continuous, which is true if the ripple
current is less than the peak current or ΔI
< IPK.
L
The inductor must carry the peak current without saturating excessively. When an inductor carries too much
current, its core material can no longer generate additional magnetic fl ux (it saturates) and the inductance
drops, sometimes very rapidly with increasing current.
This condition allows the inductor current to increase
at a very high rate, leading to high ripple current and
decreased overload protection.
Inductor vendors provide current ratings for power inductors. These are based on either the saturation current or
on the RMS current that the inductor can carry without
dissipating too much power. In some cases it is not clear
which of these two determine the current rating. Some data
sheets are more thorough and show two current ratings,
one for saturation and one for dissipation. For LT1934 applications, the RMS current rating should be higher than
the load current, while the saturation current should be
higher than the peak inductor current calculated above.
Input Capacitor
Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage ripple
at the LT1934 and to force this switching current into
a tight local loop, minimizing EMI. The input capacitor
must have low impedance at the switching frequency to
do this effectively. A 2.2μF ceramic capacitor (1μF for the
LT1934-1) satisfi es these requirements.
If the input source impedance is high, a larger value capacitor may be required to keep input ripple low. In this
case, an electrolytic of 10μF or more in parallel with a 1μF
ceramic is a good combination. Be aware that the input
1934fd
9
LT1934/LT1934-1
APPLICATIONS INFORMATION
capacitor is subject to large surge currents if the LT1934
circuit is connected to a low impedance supply, and that
some electrolytic capacitors (in particular tantalum) must
be specifi ed for such use.
Output Capacitor and Output Ripple
The output capacitor fi lters the inductor’s ripple current and
stores energy to satisfy the load current when the LT1934
is quiescent. In order to keep output voltage ripple low, the
impedance of the capacitor must be low at the LT1934’s
switching frequency. The capacitor’s equivalent series
resistance (ESR) determines this impedance. Choose one
with low ESR intended for use in switching regulators. The
contribution to ripple voltage due to the ESR is approximately I
• ESR. ESR should be less than ~150mΩ for
LIM
the LT1934 and less than ~500mΩ for the LT1934-1.
The value of the output capacitor must be large enough
to accept the energy stored in the inductor without a large
change in output voltage. Setting this voltage step equal to
1% of the output voltage, the output capacitor must be:
C
OUT
> 50 • L • (I
LIM/VOUT
2
)
For example, an LT1934 producing 3.3V with L = 47μH
requires 33μF. This value can be relaxed if small circuit
size is more important than low output ripple.
Sanyo’s POSCAP series in B-case and C-case sizes
provides very good performance in a small package for
the LT1934. Similar performance in traditional tantalum
capacitors requires a larger package (C- or D-case).
The LT1934-1, with its lower switch current, can use a
B-case tantalum capacitor.
With a high quality capacitor fi ltering the ripple current
from the inductor, the output voltage ripple is determined
by the hysteresis and delay in the LT1934’s feedback
comparator. This ripple can be reduced further by adding
a small (typically 10pF) phase lead capacitor between the
output and the feedback pin.
Ceramic Capacitors
Ceramic capacitors are small, robust and have very low
ESR. However, ceramic capacitors can cause problems
when used with the LT1934.
Not all ceramic capacitors are suitable. X5R and X7R
types are stable over temperature and applied voltage
and give dependable service. Other types (Y5V and Z5U)
have very large temperature and voltage coeffi cients of
capacitance. In the application circuit they may have only
a small fraction of their nominal capacitance and voltage
ripple may be much larger than expected.
Ceramic capacitors are piezoelectric. The LT1934’s switching frequency depends on the load current, and at light
loads the LT1934 can excite the ceramic capacitor at audio
frequencies, generating audible noise. If this is unacceptable, use a high performance electrolytic capacitor at the
output. The input capacitor can be a parallel combination
of a 2.2μF ceramic capacitor and a low cost electrolytic
capacitor. The level of noise produced by the LT1934-1
10
Table 3. Capacitor Vendors
VENDOR PHONE URL PART SERIES COMMENTS
Panasonic (714) 373-7366 www.panasonic.com Ceramic,
Polymer,
Tantalum
Kemet (864) 963-6300 www.kemet.com Ceramic,
Tantalum T494, T495
Sanyo (408) 749-9714 www.sanyovideo.com Ceramic,
Polymer,
Tantalum
Murata (404) 436-1300 www.murata.com Ceramic,
AVX www.avxcorp.com Ceramic,
Tantalum
Taiyo Yuden (864) 963-6300 www.taiyo-yuden.com Ceramic
EEF Series
POSCAP
TPS Series
1934fd
APPLICATIONS INFORMATION
LT1934/LT1934-1
when used with ceramic capacitors will be lower and may
be acceptable.
A fi nal precaution regarding ceramic capacitors concerns
the maximum input voltage rating of the LT1934. A ceramic
input capacitor combined with trace or cable inductance
forms a high quality (under damped) tank circuit. If the
LT1934 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding
the LT1934’s rating. This situation is easily avoided; see
the Hot Plugging Safely section.
Catch Diode
A 0.5A Schottky diode is recommended for the catch
diode, D1. The diode must have a reverse voltage rating
equal to or greater than the maximum input voltage. The
ON Semiconductor MBR0540 is a good choice; it is rated
for 0.5A forward current and a maximum reverse voltage
of 40V.
Schottky diodes with lower reverse voltage ratings usually have a lower forward drop and may result in higher
effi ciency with moderate to high load currents. However,
these diodes also have higher leakage currents. This leakage
current mimics a load current at the output and can raise
the quiescent current of the LT1934 circuit, especially at
elevated temperatures.
BOOST Pin Considerations
Capacitor C3 and diode D2 are used to generate a boost
voltage that is higher than the input voltage. In most cases
a 0.1μF capacitor and fast switching diode (such as the
1N4148 or 1N914) will work well. Figure 2 shows two
ways to arrange the boost circuit. The BOOST pin must
be more than 2.5V above the SW pin for best effi ciency.
For outputs of 3.3V and above, the standard circuit (Figure 2a) is best. For outputs between 2.8V and 3V, use a
0.22μF capacitor and a small Schottky diode (such as the
BAT-54). For lower output voltages the boost diode can be
tied to the input (Figure 2b). The circuit in Figure 2a is more
effi cient because the BOOST pin current comes from a lower
voltage source. You must also be sure that the maximum
voltage rating of the BOOST pin is not exceeded.
The minimum operating voltage of an LT1934 application is limited by the undervoltage lockout (~3V) and by
D2
BOOST
V
IN
V
BOOST
MAX V
V
IN
V
BOOST
MAX V
Figure 2. Two Circuits for Generating the Boost Voltage
LT1934
V
IN
GND
– VSW V
VIN + V
BOOST
D2
BOOST
LT1934
V
IN
GND
– VSW V
2V
BOOST
SW
OUT
OUT
(2a)
SW
IN
IN
(2b)
C3
V
OUT
C3
V
OUT
1934 F02
the maximum duty cycle as outlined above. For proper
start-up, the minimum input voltage is also limited by the
boost circuit. If the input voltage is ramped slowly, or the
LT1934 is turned on with its SHDN pin when the output
is already in regulation, then the boost capacitor may not
be fully charged. Because the boost capacitor is charged
with the energy stored in the inductor, the circuit will rely
on some minimum load current to get the boost circuit
running properly. This minimum load will depend on input
and output voltages, and on the arrangement of the boost
circuit. The minimum load generally goes to zero once the
circuit has started. Figure 3 shows a plot of minimum load
to start and to run as a function of input voltage. In many
cases the discharged output capacitor will present a load
to the switcher which will allow it to start. The plots show
the worst-case situation where V
is ramping very slowly.
IN
Use a Schottky diode (such as the BAT-54) for the lowest
start-up voltage.
At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This
reduces the minimum input voltage to approximately
300mV above V
. At higher load currents, the inductor
OUT
current is continuous and the duty cycle is limited by the
1934fd
11
LT1934/LT1934-1
LOAD CURRENT (mA)
3.5
INPUT VOLTAGE (V)
4.0
4.5
5.0
5.5
6.0
0.1 10 100
1934 G12
3.0
1
LT1934
V
OUT
= 3.3V
T
A
= 25°C
BOOST DIODE TIED TO OUTPUT
VIN TO START
VIN TO RUN
LOAD CURRENT (mA)
5
INPUT VOLTAGE (V)
6
7
8
0.1 10 100
1934 G13
4
1
LT1934
V
OUT
= 5V
T
A
= 25°C
BOOST DIODE TIED TO OUTPUT
VIN TO START
VIN TO RUN
V
IN
BOOST
GND FB
SHDN SW
5
D4
V
IN
4
1
6
23
1M
100k
LT1934
1934 F07
V
OUT
BACKUP
D4: MBR0530
APPLICATIONS INFORMATION
Minimum Input Voltage V
Minimum Input Voltage V
OUT
OUT
= 3.3V
= 5V
to VIN), then the LT1934’s internal circuitry will pull its
quiescent current through its SW pin. This is fi ne if your
system can tolerate a few mA in this state. If you ground
the SHDN pin, the SW pin current will drop to essentially
zero. However, if the V
pin is grounded while the output
IN
is held high, then parasitic diodes inside the LT1934 can
pull large currents from the output through the SW pin
and the V
pin. Figure 4 shows a circuit that will run only
IN
when the input voltage is present and that protects against
a shorted or reversed input.
Figure 4. Diode D4 Prevents a Shorted Input from Discharging
a Backup Battery Tied to the Output; It Also Protects the Circuit
from a Reversed Input. The LT1934 Runs Only When the Input
is Present
Figure 3. The Minimum Input Voltage Depends
on Output Voltage, Load Current and Boost Circuit
maximum duty cycle of the LT1934, requiring a higher
input voltage to maintain regulation.
Shorted Input Protection
If the inductor is chosen so that it won’t saturate excessively, an LT1934 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT1934 is absent. This may occur in battery charging applications or in battery backup systems where a battery
or some other supply is diode OR-ed with the LT1934’s
output. If the V
is held high (either by a logic signal or because it is tied
12
pin is allowed to fl oat and the SHDN pin
IN
PCB Layout
For proper operation and minimum EMI, care must be
taken during printed circuit board layout. Figure 5 shows
the high current paths in the buck regulator circuit. Note
that large, switched currents fl ow in the power switch,
the catch diode (D1) and the input capacitor (C2). The
loop formed by these components should be as small as
possible. Furthermore, the system ground should be tied
to the regulator ground in only one place; this prevents
the switched current from injecting noise into the system
ground. These components, along with the inductor and
output capacitor, should be placed on the same side of
the circuit board, and their connections should be made
on that layer. Place a local, unbroken ground plane below
these components, and tie this ground plane to system
ground at one location, ideally at the ground terminal of
the output capacitor C1. Additionally, the SW and BOOST
nodes should be kept as small as possible. Finally, keep
the FB node as small as possible so that the ground pin
1934fd
LT1934/LT1934-1
SHUTDOWN
VIAS TO LOCAL GROUND PLANE
OUTLINE OF LOCAL GROUND PLANE
V
IN
V
OUT
1934 F06
SYSTEM
GROUND
V
IN
SW
GND
(5a)
V
IN
V
SW
C2 D1 C1
1934 F05
L1
SW
GND
(5c)
V
IN
SW
GND
(5b)
I
C1
APPLICATIONS INFORMATION
Figure 5. Subtracting the Current When the Switch is On (a) from the Current When the Switch is Off (b) Reveals the Path of the High
Frequency Switching Current (c). Keep This Loop Small. The Voltage on the SW and BOOST Nodes Will Also be Switched; Keep These
Nodes as Small as Possible. Finally, Make Sure the Circuit is Shielded with a Local Ground Plane
Figure 6. A Good PCB Layout Ensures Proper, Low EMI Operation
and ground traces will shield it from the SW and BOOST
nodes. Figure 6 shows component placement with trace,
ground plane and via locations. Include two vias near
the GND pin of the LT1934 to help remove heat from the
LT1934 to the ground plane.
Hot Plugging Safely
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT1934 and LT1934-1 circuits. However, these capacitors can cause problems if the LT1934
is plugged into a live supply (see Linear Technology
Application Note 88 for a complete discussion). The low
loss ceramic capacitor combined with stray inductance in
series with the power source forms an under damped tank
circuit, and the voltage at the V
pin of the LT1934 can
IN
ring to twice the nominal input voltage, possibly exceeding
the LT1934’s rating and damaging the part. If the input
supply is poorly controlled or the user will be plugging
the LT1934 into an energized supply, the input network
should be designed to prevent this overshoot.
1934fd
13
LT1934/LT1934-1
APPLICATIONS INFORMATION
Figure 7 shows the waveforms that result when an LT1934
circuit is connected to a 24V supply through six feet of
24-gauge twisted pair. The fi rst plot is the response with
a 2.2μF ceramic capacitor at the input. The input voltage
rings as high as 35V and the input current peaks at 20A.
One method of damping the tank circuit is to add another
capacitor with a series resistor to the circuit. In Figure 7b
CLOSING SWITCH
SIMULATES HOT PLUG
+
LOW
IMPEDANCE
ENERGIZED
24V SUPPLY
10μF
35V
AI.EI.
I
IN
STRAY
INDUCTANCE
DUE TO 6 FEET
(2 METERS) OF
TWISTED PAIR
+
V
IN
LT1934
2.2μF
(7a)
LT1934
2.2μF
an aluminum electrolytic capacitor has been added. This
capacitor’s high equivalent series resistance damps the
circuit and eliminates the voltage overshoot. The extra
capacitor improves low frequency ripple fi ltering and can
slightly improve the effi ciency of the circuit, though it is
likely to be the largest component in the circuit. An alternative solution is shown in Figure 7c. A 1Ω resistor is added
V
IN
10V/DIV
I
IN
10A/DIV
10μs/DIV
0.1μF
1Ω
4.7Ω
LT1934
2.2μF0.1μF
LT1934-1
1μF
LT1934-1
1μF
(7b)
(7c)
(7d)
(7e)
1934 F07
14
Figure 7. A Well Chosen Input Network Prevents Input Voltage Overshoot and
Ensures Reliable Operation When the LT1934 is Connected to a Live Supply
1934fd
APPLICATIONS INFORMATION
LT1934/LT1934-1
in series with the input to eliminate the voltage overshoot
(it also reduces the peak input current). A 0.1μF capacitor
improves high frequency fi ltering. This solution is smaller
and less expensive than the electrolytic capacitor. For high
input voltages its impact on effi ciency is minor, reducing
effi ciency less than one half percent for a 5V output at full
load operating from 24V.
Voltage overshoot gets worse with reduced input capacitance. Figure 7d shows the hot plug response with a 1μF
ceramic input capacitor, with the input ringing above 40V.
The LT1934-1 can tolerate a larger input resistance, such
as shown in Figure 7e where a 4.7Ω resistor damps the
voltage transient and greatly reduces the input current
glitch on the 24V supply.
High Temperature Considerations
The die temperature of the LT1934 must be lower than the
maximum rating of 125°C. This is generally not a concern
unless the ambient temperature is above 85°C. For higher
temperatures, care should be taken in the layout of the
circuit to ensure good heat sinking of the LT1934. The
maximum load current should be derated as the ambient
temperature approaches 125°C.
The die temperature is calculated by multiplying the LT1934
power dissipation by the thermal resistance from junction
to ambient. Power dissipation within the LT1934 can be
estimated by calculating the total power loss from an
effi ciency measurement and subtracting the catch diode
loss. The resulting temperature rise at full load is nearly
independent of input voltage. Thermal resistance depends
on the layout of the circuit board, but a value of 150°C/W
is typical for the TSOT-23 and 75°C/W for the DFN.
The temperature rise for an LT1934 (TSOT-23) producing
5V at 250mA is approximately 25°C, allowing it to deliver
full load to 100°C ambient. Above this temperature the
load current should be reduced. For 3.3V at 250mA the
temperature rise is 15°C. The DFN temperature rise will
be roughly one-half of these values.
Finally, be aware that at high ambient temperatures the
external Schottky diode, D1, is likely to have signifi cant
leakage current, increasing the quiescent current of the
LT1934 converter.
Outputs Greater Than 6V
For outputs greater than 6V, tie a diode (such as a 1N4148)
from the SW pin to V
above V
output circuit in Typical Applications shows the location of
this diode. Also note that for outputs above 6V, the input
voltage range will be limited by the maximum rating of
the BOOST pin. The 12V circuit shows how to overcome
this limitation using an additional Zener diode.
during discontinuous mode operation. The 12V
IN
to prevent the SW pin from ringing
IN
1934fd
15
LT1934/LT1934-1
TYPICAL APPLICATIONS
3.3V Step-Down Converter
D2
V
4.5V TO 34V
ON OFF
V
6.5V TO 34V
ON OFF
BOOST
IN
C2
1μF
V
IN
LT1934-1
SHDN
GND
C1: TAIYO YUDEN JMK316BJ226ML
C2: TAIYO YUDEN GMK316BJ105ML
D1: ZETEX ZHCS400 OR ON SEMI MBR0540
D2: CENTRAL CMDSH-3
L1: COILCRAFT DO1608C-104 OR
WURTH ELECTRONICS WE-PD4 TYPE S
5V Step-Down Converter
BOOST
IN
C2
1μF
V
IN
LT1934-1
SHDN
GND
SW
SW
0.1μF
D1
FB
0.1μF
D1
FB
10pF
10pF
L1
100μH
D2
L1
150μH
1M
604k
1M
332k
V
OUT
3.3V
45mA
+
C1
22μF
1934 TA04
V
OUT
5V
45mA
+
C1
22μF
16
C1: TAIYO YUDEN JMK316BJ226ML
C2: TAIYO YUDEN GMK316BJ105ML
D1: ZETEX ZHCS400 OR ON SEMI MBR0540
D2: CENTRAL CMPD914
L1: COILCRAFT DO1608C-154 OR
WURTH ELECTRONICS WE-PD4 TYPE S
1934 TA05
1934fd
TYPICAL APPLICATIONS
LT1934/LT1934-1
1.8V Step-Down Converter
D2
V
3.6V TO 16V
ON OFF
BOOST
IN
2.2μF
C2
V
IN
LT1934
SHDN
GND
C1: SANYO 2R5TPB100M
C2: TAIYO YUDEN EMK316BJ225ML
D1: ZETEX ZHCS400 OR ON SEMI MBR0540
D2: CENTRAL CMPD914
L1: SUMIDA CR43-330
0.1μF
SW
D1
FB
L1
33μH
147k
332k
+
Loop Powered 3.3V Supply with Additional Isolated Output
10μF
D2
ISOLATED
OUT
3V
3mA
V
14V TO 32V
<3.6mA
D3
+
L1B
50μH
•
C1
L1A
50μH
•
1M
D1
715k
D4
10V
V
IN
LT1934-1
SHDN
BOOST
SW
FB
GND
IN
1μF
390k
10pF
C1
100μF
1934 TA06
+
V
OUT
1.8V
250mA
V
3V
9mA
33μF
OUT
D1: ON SEMICONDUCTOR MBR0540
D2, D3: BAT54
D4: CENTRAL CMPZ5240B
L1: COILTRONICS CTX50-1
ZENER DIODE D4 PROVIDES AN UNDERVOLTAGE LOCKOUT,
REDUCING THE INPUT CURRENT REQUIRED AT START-UP
1934 TA08
1934fd
17
LT1934/LT1934-1
BATTERY VOLTAGE (V)
2.5
CHARGE CURRENT (mA)
200
300
4.5
1934 TA07b
100
0
3
3.5
4
500
400
VIN = 12V
V
IN
= 8V
V
IN
= 24V
TYPICAL APPLICATIONS
Standalone 350mA Li-Ion Battery Charger
D2
V
7V TO 28V
0.1μF
L1
47μH
D1
332k
1M
+
V
SHDN
C5
10μF
IN
BOOST
SW
LT1934
FB
GND
D3
IN
C2
1μF
+
C1: SANYO 6TPB47M (619) 661-6835
C2: TAIYO YUDEN GMK316BJ105ML (408) 573-4150
D1, D3: ON SEMICONDUCTOR MBR0540 (602) 244-6600
D2: CENTRAL CMDSH-3 (516) 435-1110
L1: SUMIDA CR43-470 (847) 956-0667
1k 10k
1k
CHRG
ACPR
C1
47μF
C
TIMER
0.1μF
CHARGE STATUS
AC PRESENT
V
IN
LTC4052
SENSE
GATE
BAT
GNDTIMER
0.047μF
0.022μF
350mA
1-CELL 4.2V
Li-Ion
BATTERY
1934 TA07a
18
12V Step-Down Converter
D2
D4
BOOST
V
IN
15V TO 32V
ON OFF
C2
2.2μF
V
IN
SHDN
C1: KEMET T495D226K020AS
C2: TAIYO YUDEN GMK325BJ225MN
D1: ON SEMI MBR0540
D2, D4: CENTRAL CMPD914
D3: CENTRAL CMPZ5234B 6.2V ZENER
L1: TDK SLF6028T-101MR42
SW
LT1934
FB
GND
0.1μF
100μH
D1
D3
L1
V
OUT
12V
170mA
866k
+
C1
100k
22μF
1934 TA09
1934fd
PACKAGE DESCRIPTION
LT1934/LT1934-1
S6 Package
6-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1636)
0.62
MAX
3.85 MAX
2.62 REF
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.20 BSC
DATUM ‘A’
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
0.95
REF
1.22 REF
1.4 MIN
0.30 – 0.50 REF
2.80 BSC
0.09 – 0.20
(NOTE 3)
1.50 – 1.75
(NOTE 4)
1.00 MAX
0.95 BSC
0.80 – 0.90
PIN ONE ID
2.90 BSC
(NOTE 4)
1.90 BSC
0.30 – 0.45
6 PLCS (NOTE 3)
0.01 – 0.10
S6 TSOT-23 0302 REV B
6-Lead Plastic DFN (2mm × 3mm)
(Reference LTC DWG # 05-08-1715)
0.70 ±0.05
3.55 ±0.05
1.65 ±0.05
(2 SIDES)
2.15 ±0.05
PACKAGE
OUTLINE
0.25 ± 0.05
0.50 BSC
1.35 ±0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (TBD)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
(2 SIDES)
PIN 1 BAR
TOP MARK
(SEE NOTE 6)
DCB Package
2.00 ±0.10
(2 SIDES)
0.200 REF
3.00 ±0.10
(2 SIDES)
0.75 ±0.05
0.00 – 0.05
R = 0.115
R = 0.05
1.65 ± 0.10
(2 SIDES)
TYP
TYP
BOTTOM VIEW—EXPOSED PAD
3
1.35 ±0.10
(2 SIDES)
64
1
0.50 BSC
0.40 ± 0.10
PIN 1 NOTCH
R0.20 OR 0.25
s 45° CHAMFER
0.25 ± 0.05
(DCB6) DFN 0405
1934fd
19
LT1934/LT1934-1
TYPICAL APPLICATION
5V Step-Down Converter
D2
0.1μF
D1
V
6.5V TO 34V
ON OFF
IN
BOOST
V
C2
2.2μF
IN
SHDN
C1: SANYO 6TPB68M
C2: TAIYO YUDEN GMK325BJ225MN
D1: ZETEX ZHCS400 OR ON SEMI MBR0540
D2: CENTRAL CMPD914
L1: SUMIDA CDRH5D28-680
SW
LT1934
FB
GND
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Linear Technology Corporation
20
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
), 1.4MHz, High Effi ciency
OUT
), 100kHz, High Effi ciency
OUT
), 1.25MHz, High Effi ciency
OUT
), 200kHz, High Effi ciency
), 1.25MHz, High Effi ciency
), 200kHz, High Effi ciency
OUT
), 550kHz, Synchronous
), 500kHz, High Effi ciency
), 1.5MHz, Synchronous
), 1.5MHz, Synchronous
), 200kHz, High Effi ciency
OUT
●
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VIN = 3.6V to 25V, V
ThinSOT Package
VIN = 7.4V to 60V, V
S8 Package
VIN = 3V to 25V, V
S8, TSSOP16E Packages
VIN = 5.5V to 60V, V
TSSOP16/E Package
VIN = 3V to 25V; V
MS8/E Packages
VIN = 7.4V to 40V; V
N8, S8 Packages
VIN = 2.7V to 10V; V
MS8 Package
VIN = 2.7V to 10V; V
TSSOP16 Package
V
= 5.5V to 60V, V
IN
TSSOP16/E Package
VIN = 2.7V to 6V, V
ThinSOT Package
VIN = 2.5V to 5.5V, V
ThinSOT Package
VIN = 2.5V to 5.5V, V
MS Package
VIN = 2.5V to 5.5V, V
TSSOP16E Package
VIN = 5.5V to 60V, V
TSSOP16E Package
10pF
L1
68μH
1M
+
332k
= 1.25V, IQ = 1.9mA, I
OUT
= 1.24V, IQ = 3.2mA, I
OUT
= 1.2V, IQ = 1mA, I
OUT
= 1.2V, IQ = 2.5mA, I
OUT
= 1.2V, IQ = 1mA, I
OUT
= 1.24V, IQ = 3.2mA, I
OUT
= 0.8V, IQ = 10μA, I
OUT
= 0.8V, IQ = 15μA, I
OUT
= 1.2V, IQ = 2.5mA, I
OUT
= 0.8V, IQ = 20μA, I
OUT
= 0.6V, IQ = 20μA, I
OUT
= 0.8V, IQ = 60μA, I
OUT
= 0.8V, IQ = 60μA, I
OUT
= 1.2V, IQ = 2.5mA, I
OUT
V
5V
250mA
C1
68μF
1934 TA03
OUT
= <1μA,
SD
= 2.5μA,
SD
= 15μA,
SD
= 25μA,
SD
= 6μA,
SD
= 30μA,
SD
= <1μA,
SD
= <1μA,
SD
= 25μA,
SD
= <1μA,
SD
= <1μA,
SD
= <1μA,
SD
= <1μA,
SD
= 30μA,
SD
LT 0708 REV D• PRINTED IN USA
© LINEAR TECHNOLOGY CORPORATION 2002
1934fd
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