LINEAR TECHNOLOGY LT1738 Technical data

Ultralow Noise DC/DC Controller

FEATURES

■

Greatly Reduced Conducted and Radiated EMI

■

Low Switching Harmonic Content

■

Independent Control of Output Switch Voltage and
Current Slew Rates

■

Greatly Reduced Need for External Filters

■

Single N-Channel MOSFET Driver

■

20kHz to 250kHz Oscillator Frequency

■

Easily Synchronized to External Clock

■

Regulates Positive and Negative Voltages

■

Easier Layout Than with Conventional Switchers

U

APPLICATIO S

■

Power Supplies for Noise Sensitive Communication
Equipment

■

EMI Compliant Offline Power Supplies

■

Precision Instrumentation Systems

■

Isolated Supplies for Industrial Automation

■

Medical Instruments

■

Data Acquisition Systems

LT1738

Slew Rate Controlled

U

DESCRIPTIO

The LT®1738 is a switching regulator controller designed
to lower conducted and radiated electromagnetic interfer­ence (EMI). Ultralow noise and EMI are achieved by
controlling the voltage and current slew rates of an exter­nal N-channel MOSFET switch. Current and voltage slew
rates can be independently set to optimize harmonic
content of the switching waveforms vs efficiency. The
LT1738 can reduce high frequency harmonic power by as
much as 40dB with only minor losses in efficiency.

The LT1738 utilizes a current mode architecture opti­mized for single switch topologies such as boost, flyback
and Cuk. The IC includes gate drive and all necessary
oscillator, control and protection circuitry. Unique error
amp circuitry can regulate both positive and negative
voltages. The internal oscillator may be synchronized to
an external clock for more accurate placement of switch­ing harmonics.

Protection features include gate drive lockout for low VIN,
soft-start, output current limit, short-circuit current limit­ing, gate drive overvoltage clamp and input supply
undervoltage lockout.

, LTC and LT are registered trademarks of Linear Technology Corporation.

TYPICAL APPLICATIO

Ultralow Noise 5V to 12V Converter

GCL

GATE

PGND

NFB

CAP

22µH

3

5pF

1

4

CS

20

9

FB

101113

5V

V

IN

100µF

1.3nF

25k

25k

0.22µF

+

14 2

16.9k

3.3k

16

3.3k

15

22nF

12

1.5k

10nF

P3

17

V

IN

SHDN

5

V5

6

SYNC

7

C

T

LT1738

8

R

T

R

VSL

R

CSL

V

C

GND

SS

MBRD620

Si9426

25mΩ

U

4 × 150µF

OSCON

21.5k

2.5k

12V Output Noise

10µH

+

150µF

OSCON

OPTIONAL

AB

12V
1A

+

POINT A

ON SCHEMATIC

500µV/DIV

POINT B

ON SCHEMATIC

50mV/DIV

1738 TA01

(Bandwidth = 100MHz)

5µs/DIV

1738 TA01a

400µV

1738fa

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1

LT1738

1
2
3
4
5
6
7
8
9

10

TOP VIEW

G PACKAGE

20-LEAD PLASTIC SSOP

20
19
18
17
16
15
14
13
12
11

GATE

CAP
GCL

CS
V5

SYNC

C

T

R

T

FB

NFB

PGND
NC
NC
V

IN

R

VSL

R

CSL

SHDN
SS
V

C

GND

WW

W

ABSOLUTE AXI U RATI GS

U

PACKAGE/ORDER I FOR ATIO

(Note 1)

Supply Voltage (VIN)................................................ 20V

Gate Drive Current .....................................Internal Limit

V5 Current .................................................Internal Limit

SHDN Pin Voltage.................................................... 20V

Feedback Pin Voltage (Trans. 10ms) ...................... ±10V

Feedback Pin Current............................................ 10mA

Negative Feedback Pin Voltage (Trans. 10ms)........ ±10V

CS Pin.......................................................................... 5V

GCL Pin ..................................................................... 16V

SS Pin.......................................................................... 3V

Operating Junction Temperature Range

(Note 3) ............................................ –40°C to 125°C

Storage Temperature Range ................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec).................. 300°C

ELECTRICAL CHARACTERISTICS

temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VC = 0.9V, VFB = V
other pins open unless otherwise noted.

The ● denotes the specifications which apply over the full operating

Consult LTC Marketing for parts specified with wider operating temperature ranges.

T

= 150°C, θJA = 110°C/W

JMAX

REF

, R

VSL

UUW

ORDER PART

NUMBER

LT1738EG
LT1738IG

, R

= 16.9k, RT = 16.9k and

CSL

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
Error Amplifiers

V

REF

I

FB

FB

REG

V

NFR

I

NFR

NFB
g

m

I

ESK

I

ESRC

V

CLH

V

CLL

A

V

FB

OV

I

SS

Reference Voltage Measured at Feedback Pin ● 1.235 1.250 1.265 V
Feedback Input Current VFB = V
Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 20V ● 0.012 0.03 %/V
Negative Feedback Reference Voltage Measured at Negative Feedback Pin ● –2.56 –2.50 –2.45 V

Negative Feedback Input Current V
Negative Feedback Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 20V ● 0.009 0.03 %/V

REG

Error Amplifier Transconductance ∆IC = ±50µA 1100 1500 2200 µmho

Error Amp Sink Current VFB = V
Error Amp Source Current VFB = V
Error Amp Clamp Voltage High Clamp, VFB = 1V 1.27 V
Error Amp Clamp Voltage Low Clamp, VFB = 1.5V 0.12 V
Error Amplifier Voltage Gain 180 250 V/V
FB Overvoltage Shutdown Outputs Drivers Disabled 1.47 V
Soft-Start Charge Current VSS = 1V 9.0 12 µA

REF

with Feedback Pin Open

= V

NFB

NFR

+ 150mV, VC = 0.9V ● 120 200 350 µA

REF

– 150mV, VC = 0.9V ● 120 200 350 µA

REF

● 250 1000 nA

–37 –25 µA

● 700 2500 µmho

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2

LT1738

ELECTRICAL CHARACTERISTICS

temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VC = 0.9V, VFB = V

The ● denotes the specifications which apply over the full operating

, R

, R

REF

VSL

= 16.9k, RT = 16.9k and

CSL

other pins open unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
Oscillator and Sync

f

MAX

f

SYNC

V

SYNC

R

SYNC

Gate Drive

DC

MAX

VG

ON

VG

OFF

IG

SO

IG

SK

V

INUVLO

Current Sense

t

IBL

V

SENSE

V

SENSEF

Slew Control for the Following Slew Tests See Test Circuit in Figure 1b

V

SLEWR

V

SLEWF

VI

SLEWR

VI

SLEWF

Supply and Protection

V

INMIN

I

VIN

V

SHDN

∆V

SHDN

I

SHDN

V5 5V Reference Voltage 6.5V ≤ VIN ≤ 20V, IV5 = 5mA 4.85 5 5.20 V

IV5

SC

Note 1: Absolute Maximum Ratings are those values beyond which the
life of a device may be impaired.

Note 2: Supply current specification includes loads on each gate as in
Figure 1a. Actual supply currents vary with operating frequency, operating
voltages, V5 load, slew rates and type of external FET.

Note 3: The LT1738E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications from –40°C to 125°C are assured by

Max Switch Frequency 250 kHz
Synchronization Frequency Range Oscillator Frequency = 250kHz 290 kHz
SYNC Pin Input Threshold ● 0.7 1.4 2
SYNC Pin Input Resistance 40 kΩ

Maximum Switch Duty Cycle R

VSL

= R

= 3.9k, ● 90 93.5 %

CSL

Osc Frequency = 25kHz

Gate On Voltage VIN = 12, GCL = 12 10 10.4 10.7 V

V

= 12, GCL = 8 7.6 7.9 8.1 V

IN

Gate Off Voltage VIN = 12V 0.2 0.35 V
Max Gate Source Current VIN = 12V 0.3 A
Max Gate Sink Current VIN = 12V 0.3 A
Gate Drive Undervoltage Lockout (Note 5) V

= 6.5V 7.3 7.5 V

GCL

Switch Current Limit Blanking Time 100 ns
Sense Voltage Shutdown Voltage VC Pulled Low ● 86 103 120 mV
Sense Voltage Fault Threshold 220 300 mV

Output Voltage Slew Rising Edge R
Output Voltage Slew Falling Edge R
Output Current Slew Rising Edge (CS pin V) R
Output Current Slew Falling Edge (CS pin V) R

Minimum Input Voltage (Note 4) V
Supply Current (Note 3) R

R

VSL

VSL

VSL

VSL

GCL

VSL
VSL

= R

= 17k 26 V/µs

CSL

= R

= 17k 19 V/µs

CSL

= R

= 17k 2.1 V/µs

CSL

= R

= 17k 2.1 V/µs

CSL

= V

IN

= R

= 17k V

CSL

= R

= 17k V

CSL

= 12 ● 12 40 mA

IN

= 20 35 55 mA

IN

● 2.55 3.6 V

Shutdown Turn-On Threshold ● 1.31 1.39 1.48 V
Shutdown Turn-On Voltage Hysteresis ● 50 110 180 mV
Shutdown Input Current Hysteresis ● 10 24 35 µA

6.5V ≤ V

≤ 20V, IV5 = –5mA 4.80 5 5.15 V

IN

5V Reference Short-Circuit Current VIN = 6.5V Source 10 mA

= 6.5V Sink –10 mA

V

IN

design, characterization and correlation with statistical process
controls.The LT1738I is guaranteed and tested to meet performance
specifications from – 40°C to 125°C.

Note 4: Output gate drive is enabled at this voltage. The GCL voltage will
also determine driver activity.

Note 5: Gate drive is ensured to be on when V

is greater than max value.

IN

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3

LT1738

UW

TYPICAL PERFOR A CE CHARACTERISTICS

Feedback Voltage and Input
Current vs Temperature

1.260

1.258

1.256

1.254

1.252

1.250

1.248

1.246

FEEDBACK VOLTAGE (V)

1.244

1.242

1.240
–50 –25 0 25 50 75 100 125 150

TEMPERATURE (°C)

Feedback Overvoltage Shutdown
vs Temperature

1.70

1.65

1.60

1.55

1.50

1.45

1.40

1.35

FEEDBACK VOLTAGE (V)

1.30

1.25

1.20
–50 –25 0 25 50 75 100 125 150

TEMPERATURE (°C)

1738 G03

750
700
650

FB INPUT CURRENT (nA)

600
550
500
450
400
350
300
250

1738 G01

Error Amp Transconductance vs
Temperature

2000
1900
1800
1700
1600
1500
1400
1300
1200

TRANSCONDUCTANCE (µmho)

1100
1000

–50 –25 0 25 50 75 100 125 150

TEMPERATURE (°C)

Negative Feedback Voltage and
Input Current vs Temperature

2.480

2.485

2.490

2.495

2.500

2.505

2.510

NEGATIVE FEEDBACK VOLTAGE (V)

2.515

2.520
–50 –25 0 25 50 75 100 125 150

TEMPERATURE (°C)

Error Amp Output Current vs
Feedback Pin Voltage from Nominal

500
400
300
200
100

0

–100

CURRENT (µA)

–200
–300
–400
–500

–400 –300 –200 –100 0 100 200 300 400

FEEDBACK PIN VOLTAGE FROM NOMINAL (mV)

1738 G04

1738 G02

3.2

3.0

2.8

2.6

2.4

2.2

2.0

1.8

1.6

–40°C

125°C

NFB INPUT CURRENT (µA)

25°C

1738 G05

VC Pin Threshold and Clamp
Voltage vs Temperature

1.4

1.2

1.0

0.8

0.6

PIN VOLTAGE (V)

C

V

0.4

0.2

0

–50 –25 0 25 50 75 100 125 150

CLAMP

THRESHOLD

TEMPERATURE (°C)

4

1738 G06

CS Pin Trip Voltage and CS Fault
Voltage vs Temperature

240

220

200

180

160

140

CS PIN VOLTAGE (mV)

120

100

80

–50 –25 0 25 50 75 100 125 150

FAULT

TRIP

TEMPERATURE (°C)

1738 G07

SHDN Pin On and Off Thresholds
vs Temperature

1.50

1.45

1.40

1.35

SHDN PIN VOLTAGE (V)

1.30

1.25
–50 –25 0 25 50 75 100 125 150

ON

OFF

TEMPERATURE (°C)

1738 G08

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UW

TYPICAL PERFOR A CE CHARACTERISTICS

LT1738

SHDN Pin Hysteresis Current vs
Temperature VIN Current vs Temperature

27

25

23

21

19

SHDN PIN CURRENT (µA)

17

15

–50 –25 0 25 50 75 100 125 150

TEMPERATURE (°C)

1738 G09

18

VIN = 20 R

VSL

= 120kHz

0

TEMPERATURE (°C)

VIN = 12 R

VSL

, R

CSL

17

16

15

14

CURRENT (mA)

VIN = 12 R

13

IN

V

12

11

USING LOAD OF FIGURE 1A
f

OSC

10

–50 –25 25 75 125

Gate Drive High Voltage vs

Slope Compensation

110

100

90

80

70

PERCENT OF MAX CS VOLTAGE

60

50

20 40 60 80 100

0

DUTY CYCLE (%)

VC PIN = 0.9V

= 25°C

T

A

1738 G12

Temperature

10.7

10.6

10.5

10.4

10.3

10.2

10.1

10.0

9.90

GATE DRIVE A/B PIN VOLTAGE (V)

9.80

9.70
–50 –25 0 25 50 75 100 125 150

GCL = 12V

GCL = 6V

TEMPERATURE (°C)

= 17k

50

VSL

, R

, R

CSL

= 17k

CSL

VIN = 12V
NO LOAD

= 4.85k

100

1738 G10

1738 G13

150

CS Pin to VC Pin Transfer
Function

1.6
TA = 25°C

1.4

1.2

1.0

0.8

0.6

PIN VOLTAGE (V)

C

V

0.4

0.2

0

20 40 60 80 100 120

0

Gate Drive Low Voltage vs
Temperature

6.5

0.50

6.4

6.3

6.2

6.1

6.0

5.9

5.8

5.7

5.6

5.5

VIN = 12V

0.45
NO LOAD

0.40

0.35

0.30

0.25

0.20

0.15

0.10

GATE DRIVE A/B PIN VOLTAGE (V)

0.05

0

–50 –25 0 25 50 75 100 125 150

CS PIN VOLTAGE (mV)

1738 G11

TEMPERATURE (°C)

1738 G14

Gate Drive Undervoltage Lockout
Voltage vs Temperature

7.3
GCL = 6V

7.2

7.1

7.0

6.9

6.8

6.7

PIN VOLTAGE (V)

IN

6.6

V

6.5

6.4

6.3

–50 –25 0 25 50 75 100 125 150

TEMPERATURE (°C)

1738 G15

Soft-Start Current vs Temperature

9.5
SS VOLTAGE = 0.9V

9.3

9.1

8.9

8.7

8.5

8.3

8.1

SS PIN CURRENT (µA)

7.9

7.7

7.5

–50 –25 0 25 50 75 100 125 150

TEMPERATURE (°C)

1738 G16

V5 Voltage vs Load Current

5.08

5.06

5.04

5.02

5.00

V5 PIN VOLTAGE (V)

4.98

4.96
–10 –5 0 51015

–15

T = 125°C

T = 25°C

T = –40°C

LOAD CURRENT (mA)

1738 G17

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5

LT1738

PI FU CTIO S

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Part Supply

V5 (Pin 5): This pin provides a 5V output that can sink or

source 10mA for use by external components. V5 source
current comes from V
must be greater than 6.5V in order for this voltage to be in
regulation. If this pin is used, a small capacitor (<1µF) may
be placed on this pin to reduce noise. This pin can be left
open if not used.

GND (Pin 11): Signal Ground. The internal error amplifier,
negative feedback amplifier, oscillator, slew control cir­cuitry, V5 regulator, current sense and the bandgap refer­ence are referred to this ground. Keep the connection to
this pin, the feedback divider and VC compensation net­work free of large ground currents.

SHDN (Pin 14): The shutdown pin can disable the switcher.
Grounding this pin will disable all internal circuitry.

Increasing SHDN voltage will initially turn on the internal
bandgap regulator. This provides a precision threshold for
the turn on of the rest of the IC. As SHDN increases past

1.39V the internal LDO regulator turns on, enabling the
control and logic circuitry.

24µA of current is sourced out of the pin above the turn on
threshold. This can be used to provide hysteresis for the
shutdown function. The hysteresis voltage will be set by
the Thevenin resistance of the resistor divider driving this
pin times the current sourced out. Above approximately

2.1V the hysteresis current is removed. There is approxi­mately 0.1V of voltage hysteresis on this pin as well.

The pin can be tied high (to V

V

(Pin 17): Input Supply. All supply current for the part

IN

comes from this pin including gate drive and V5 regulator.
Charge current for gate drive can produce current pulses
of hundreds of milliamperes. Bypass this pin with a low
ESR capacitor.

. Sink current goes to GND. V

IN

for instance).

IN

IN

PGND (Pin 20): Power Driver Ground. This ground comes
from the MOSFET gate driver. This pin can have several
hundred milliamperes of current on it when the external
MOSFET is being turned off.

Oscillator

SYNC (Pin 6): The SYNC pin can be used to synchronize

the part to an external clock. The oscillator frequency
should be set close to the external clock frequency.
Synchronizing the clock to an external reference is useful
for creating more stable positioning of the switcher volt­age and current harmonics. This pin can be left open or
tied to ground if not used.

CT (Pin 7): The oscillator capacitor pin is used in conjunc­tion with RT to set the oscillator frequency. For RT = 16.9k:

C

(nf) = 129/f

OSC

RT (Pin 8): A resistor to ground sets the charge and
discharge currents of the oscillator capacitor. The nomi­nal value is 16.9k. It is possible to adjust this resistance
±25% to set oscillator frequency more accurately.

OSC

(kHz)

Gate Drive

GATE (Pin 1): This pin connects to the gate of an external

N-channel MOSFET. This driver is capable of sinking and
sourcing at least 300mA.

The GCL pin sets the upper voltage of the gate drive. The
GATE pin will not be activated until V
voltage as defined by the GCL pin (gate undervoltage
lockout).

The gate drive output has current limit protection to safe
guard against accidental shorts.

GCL (Pin 3): This pin sets the maximum gate voltage to the
GATE pin to the MOSFET gate drive. This pin should be
either tied to a zener, a voltage source, or VIN.

reaches a minimum

IN

When V
undervoltage lockout where the gate driver is driven low.
This, along with gate drive undervoltage lockout, prevents
unpredictable behavior during power up.

is below 2.55V the part will go into supply

IN

6

If the pin is tied to a zener or a voltage source, the
maximum gate drive voltage will be approximately V
– 0.2V. If it is tied to VIN, the maximum gate voltage is
approximately VIN – 1.6.

GCL

1738fa

PI FU CTIO S

LT1738

UUU

Approximately 50µA of current can be sourced from this
pin if V

This pin also controls undervoltage lockout of the gate
drive. If the pin is tied to a zener or voltage source, the gate
drive will not be enabled until VIN > V
is tied to VIN, then undervoltage lockout is disabled.

There is an internal 19V zener tied from this pin to ground
to provide a fail-safe for maximum gate voltage.

GCL

< V

– 0.8V.

IN

+ 0.8V. If this pin

GCL

Slew Control

CAP (Pin 2): This pin is the feedback node for the external

voltage slewing capacitor. Normally a small 1pf to 5pf
capacitor is connected from this pin to the drain of the
MOSFET.

The voltage slew rate is inversely proportional to this
capacitance and proportional to the current that the part
will sink and source on this pin. That current is inversely
proportional to R

R

(Pin 15): A resistor to ground sets the current slew

CSL

rate for the external drive MOSFET during switching. The
minimum resistor value is 3.3k and the maximum value is
68k. The time to slew between on and off states of the
MOSFET current will determine how the di/dt related
harmonics are reduced. This time is proportional to R
and RS (the current sense resistor) and maximum current.
Longer times produce a greater reduction of higher fre­quency harmonics.

R

(Pin 16): A resistor to ground sets the voltage slew

VSL

rate for the drain of the external drive MOSFET. The
minimum resistor value is 3.3k and the maximum value is
68k. The time to slew between on and off states on the
MOSFET drain voltage will determine how dv/dt related
harmonics are reduced. This time is proportional to R
CV and the input voltage. Longer times produce more
rolloff of harmonics. CV is the equivalent capacitance from
CAP to the drain of the MOSFET.

VSL

.

CSL

,

VSL

Switch Mode Control

SS (Pin 3): The SS pin allows for ramping of the switch

current threshold at startup. Normally a capacitor is placed
on this pin to ground. An internal 9µA current source will
charge this capacitor up. The voltage on the VC pin cannot

exceed the voltage on SS. Thus peak current will ramp up
as the SS pin ramps up. During a short circuit fault the SS
pin will be discharged to ground thus reinitializing soft­start.

When SS is below the VC clamp voltage the VC pin will
closely track the SS pin.

This pin can be left open if not used.
CS (Pin 4): This is the input to the current sense amplifier.

It is used for both current mode control and current
slewing of the external MOSFET. Current sense is accom­plished via a sense resistor (RS) connected from the
source of the external MOSFET to ground. CS is connected
to the top of RS. Current sense is referenced to the GND
pin.

The switch maximum operating current will be equal to

0.1V/RS. At CS = 0.1V, the gate driver will be immediately
turned off (no slew control).

If CS = 0.22V in addition to the drivers being turned off, V
and SS will be discharged to ground (short-circuit protec­tion). This will hasten turn off on subsequent cycles.

FB (Pin 9): The feedback pin is used for positive voltage
sensing. It is the inverting input to the error amplifier. The
noninverting input of this amplifier connects internally to
a 1.25V reference.

If the voltage on this pin exceeds the reference by 220mV,
then the output driver will immediately turn off the external
MOSFET (no slew control). This provides for output over­voltage protection

When this input is below 0.9V then the current sense
blanking will be disabled. This will assist start up.

NFB (Pin 10): The negative feedback pin is used for
sensing a negative output voltage. The pin is connected to
the inverting input of the negative feedback amplifier
through a 100k source resistor. The negative feedback
amplifier provides a gain of –0.5 to the FB pin. The nominal
regulation point would be –2.5V on NFB. This pin should
be left open if not used.

If NFB is being used then overvoltage protection will occur
at 0.44V below the NFB regulation point.

At NFB < –1.8 current sense blanking will be disabled.

C

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LT1738

PI FU CTIO S

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TEST CIRCUITS

VC (Pin 12): The compensation pin is used for frequency
compensation and current limiting. It is the output of the
error amplifier and the input of the current comparator.
Loop frequency compensation can be performed with an
RC network connected from the VC pin to ground. The
voltage on VC is proportional to the switch peak current.
The normal range of voltage on this pin is 0.25V to 1.27V.
However, during slope compensation the upper clamp
voltage is allowed to increase with the compensation.

During a short-circuit fault the VC pin will be discharged to
ground.

W

BLOCK DIAGRA

V

NFB

100k 50k

10

+

NEGATIVE
FEEDBACK

AMP

–

SHDN R

IN

14 17 5 15

REGULATOR

V

REG

20mA

5pF

CAP

GATE

2

IN5819

ZVN3306A

10

+
–

1738 F01a

Figure 1a. Typical Test Circuitry

V5

TO
DRIVERS

R

VSL

CSL

16

0.9A

5pF

CAP

GATE

CS

0.1

IN5819

Si4450DY

10

+
–

1738 F01b

Figure 1b. Test Circuit for Slew

GCL

3

CAP

2

9

FB

–

ERROR

+

AMP

SLEW

CONTROL

GATE

1

+

1.25V

V

12

C

PGND

20

–

4

SS

13

COMP

+

SENSE

AMP

+

CS

–

SQ

FF

R

8

T

OSCILLATOR

C

7

T

SYNC GND

R

SUB

116

1738 BD

1738fa

8

OPERATIO

LT1738

U

In noise sensitive applications switching regulators tend
to be ruled out as a power supply option due to their
propensity for generating unwanted noise. When switch­ing supplies are required due to efficiency or input/output
constraints, great pains must be taken to work around the
noise generated by a typical supply. These steps may
include pre and post regulator filtering, precise synchro­nization of the power supply oscillator to an external clock,
synchronizing the rest of the circuit to the power supply
oscillator or halting power supply switching during noise
sensitive operations. The LT1738 greatly simplifies the
task of eliminating supply noise by enabling the design of
an inherently low noise switching regulator power supply.

The LT1738 is a fixed frequency, current mode switching
regulator with unique circuitry to control the voltage and
current slew rates of the output switch. Current mode
control provides excellent AC and DC line regulation and
simplifies loop compensation.

Slew control capability provides much greater control
over the power supply components that can create con­ducted and radiated electromagnetic interference. Com­pliance with EMI standards will be an easier task and will
require fewer external filtering components.

The LT1738 uses an external N-channel MOSFET as the
power switch. This allows the user to tailor the drive
conditions to a wide range of voltages and currents.

CURRENT MODE CONTROL

Referring to the block diagram. A switching cycle begins
with an oscillator discharge pulse, which resets the RS
flip-flop, turning on the GATE driver and the external
MOSFET. The switch current is sensed across the external
sense resistor and the resulting voltage is amplified and
compared to the output of the error amplifier (VC pin). The
driver is turned off once the output of the current sense
amplifier exceeds the voltage on the VC pin. In this way
pulse by pulse current limit is achieved.

Internal slope compensation is provided to ensure stabil­ity under high duty cycle conditions.

Output regulation is obtained using the error amp to set
the switch current trip point. The error amp is a

transconductance amplifier that integrates the difference
between the feedback output voltage and an internal

1.25V reference. The output of the error amp adjusts the
switch current trip point to provide the required load
current at the desired regulated output voltage. This
method of controlling current rather than voltage pro­vides faster input transient response, cycle-by-cycle
current limiting for better output switch protection and
greater ease in compensating the feedback loop. The V
pin is used for loop compensation and current limit
adjustment. During normal operation the VC voltage will
be between 0.25V and 1.27V. An external clamp on VC or
SS may be used for lowering the current limit.

The negative voltage feedback amplifier allows for direct
regulation of negative output voltages. The voltage on the
NFB pin gets amplified by a gain of – 0.5 and driven on to
the FB input, i.e., the NFB pin regulates to –2.5V while the
amplifier output internally drives the FB pin to 1.25V as in
normal operation. The negative feedback amplifier input
impedance is 100k (typ) referred to ground.

Soft-Start

Control of the switch current during start up can be
obtained by using the SS pin. An external capacitor from
SS to ground is charged by an internal 9µA current source.
The voltage on VC cannot exceed the voltage on SS. Thus
as the SS pin ramps up the VC voltage will be allowed to
ramp up. This will then provide for a smooth increase in
switch maximum current. SS will be discharged as a result
of the CS voltage exceeding the short circuit threshold of
approximately 0.22V.

Slew Control

Control of output voltage and current slew rates is achieved
via two feedback loops. One loop controls the MOSFET
drain dV/dt and the other loop controls the MOSFET dI/dt.

The voltage slew rate uses an external capacitor between
CAP and the MOSFET drain. This integrating cap closes the
voltage feedback loop. The external resistor R
current for the integrator. The voltage slew rate is thus
inversely proportional to both the value of capacitor and
R

.

VSL

sets the

VSL

C

1738fa

9

LT1738

OPERATIO

U

The current slew feedback loop consists of the voltage
across the external sense resistor, which is internally
amplified and differentiated. The derivative is limited to a
value set by R
proportional to both the value of sense resistor and R

The two control loops are combined internally so that a
smooth transition from current slew control to voltage
slew control is obtained. When turning on, the driver
current will slew before voltage. When turning off, voltage
will slew before current. In general it is desirable to have
R

and R

VSL

Internal Regulator

Most of the control circuitry operates from an internal 2.4V
low dropout regulator that is powered from VIN. The
internal low dropout design allows VIN to vary from 2.7V
to 20V with stable operation of the controller. When SHDN
< 1.3V the internal regulator is completely disabled.

5V Regulator

A 5V regulator is provided for powering external circuitry.
This regulator draws current from VIN and requires VIN to
be greater than 6.5V to be in regulation. It can sink or
source 10mA. The output is current limited to prevent
against destruction from accidental short circuits.

Safety and Protection Features

There are several safety and protection features on the
chip. The first is overcurrent limit. Normally the gate driver
will go low when the output of the internal sense amplifier
exceeds the voltage on the VC pin. The VC pin is clamped
such that maximum output current is attained when the CS
pin voltage is 0.1V. At that level the outputs will be
immediately turned off (no slew). The effect of this control
is that the output voltage will foldback with overcurrent.

In addition, if the CS voltage exceeds 0.22V, the VC and SS
pins will be discharged to ground, resetting the soft-start
function. Thus if a short is present this will allow for faster
MOSFET turnoff and less MOSFET stress.

If the voltage on the FB pin exceeds regulation by approxi­mately 0.22V, the outputs will immediately go low. The
implication is that there is an overvoltage fault.

. The current slew rate is thus inversely

CSL

of similar value.

CSL

CSL.

The voltage on GCL determines two features. The first is
the maximum gate drive voltage. This will protect the
MOSFET gate from overvoltage.

With GCL tied to a zener or an external voltage source then
the maximum gate driver voltage is approximately
V

␣ – 0.2V. If GCL is tied to VIN, then the maximum gate

GCL

voltage is determined by VIN and is approximately
VIN – 1.6V. There is an internal 19V zener on the GCL pin
that prevents the gate driver pin from exceeding approxi­mately 19V.

In addition, the GCL voltage determines undervoltage
lockout of the gate drive. This feature disables the gate
driver if VIN is too low to provide adequate voltage to turn
on the MOSFET. This is helpful during start up to insure the
MOSFET has sufficient gate drive to saturate.

If GCL is tied to a voltage source or zener less than 6.8V,
the gate driver will not turn on until VIN exceeds GCL
voltage by 0.8V. For V
insured to be off for VIN < 7.3V and it will be turned on by
V

+ 0.8V.

GCL

If GCL is tied to VIN, the gate driver is always on
(undervoltage lockout is disabled).

The gate drive has current limits for the drive currents. If
the sink or source current is greater than 300mA then the
current will be limited.

The V5 regulator also has internal current limiting that will
only guarantee ±10mA output current.

There is also an on chip thermal shutdown circuit that will
turn off the output in the event the chip temperature rises
to dangerous levels. Thermal shutdown has hysteresis
that will cause a low frequency (<1kHz) oscillation to occur
as the chip heats up and cools down.

The chip has an undervoltage lockout feature that will
force the gate driver low in the event that VIN drops below

2.5V. This insures predictable behavior during start up and
shut down. SHDN can be used in conjuction with an
external resistor divider to completely disable the part if
the input voltage is too low. This can be used to insure
adequate voltage to reliably run the converter. See the
section in Applications Information.

Table 1 summarizes these features.

above 6.5V, the gate drive is

GCL

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OPERATIO

Table 1. Safety and Protection Features

FEATURE FUNCTION EFFECT on GATE DRIVER SLEW CONTROL EFFECT on VC, SS

Maximum Current Fault Turn Off FET at Maximum Immediately Goes Low Overridden None

Switch Current (V

Short-Circuit Fault Turn Off FET and Reset V

for Short-Circuit (V

Overvoltage Fault Turn Off Driver If FB > V

(Output Overvoltage)

GCL Clamp Set Max Gate Voltage to Prevent Limits Max Voltage None None

FET Gate Breakdown

Gate Drive Disable Gate Drive When V
Undervoltage Lockout Is Too Low. Set Via GCL Pin

Thermal Shutdown Turn Off Driver If Chip Immediately Goes Low Overridden None

Temperature Is Too Hot
VIN Undervoltage Lockout Disable Part When VIN ≅ 2.55V Immediately Goes Low Overridden None
Gate Drive Source and Sink Current Limit Limit Gate Drive Current Limit Drive Current None None
V5 Source/Sink Current Limit Limit Current from V5 None None None
Shutdown Disable Part When SHDN <1.3V

= 0.1)

SENSE

C

= 0.22) to GND

SENSE

+ 0.22V Immediately Goes Low Overridden None

REG

IN

Immediately Goes Low Overridden Discharge VC, SS

Immediately Goes Low Overridden None

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APPLICATIO S I FOR ATIO

Reducing EMI from switching power supplies has tradi­tionally invoked fear in designers. Many switchers are
designed solely on efficiency and as such produce wave­forms filled with high frequency harmonics that then
propagate through the rest of the system.

The LT1738 provides control over two of the more impor­tant variables for controlling EMI with switching inductive
loads: switch voltage slew rate and switch current slew
rate. The use of this part will reduce noise and EMI over
conventional switch mode controllers. Because these
variables are under control, a supply built with this part will
exhibit far less tendency to create EMI and less chance of
encountering problems during production.

It is beyond the scope of this data sheet to get into EMI
fundamentals. Application Note 70 contains much infor­mation concerning noise in switching regulators and
should be consulted.

Oscillator Frequency

The oscillator determines the switching frequency and
therefore the fundamental positioning of all harmonics.
The use of good quality external components is important

to ensure oscillator frequency stability. The oscillator is of
a sawtooth design. A current defined by external resistor
RT is used to charge and discharge the capacitor CT . The
discharge rate is approximately ten times the charge rate.

By allowing the user to have control over both compo­nents, trimming of oscillator frequency can be more easily
achieved.

The external capacitance CT is chosen by:

CnF

()

=

T

where f is the desired oscillator frequency in kHz. For R

2180

f kHz R k

()•()

Ω

T

T

equal to 16.9k, this simplifies to:

CnF

()

T

129

=

f kHz

()

e.g., CT = 1.29nF for f = 100kHz

Nominally RT should be 16.9k. Since it sets up current, its
temperature coefficient should be selected to compliment
the capacitor. Ideally, both should have low temperature
coefficients.

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APPLICATIO S I FOR ATIO

Oscillator frequency is important for noise reduction in
two ways. First the lower the oscillator frequency the lower
the waveform’s harmonics, making it easier to filter them.
Second the oscillator will control the placement of the
output voltage harmonics which can aid in specific prob­lems where you might be trying to avoid a certain fre­quency bandwidth.

Oscillator Sync

If a more precise frequency is desired (e.g., to accurately
place harmonics) the oscillator can be synchronized to an
external clock. Set the RC timing components for an
oscillator frequency 10% lower than the desired sync
frequency.

Drive the SYNC pin with a square wave (with greater than
2V amplitude). The rising edge of the sync square wave
will initiate clock discharge. The sync pulse should have a
minimum pulse width of 0.5µs.

Be careful in sync’ing to frequencies much different from
the part since the internal oscillator charge slope deter­mines slope compensation. It would be possible to get into
subharmonic oscillation if the sync doesn’t allow for the
charge cycle of the capacitor to initiate slope compensa­tion. In general, this will not be a problem until the sync
frequency is greater than 1.5 times the oscillator free-run
frequency.

Slew Rate Setting

The primary reason to use this part is to gain advantage of
lower EMI and noise due to the slew control. The rolloff in
higher frequency harmonics has its theoretical basis with
two primary components. First, the clock frequency sets
the fundamental positioning of harmonics and second, the
associated normal frequency rolloff of harmonics.

This part creates a second higher frequency rolloff of
harmonics that inversely depends on the slew time, the
time that voltage or current spends between the off state
and on state. This time is adjustable through the choice of
the slew resistors, the external resistors to ground on the
R

and R

VSL

the external voltage feedback capacitor CV (from CAP to
the MOSFET drain) and the sense resistor. Lower slew

pins and the external components used for

CSL

rates (longer slew times, lower rolloff frequency for har­monics ) are created with higher values of R
and the current sense resistor.

Setting the voltage and current slew rates should be done
empirically. The most practical way of determining these
components is to set CV and the sense resistor value.
Then, start by making R
in series with 3.3k. Starting from the lowest resistor
setting (fast slew) adjust the pots until the noise level
meets your guidelines. Note that slower slewing wave­forms will dissipate more power so that efficiency will
drop. You can monitor this as you make your slew adjust­ment by measuring input and output voltage and their
respective currents. Monitor the MOSFET temperature as
slew rates are slowed. The MOSFET will heat up as
efficiency decreases.

Measuring noise should be done carefully. It is easy to
introduce noise by poor measurement techniques. Con­sult AN70 for recommended measurement techniques.
Keeping probe ground leads very short is essential.

Usually it will be desirable to keep the voltage and current
slew resistors approximately the same. There are circum­stances where a better optimization can be found by
adjusting each separately, but as these values are sepa­rated further, a loss of independence of control may occur.

It is possible to use a single slew setting resistor. In this
case the R
with a value of 1.8k to 34k (one half the individual resis­tors) can then be tied from these pins to ground.

In general only the R
ment of current slew. The current slew time also depends
on the current sense resistor but this resistor is normally
set with consideration of the maximum current in the
MOSFET.

Setting the voltage slew also involves selection of the
capacitor CV. The voltage slew time is proportional to the
output voltage swing (basically input voltage), the external
voltage feedback capacitor and the R
higher input voltages smaller capacitors will be used with
lower R

and R

VSL

values. For a starting point use Table␣ 2.

VSL

CSL

, R

VSL

pins are tied together. A resistor

value will be available for adjust-

CSL

each a 50k resistor pot

CSL

VSL

, R

CSL

, C

V

VSL

value. Thus at

12

1738fa

LT1738

FB PIN

1738 F03

V

OUT

R2

R1

NFB PIN

I

NFB

1738 F04

–V

OUT

R2

R1

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APPLICATIO S I FOR ATIO

Table 2

INPUT VOLTAGE CAPACITOR VALUE

< 25V 5pF
50V 2.5pF
100V 1pF

Smaller value capacitors can be made in two ways. The
first is simply combining two capacitors in series. The
equivalent capacitance is then (C1 • C2)/(C1 + C2).

The second method makes use of a capacitor divider. Care
should be taken that the voltage rating of the capacitor
satisfies the full voltage swing thus essentially the same
rating as the MOSFET.

The equivalent slew capacitance for Figure 2 is
(C1 • C2)/(C1 + C2 + C3).

MOSFET DRAIN

CAP

C2

C1

C3

1738 F02

Figure 3

Negative Output Voltage Setting

Negative output voltage can be sensed using the NFB pin.
In this case regulation will occur when the NFB pin is at
–2.5V. The nominal input bias current for the NFB is –25µA
(I

), which needs to be accounted for in setting up the

NFB

divider.
Referring to Figure 4, R1 is chosen such that:

RR



12

=





V

−

OUT

RA

25 2 25

+µ

.•

25

.






A suggested value for R2 is 2.5k. The NFB pin is normally
left open if the FB pin is being used.

Figure 2

Positive Output Voltage Setting

Sensing of a positive output voltage is usually done using
a resistor divider from the output to the FB pin. The
positive input to the error amp is connected internally to a

1.25V bandgap reference. The FB pin will regulate to this
voltage.

Referring to Figure 3, R1 is determined by:

RR

12






125

OUT

.



1=−





V

The FB bias current represents a small error and can
usually be ignored for values of R1||R2 up to 10k.

One word of caution, sometimes a feedback zero is added
to the control loop by placing a capacitor across R1. If the
feedback capacitively pulls the FB pin above the internal
regulator voltage (2.4V), output regulation may be dis­rupted. A series resistance with the feedback pin can
eliminate this potential problem. There is an internal clamp
on FB that clamps at 0.7V above the regulation voltage that
should also help prevent this problem.

Figure 4

Dual Polarity Output Voltage Sensing

Certain applications may benefit from sensing both posi­tive and negative output voltages. When doing this each
output voltage resistor divider is individually set as previ­ously described. When both FB and NFB pins are used, the
LT1738 will act to prevent either output from going
beyond its set output voltage. The highest output (lightest
load) will dominate control of the regulator. This technique
would prevent either output from going unregulated high
at no load. However, this technique will also compromise
output load regulation.

Shutdown

If SHDN is pulled low, the regulator will turn off. As the
SHDN pin voltage is increased from ground the internal
bandgap regulator will be powered on. This will set a 1.39V
threshold for turn on of the internal regulator that runs

1738fa

13

LT1738

VC PIN

1738 F05

R

VC

2k

C

VC

0.01µF

C

VC2

4.7nF

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APPLICATIO S I FOR ATIO

most of the control circuitry of the regulator. Note after the
control circuitry powers on, gate driver activity will depend
on the voltage of VIN with respect to the voltage on GCL.

As the SHDN pin enables the internal regulator a 24µA
current will be sourced from the pin that can provide
hysteresis for undervoltage lockout. This hysteresis can
be used to prevent part shutdown due to input voltage sag
from an initial high current draw.

In addition to the current hysteresis, there is also approxi­mately 100mV of voltage hysteresis on the SHDN pin.

When the SHDN pin is greater than 2.2V, the hysteretic
current from the part will be reduced to essentially zero.

If a resistor divider is used to set the turn on threshold then
the resistors are determined by the following equations:

RA RB

+



V

=



ON SHDN



VRA

HYST

=

Reworking these equations yields:

VV VV

()

RA

RB

HYST SHDN ON SHDN

=

VV VV

()

HYST SHDN ON SHDN

=

So if we wanted to turn on at 20V with 2V of hysteresis:



V

•





RB



V

∆

SHDN

•



RA RB



••
IV

()

SHDN SHDN

••

•

IVV

[]

SHDN ON SHDN

+

−∆

•

−∆

−

()

I

SHDN






RA

RB

V

IN

SHDN

Frequency Compensation

Loop frequency compensation is accomplished by way of
a series RC network on the output of the error amplifier
(VC␣ pin).

Figure 5

Referring to Figure 5, the main pole is formed by capacitor
CVC and the output impedance of the error amplifier
(approximately 400kΩ). The series resistor RVC creates a
“zero” which improves loop stability and transient re­sponse. A second capacitor C

, typically one-tenth the

VC2

size of the main compensation capacitor, is sometimes
used to reduce the switching frequency ripple on the V

C

pin. VC pin ripple is caused by output voltage ripple
attenuated by the output divider and multiplied by the error
amplifier. Without the second capacitor, VC pin ripple is:

V

CPINRIPPLE

where V

125.• • •

=

= Output ripple (V

RIPPLE

RIPPLE VC

V

OUT

)

P-P

VgmR

gm = Error amplifier transconductance
RVC = Series resistor on VC pin
V

= DC output voltage

OUT

VVVV

2 1 39 20 0 1

RA

RB

•. •.

=

VVVV

2 1 39 20 0 1

•. •.

=

µ−

24 20 1 39

−

AV

µ

24 1 39

•.

−

AV V

•.

()

=

=

175

23 4

.

.

k

k

Resistor values could be altered further by adding zeners
in the divider string. A resistor in series with SHDN pin
could further change hysteresis without changing turn on

To prevent irregular switching, VC pin ripple should be
kept below 50mV

. Worst-case VC pin ripple occurs at

P-P

maximum output load current and will also be increased if
poor quality (high ESR) output capacitors are used. The
addition of a 0.0047µF capacitor for C

pin reduces

VC2

switching frequency ripple to only a few millivolts. A low
value for RVC will also reduce VC pin ripple, but loop phase
margin may be inadequate.

voltage.

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APPLICATIO S I FOR ATIO

Setting Current Limit

The sense resistor sets the value for maximum operating
current. When the CS pin voltage is 0.1V the gate driver
will immediately go low (no slew control). Therefore the
sense resistor value should be set to RS = 0.1V/I
where I
I

SW(PEAK)

values and tolerances. Certainly it should be set below the
saturation current value for the inductor.

If the CS pin voltage is 0.22V in addition to the driver going
low, VC and SS will be discharged to ground. This is to
provide additional protection in the event of a short circuit.
By discharging VC and SS the MOSFET will not be stressed
as hard on subsequent cycles since the current trip will be
set lower.

Turn off of the MOSFET will normally be inhibited for about
100ns at the start of every turn on cycle. This is to prevent
noise from interfering with normal operation of the control­ler. This current sense blanking does not prevent the out­puts from being turned off in the event of a fault. Slewing
of the gate voltage effectively provides additional blanking.

Traces to the SENSE resistor should be kept short and wide
to minimize resistance and inductance. Large interwinding
capacitance in the transformer or high capacitance on the
drain of the MOSFET will produce a current pulse through
the sense resistor during drain voltage slewing. The mag­nitude of the pulse is C • dV/dt where C is the capacitance
and dV/dt is the voltage slew rate which is controlled by the
part. This pulse will increase the sensed current on switch
turn on and if large enough can cause premature MOSFET
turn off. If this occurs, the inductor transformer may need
a different winding technique (see AN39) or alternatively,
a blanking circuit can be used. Please contact the LTC ap­plications group for support if required.

Soft-Start

The soft-start pin is used to provide control of switching
current during startup. The maximum voltage on the V
pin is approximately the voltage on the SS pin. A current
source will linearly charge a capacitor on the SS pin. The
VC pin voltage will thus ramp up also. The approximate
time for the voltage on these pins to ramp up is
(1.31V/9µA) • CSS or approximately 146ms per µF.

SW(PEAK)

will depend on the topology and component

is the peak current in the MOSFET.

SW(PEAK)

,

C

The soft-start current will be initiated as soon as the part
turns on. Soft-start will be reinititated after a short-circuit
fault.

Thermal Considerations

Most of the IC power dissipation is derived from the V
pin. The VIN current depends on a number of factors
including: oscillator frequency; loads on V5; slew settings;
gate charge current. Additional power is dissipated if V5
sinks current and during the MOSFET gate discharge.

The power dissipation in the IC will be the sum of:

1) The RMS VIN current times V

2) V5 RMS sink current times 5V

3) The gate drive’s RMS discharge current times voltage.
Because of the strong VIN component it is advantageous

to operate the LT1738 at as low a VIN as possible.
It is always recommended that package temperature be

measured in each application. The part has an internal
thermal shutdown to minimize the chance of IC destruc­tion but this should not replace careful thermal design.

The thermal shutdown feature does not protect the exter­nal MOSFET. A separate analysis must be done for this
device to insure that it is operating within safe limits.

Once IC power dissipation, P
tion temperature is then computed as:

TJ = T

where T
thermal resistance. For the 20-pin SSOP, θ

Choosing The Inductor

For a boost converter, inductor selection involves trade­offs of size, maximum output power, transient response
and filtering characteristics. Higher inductor values pro­vide more output power and lower input ripple. However,
they are physically larger and can impede transient re­sponse. Low inductor values have high magnetizing cur­rent, which can reduce maximum power and increase
input current ripple.

+ P

AMB

is ambient temperature and θ

AMB

DIS

• θ

JA

IN

, is determined die junc-

DIS

is the package

JA

is 100°C/W.

JA

IN

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The following procedure can be used to handle these
trade-offs:

1. Assume that the average inductor current for a boost
converter is equal to load current times V
decide whether the inductor must withstand continu­ous overload conditions. If average inductor current at
maximum load current is 0.5A, for instance, a 0.5A
inductor may not survive a continuous 1.5A overload
condition. Also be aware that boost converters are not
short-circuit protected, and under output short condi­tions, only the available current of the input supply
limits inductor current.

2. Calculate peak inductor current at full load current to
ensure that the inductor will not saturate. Peak current
can be significantly higher than output current, espe­cially with smaller inductors and lighter loads, so don’t
omit this step. Powdered iron cores are forgiving
because they saturate softly, whereas ferrite cores satu­rate abruptly. Other core materials fall in between. The
following formula assumes continuous mode opera­tion but it errs only slightly on the high side for discon­tinuous mode, so it can be used for all conditions.

OUT/VIN

and

Capacitors

Correct choice of input and output capacitors can be very
important to low noise switcher performance. Noise de­pends more on the ESR of the capacitors. In addition lower
ESR can also improve efficiency.

Input capacitors must also withstand surges that occur
during the switching of some types of loads. Some solid
tantalum capacitors can fail under these surge conditions.

Design Note 95 offers more information but the following
is a brief summary of capacitor types and attributes.

Aluminum Electrolytic:

will typically only be used for higher voltage applications.
Large values will be needed for low ESR.

Specialty Polymer Aluminum:

with their series CD capacitors. While they are only avail­able for voltages below 16V, they have very low ESR and
good surge capability.

Solid Tantalum:

the maximum voltage rating is 50V. With large surge
currents the capacitor may need to be derated or you need
a special type such as the AVX TPS line.

Small size and low impedance. Typically

Low cost and higher voltage. They

Panasonic has come out



V

II

=+

PEAK OUT

L = inductance value
VIN = supply voltage
V

= output voltage

OUT

I = output current
f = oscillator frequency

3. Choose a core geometry. For low EMI problems a
closed structure should be used such as a pot core, ER
core or toroid (see AN70 appendix I).

4. Select an inductor that can handle peak current,
average current (heating effects) and fault current.

5. Finally, double check output voltage ripple.

The experts in the Linear Technology Applications depart­ment have experience with a wide range of inductor types
and can assist you in making a good choice.

OUT



V



VV V

IN OUT IN

IN

()

LfV

•••2

–

OUT






OS-CON:

able for 35V or less. Form factor may be a problem.

Ceramic:

voltage bypass. They may resonate with their ESL before
ESR becomes dominant. Recent multilayer ceramic (MLC)
capacitors provide larger capacitance with low ESR.

There are continuous improvements being made in ca­pacitors so consult with manufacturers as to your specific
needs.

Input Capacitors

The input capacitor should have low ESR at high frequen­cies since this will be an important factor concerning how
much conducted noise is generated.

There are two separate requirements for input capacitors.
The first is for the supply to the part’s VIN pin. The VIN pin
will provide current for the part itself and the gate charge
current.

Lower impedance than aluminum but only avail-

Generally used for high frequency and high

1738fa

16

LT1738

U

WUU

APPLICATIO S I FOR ATIO

The worst component from an AC point is the gate charge
current. The actual peak current depends on gate capaci­tance and slew rate, being higher for larger values of each.
The total current can be estimated by gate charge and
frequency of operation. Because of the slewing with this
part gate charge is spread out over a longer time period
than with a normal FET driver. This reduces capacitance
requirements.

Typically the current will have spikes of under 100mA
located at the gate voltage transitions. This is charge/
discharge to and from the threshold voltage. Most slewing
occurs with the gate voltage near threshold.

Since the part’s VIN will typically be under 15V many
options are available for choice of capacitor. Values of
input capacitor for just the VIN requirement will typically be
in the 50µF range with an ESR of under 0.1Ω.

In addition to the part’s supply, decoupling of the supply
to the inductor needs to be considered. If this is the same
supply as the VIN pin then that capacitor will need to be
increased. However, often with this part the inductor
supply will be a higher voltage and as such will use a
separate capacitor.

The inductor’s decoupling capacitor will see the switch
current as ripple.

The above switch current computation can be used to
estimate the capacity for these capacitors.

ESR

DC

MIN

•
f

C

=

IN

where ∆V

∆

CAP

1

V

∆

CAP

I

SW MAX

−

()

is the allowed sag on the input capacitor.
ESR is the equivalent series resistance for the cap. In
general allowed sag will be a few tenths of a volt.

Output Filter Capacitor

The output capacitor is chosen both for capacity and ESR.
The capacity must supply the load current in the switch on
state. While slew control reduces higher frequency com­ponents of the ripple current in the capacitor, the capacitor
ESR and the magnitude of the output ripple current
controls the fundamental component. ESR should also be

low to reduce capacitor dissipation. Typically ESR should
be below 0.05Ω.

The capacitance value can be computed by consideration
of desired load ripple, duty cycle and ESR.

ESR

•

DC

MIN

f

C

OUT

=

V

∆

OUT

I

∆

()

LMAX

1

−

MOSFET Selection

There is a wide variety of MOSFETs to choose from for this
part. The part will work with either normal threshold (3V to
4V) or logic level threshold devices (1V to 2V).

Select a voltage rating to insure under worst-case condi­tions that the MOSFET will not break down. Next choose an
R

sufficiently low to meet both the power dissipation

ON

capabilities of the MOSFET package as well as overall
efficiency needs of the converter.

The LT1738 can handle a large range of gate charges.
However at very large charge stability may be affected.

The power dissipation in the MOSFET depends on several
factors. The primary element is I2R heating when the
device is on. In addition, power is dissipated when the
device is slewing. An estimate for power dissipation is:





PV

=



IN



∆

2

I

+

•
I

SR




2

fI R DC

+

•••

ON



2

I

4

22

VR I

−+



IN ON





+



2

•





V

SR

2







I

∆

3

•
4















I

•







where I is the average current, ∆I is the ripple current in the
switch, ISR is the current slew rate, VSR is the voltage slew
rate, f is the oscillator frequency, DC is the duty cycle and
RON is the MOSFET on-resistance.

Setting GCL Voltage

Setting the voltage on the GCL pin depends on what type
of MOSFET is used and the desired gate drive undervolt­age lockout voltage.

1738fa

17

LT1738

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WUU

APPLICATIO S I FOR ATIO

First determine the maximum gate drive that you require.
Typically you will want it to be at least 2V greater than the
rated threshold. Higher voltages will lower the on resis­tance and increase efficiency. Be certain to check the
maximum allowed gate voltage. Often this is 20V but for
some logic threshold MOSFETs it is only 8V to 10V.

V

needs to be set approximately 0.2V above the desired

GCL

max gate threshold. In addition VIN needs to be at least

1.6V above the gate voltage.
The GCL pin can be tied to VIN which will result in a

maximum gate voltage of V
This pin also controls undervoltage lockout of the gate

drive. The undervoltage lockout will prevent the MOSFET
from switching until there is sufficient drive present.

If GCL is tied to a voltage source or zener less than 6.8V,
the gate drivers will not turn on until VIN exceeds the GCL
voltage by 0.8V. For V

GCL

insured to be off for VIN < 7.3V and they will be turned on
by V

GCL

+ 0.8V.

If GCL is tied to VIN, the gate driver is always on
(undervoltage lockout is disabled).

Approximately 50µA of current can be sourced from this
pin if VIN > V

+ 0.8V. This could be used to bias a zener.

GCL

The GCL pin has an internal 19V zener to ground that will
provide a failsafe for maximum gate voltage.

As an example say we are using a Siliconix Si4480DY
which has R

rated at 6V. To get 6V, V

DS(ON)

be set to 6.2V and VIN needs to be at least 7.6V.

Gate Driver Considerations

In general, the MOSFET should be positioned as close to
the part as possible to minimize inductance.

When the part is active the gate drive will be pulled low to
less than 0.2V. When the part is off, the gate drive contains
a 40k resistor in series with a diode to ground that will offer
passive holdoff protection. If you are using some logic
level MOSFETs this might not be sufficient. A resistor may
be placed from gate to ground, however the value should
be reasonably high to minimize DC losses and possible AC
issues.

– 1.6V.

IN

above 6.5V, the gate drive is

needs to

GCL

The gate drive source current comes from VIN. The sink
current exits through PGND. In general the decoupling cap
should be placed close to these two pins.

Switching Diodes

In general, switching diodes should be Schottky diodes.
Size and breakdown voltage depend on the specific con­verter. A lower forward drop will improve converter effi­ciency. No other special requirements are needed.

PCB LAYOUT CONSIDERATIONS

As with any switcher, careful consideration should be given
to PC board layout. Because this part reduces high fre­quency EMI, the board layout is less critical. However, high
currents and voltages still produce the need for careful
board layout to eliminate poor and erratic performance.

Basic Considerations

Keep the high current loops physically small in area. The
main loops are shown in Figure 6: the power switch loops
(A) and the rectifier loop (B). These loops can be kept small
by physically keeping the components close to one an­other. In addition, connection traces should be kept wide
to lower resistance and inductances. Components should
be placed to minimize connecting paths. Careful attention
to ground connections must also be maintained. Be care­ful that currents from different high current loops do not
get coupled into the ground paths of other loops. Using
singular points of connection for the grounds is the best
way to do this. The two major points of connection are the
bottom of the input decoupling capacitor and the bottom
of the output decoupling capacitor. Typically, the sense
resistor device PGND and device GND will tie to the bottom
of the input capacitor.

V

IN

C

IN

GATE

CS

•

Figure 6

C

V

OUT

BA

•

OUT

1738 F06

18

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LT1738

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APPLICATIO S I FOR ATIO

There are two other loops to pay attention to. The current
slew involves a high bandwidth control that goes through
the MOSFET switch, the sense resistor and into the CS pin
of the part and out the GATE pin to the MOSFET. Trace
inductance and resistance should be kept low on the GATE
drive trace. The CS trace should have low inductance.

Finally, care should be taken with the CAP pin. The part will
tolerate stray capacitance to ground on this pin (<5pFs).
However, stray capacitance to the MOSFET drain should
be minimized. This path would provide an alternate ca­pacitive path for the voltage slew.

U

PACKAGE DESCRIPTIO

G Package

20-Lead Plastic SSOP (5.3mm)

(Reference LTC DWG # 05-08-1640)

MORE HELP

AN70 contains information about low noise switchers and
measurement of noise and should be consulted. AN19 and
AN29 also have general knowledge concerning switching
regulators. Also, our Application Department is always
ready to lend a helping hand.

1.25 ±0.12

7.8 – 8.2

0.42 ±0.03 0.65 BSC

RECOMMENDED SOLDER PAD LAYOUT

5.00 – 5.60**
(.197 – .221)

0.09 – 0.25

(.0035 – .010)

NOTE:

1. CONTROLLING DIMENSION: MILLIMETERS

2. DIMENSIONS ARE IN

3. DRAWING NOT TO SCALE
*

DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED .152mm (.006") PER SIDE

**

DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED .254mm (.010") PER SIDE

0.55 – 0.95

(.022 – .037)

MILLIMETERS

(INCHES)

5.3 – 5.7

° – 8°

0

6.90 – 7.50*
(.272 – .295 )

1718 14 13 12 1115161920

12345678910

0.65

(.0256)

BSC

0.22 – 0.38

(.009 – .015)

(.291 – .323)

2.0

(.079)

0.05

(.002)

G20 SSOP 0802

7.40 – 8.20

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen­tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

1738fa

19

LT1738

U

TYPICAL APPLICATIO

Ultralow Noise 30W Offline Power Supply

DANGER: HIGH VOLTAGE

X1

1M

90VAC

TO 264VAC

X3

12V
IN755 A

165k

51k

UNLESS OTHERWISE NOTED: ALL RESISTORS 1206, 5%
BR1: GENERAL INSTRUMENTS W06G
C2, C3, C4: SANYO MV-GX

INPUT FILTER IS REQUIRED TO ATTENUATE SWITCHING FREQUENCY HARMONICS AND PASS FCC CLASS B (LT1738 DOES NOT ATTENUATE THESE LOW FREQUENCY HARMONICS)
MAIN ADVANTAGE WITH LT1738 IS IT MAKES SUPPRESSING THE HIGH FREQUENCY NOISE AND EMI EASY. THIS IS PARTICULARLY USEFUL FOR MEDICAL DEVICES BECAUSE
THE AC LINE TO EARTH GND CAPS ON THE INPUT FILTER CAN BE ELIMINATED; ALLOWING THE DEVICE TO PASS THE EARTH GND LEAKAGE CURRENT MEDICAL SPECIFICATIONS.

250VAC

1M

510k

2N2222

0.1µF

“X2”

2N2222

510k

165k

V

BIAS

L1

+

100µF

BR1

100k
2W

+

56µF
35V

1.8nF

19.6k

3.9k

3.9k

17 19 10

V

IN

14

SHDN

5

V5

6

SYNC

7

C

T

8

R

T

16

R

VSL

15

R

CSL

9

F

B

SS

13 11 3

10Ω

NC NFB

CAP

GATE

U3

LT1738

GND

10nF

D1: MBR20200CT
L1: HM18-10001
T1: PREMIER MAGNETICS POL-15033

PGND

GCL

V

BIAS

CS

NC

V

C

400V

470pF

5pF

2

1

4

18

20

12

P6KE200A

MUR160

KA

D4

BA521

5pF

600V

MTP2N60E

0.068Ω

1/2W

ISO1

CNY17-3

6
5

4

NOTE: PIN 2 OF LT1738 MUST BE LAID OUT
AWAY FROM FAST SLEWING NODES

T1

11

1

7

3
6

12

5

8

1k

1

3

2

D1

A1

A2

+

C2

330µF

25V

0.1µF

3

+

COMP

V

U2

LT1431

COLL

G-F G-S

65

1k

2

REF

R

TOP

RMIO

+

C3
330µF
25V

8
4
7

+V

OUT

12V

2.5A

C4

+

330µF
25V

–V

OUT

V

OUT

0.22µF

38.3k
1%

10k
1%

1k

1738 TA02

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DirectSense is a trademark of Linear Technology Corporation.

1738fa

LT/TP 1202 1K REV A • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 2001

20

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone]

●

www.linear.com