Fairchild AN-7511 User Manual

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Insulated-Gate Transistors Simplify AC-Motor

Application Note September 1993 AN-7511

An IGT’s few input requirements and low On-state resistance simplify drive circuitry and increase power efficiency in motorcontrol applications. The voltage-controlled, MOSFET-like input and transfer characteristics of the insulated-gate transistor (IGT) (see EDN, September 29, 1983, pg 153 for IGT details) simplify power-control circuitry when compared with bipolar devices. Moreover, the IGT has an input capacitance mirroring that of a MOSFET that has only one-third the powerhandling capability. These attributes allow you to design simple, low-power gate-drive circuits using isolated or level-shifting techniques. What’s more, the drive circuit can control the IGT’s switching times to suppress EMI, reduce oscillation and noise, and eliminate the need for snubber networks.

Use Optoisolation T o Avoid Ground Loops

The gate-drive techniques described in the following sections illustrate the economy and flexibility the IGT brings to power control: economy, because you can drive the device’s gate directly from a preceding collector, via a resistor network, for example; flexibil ity, because you can choose the drive circuit’s impedance to yield a desired turn-off time, or you can use a switchable im pedance that causes the IGT to act as a chargecontrolled device requiring less than 10 nanocoulombs of drive charge for full turn-on.

Take Some Driving Lessons

Note the IGT’s straightforward drive compatibility with CMOS, NMOS and open-collector TTL/HTL logic circuits in the common-emitter configuration Figure 1A. R off time, and the sum of R

and the parallel combination of R

and R2 sets the turn-on time. Drive-circuit requirements, however, are more complex in the common-collector configuration Figure 1B.

In this floating-gate-supply floating-control drive scheme, R controls the gate supply’s power loss, R2 governs the turn-off time, and the sum of R

and R2 sets the turn-on time. Figure

1C shows another common-collector configuration employing a bootstrapped gate supply. In this configuration, R3 defines the turn-off time, while the sum of R

on time. Note that the gate’s very low leakage allows the use of low-consumption bootstrap supplies using very low-value capacitors. Figure 1 shows two of an IGT’s strong points. In the common-emitter Figure 1A, TTL or MOS-logic circuits can drive the device directly. In the common-collector mode, you’ll need lev el s hifting, using either a second power suppl y Figure 1B or a bootstrapping scheme Figure 1C.

controls the turn-

and R3 controls the turn-

Speed Control

OFF

FIGURE 1A.SIMPLE DRIVING AND TRANSITION-TIME

R1 CONTROLS GATE SUPPLY POWER LOSS

CONTROLS t

R1 + R2 CONTROLS t

CONTROL

FIGURE 1B. A

OFF ON

FIGURE 1C. BOOTSTRAPPING SCHEME

CONTROL

OFF

INPUT

OFF

SECOND POWER SUPPLY

LOAD

VCCR

------------------- -

≤≤

R1R

R3 CONTROLS t

15V

LOAD

VCCR

------------------- -

≤≤

R1R

R3 CONTROLS t R2 + R3 CONTROLS t

-------------------------------------------------«

CEOIGES

25V

OFF

LOAD

25V

OFF

through the logic circuit’s ground can create problems. Optoisolation can solve this problem (Figure 2A.) Because of the high common-mode dV/dt possible in this configuration, you should use an optoisolator with ver y low isolation capacitance; the H11AV specs 0.5pF maximum.

In the common-collector circuits, power-switch current flowing

Application Note 7511

For optically isolated “relay-action” switching, it makes sense to replace the phototransistor optocoupler with an H11L1 Schmitt-trigger optocoupler (Figure 2B).) For applications requiring extremely high isolation, you can use an optical f iber to provide the signal to the gate-control photodetector. These circuit examples use a gate-discharge resistor to control the IGT’s turn-off time. To exploit fully the IGT’s safe operating area (SOA), this resistor allows time for the device’s minority carriers to recombine. Furthermore, the recombination occurs without any current crowding that could cause hot-spot formation or latch-up pnpn action. For very fast turn-off, you can use a minimal snubber network, which allows the saf e use of lower value gate resistors and higher collector currents.

CONTROL

INPUT

OFF ON

FIGURE 2A. AVOID GROUND-LOOP PROBLEMS BY USING AN

OPTOISOLATOR. THE ISOLATOR IGNORES SYSTEM GROUND CURRENTS AND ALSO PROVIDES HIGH COMMON-MODE RANGE.

H11AV2

LOAD

directly from TTL levels, thanks to its 1.2V, 20mA input parameters.

Available photovoltaic couplers have an output-current capability of approximately 100µA. Combined with approximately 100kΩ equivalent shunt impedance and the IGT’s input capacitance, this current level yields very long switching times. These transition times (typically ranging to 1 msec) vary with the photovoltaic coupler’s drive current and the IGT’s Miller-effect equivalent capacitance.

Figure 3 illustrates a typical photovoltaic-coupler drive along with its transient response. In some applications, the photovoltaic element can charge a storage capacitor that’s subsequently switched with a phototransistor isolator. This isolator technique - similar to that used in bootstrap circuits pro v id e s rapid tu rn- o n and tu rn - o f f w hi l e maintai n in g s m a l l s i ze, good isolation and low cost.

In common-collector applications involving high-voltage, reactive-load switching, capacitive currents in the low-level logic circuits can flow through the isolation capacitance of the control element (eg, a pulse transformer, optoisolator, piezoelectric coupler or level-shift transistor). These currents can cause undesirable effects in the logic circuitry, especially in highimpedance, low-signal-level CMOS circuits.

OFF

CONTROL

INPUT

DIG22

IGT

VCC = 300V

43k

1N5061

CONTROL

INPUT

OFF ON

FIGURE 2B. A SCHMITT-TRIGGER OPTOIS OLATOR YIELDS

10µF

35V

H11L1

“SNAP-ACTION” TRIGGERING SIMILAR TO THAT OF A RELAY.

5.6k 5.6k

5.6k

LOAD

Pulse-Transformer Drive Is Cheap And Efficient

Photovoltaic couplers provide yet another means of driving the IGT. Typically, these devices contain an array of small silicon photovoltaic cells, illuminated by an infrared diode through a transparent dielectric. The photovoltaic coupler provides an isolated, controlled, remote dc supply without the need for oscillators, rectifiers or filters. What’s more, you can drive it

OUTPUT

CURRENT

INPUT

CURRENT

FIGURE 3. AS ANOTHER OPTICAL-DRIVE OPTION, A PHOTO-

VOLTAIC COUPLER PROVIDES AN ISOLATED, REMOTE DC SU PPIY TO THE IGT’S INPUT. ITS LOW 100µA OUTPUT, HOWEVER, YIELDS LONG IGT TURN-ON AND TURN-OFF TIMES.

012ms

The solution? Use fiber-optic components Figure 4 to eliminate the problems completely. As an added feature, this lowcost technique provides physical separation between the power and logic circuitry, thereby eliminating the effects of radiated EMI and high-flux magnetic fields typically found near power-switching circuits. You could use this method with a bootstrap-supply circuit, although the fiber-optic system’s reduced transmission efficiency could require a gain/speed trade-off. The added bipolar signal transistor minimizes the pot enti al for compromise.

CONTROL

)

INPUT

OFF

EMITTER

(DISCONNECTED)

Application Note 7511

1N914

GFOE1A1

2N5354

10M (30FT)

QSF2000C

(W/CONNECTORS)

GFOD1A1

IGT

DETECTOR

(CONNECTED

FIGURE 4. ELIMINATE EMI IN HIGH-FLUX OR NOISE ENVI-

RONMENTS BY USING FIBER-OPTIC COMPO NENTS. THESE PARTS ALSO ALLEVIAT PROBLEMS ARISING FROM CAPACITIVE COU PLING IN ISOLATION ELEMENTS.

iezos Pare Prices

ACOUSTIC WAVE

OUTPUT VOLTAGE

OSCILLATOR

FIGURE 5A. YIELDING 4-kV ISOLATION, A PIEZOELECTRIC COUPLER PROVIDES TRANSFORMER-LIKE PERFORMANCE AND AN

ISOLATED POWER SUPPLY.

2.5k

3.3k

2.7k

0.001 µF

18V

NE555

0.001 µF

PZ61343

D33D21

1N914

IGT

4.7k

1N914

FIGURE 5B. THIS CIRCUIT PROVIDES THE DRIVE FOR THIS ARTICLE’S MOTOR-CONTROL CIRCUIT.

Application Note 7511

A piezoelectric coup ler operationally similar to a pulse-train drive transformer, but potentially less costly in high volume is a small, ef ficient device with iso lation capability ranging to 4kV. What’s more, unlike optocouplers, they require no auxiliary power supply. The piezo element is a ceramic component in which electrical energy is converted to mechanical energy, transmitted as an acoustic wave, and then reconver ted to electrical energy at the output terminals Figure 5A.

The piezo element’s maximum coupling efficiency occurs at its resonant frequency, so the control oscillator must operate at that freq uen cy. For example, the PZT61343 piezo c oup ler in Figure 5B’s driver circuit requires a 108kHz, ±1%-accurate astable multivibrator to maximize mechanical oscillations in the ceramic material. This piezo element has a 1W max power handling capability and a 30mA p-p max secondar y current rating. The 555 timer shown provides compatible waveforms while the R C ne twork sets the frequency.

Isolate With Galvanic Impunity

Do you require tried and true isolation? Then use transformers; the IGT’s low gate requirements simplify the design of independent, transformer-coupled gate-drive supplies. The supplies can directly drive the gate and its discharge resistor Figure 6, or they can simply replace the level-shifting supplies of Figure 2. It’s good practice to use pulse transformers in drive circuitry, both for IGT’s and MOSFETs, because these components are economical, rugged and hi ghly reliable.

OFF

TRANSFORMER

CONTROL

INPUT

PULSE

1N914

1N914 2N5232

IGT

FIGURE 6A. PROVIDING HIGH ISOLA TION A T LO W COST , PUL SE

TRANSFORMERS ARE IDEAL FOR DRIVING THE IGT. AT SUFFICIENTLY HIGH FREQUENCIES, C CAN BE THE IGT’S GATE-EMITTER CAPA CITANCE ALONE.

ON OFF

CONTROL

INPUT

1N914

IGT

1N914 RC = 3µSEC

FIGURE 6B. A HIGH-FREQUEN CY OSCI LL ATOR IN THE TRANS -

FORMER’S PRIMARY YIELDS UNLIMITED ONTIME CAP ABILITY.

In the pulse-on, pulse-off method Figure 6A, C1 stores a positive pulse, holding the IGT on. At moderate frequencies (several hundred Hertz and above), the gate-emitter capacitance alone can store enough energy to keep the IGT on; lower frequencies require an additional external capacitor. Use of the common-base n-p-n bipolar transistor to discharge the capacitance minimizes circuit loading on the capacitor. This action extends continuous on-time capability without capacitor refreshing; it also controls the gate-discharge time via the 1kΩ emitter resistor.

VARIABLE

220V AC

3φ 60Hz

THREE-PHASE

BRIDGE

RECTIFIER

LOW VOLTAGE

TRANSFORMER

RECTIFIER

FILTER

SIGNAL PATH ISOLATOR

EG: OPTOCOUPLIER PIEZO COUPLER

24V DC

SWITCHING

REGULATOR

POWER SUPPLY

FOR CONTROL

CIRCUITS

VOLTAGE ENABLE

ADJUST VOLTAGE

VOLTAGE

CONTROLLED

OSCILATOR

24V

III

MOTOR

CONTROL

LOGIC

DC VOLTAGE

TIMING

AND DRIVE

SHUT DOWN DRIVE OSCILLATOR

THREE-PHASE

INVERTER

ENABLE

LOWER

LEGS

OVERLOAD

PROTECTION

IGT

3φ

INDUCTION

MOTOR

CURRENT SENSE SIGNAL

TACHOMETER

FEEDBACK

FIGURE 8. THIS 6-STEP 3-PHASE-MOTOR DRIVE USES THE IGT-DRIVE T ECHNIQUES DESCRIBED IN THE TEXT. THE REGULATOR AD-

JUSTS THE OUTPUT DEVICES’ INPUT LEVELS; THE VOLTAGE-CONTROLLED OSCILLATOR VARIES THE SWITCHING FREQUENCY AND ALSO PROVIDES THE CLOCK FOR THE 3-PHASE TIMING LOGIC. THE V/F RATIO STAYS CONSTANT TO MAINTAIN CONSTANT TORQUE REGARDLESS OF SPEED.

Application Note 7511

Piezoelectric Couplers Provide 4-kV Isolation

Using a high-frequency oscillator fo r pulse-train drive Figure 6B yields unlimited on-time capability. However, the scheme requires an oscillator that can be tur ned on and off by the control logic. A diode or zener clamp across the transformer’s primary will limit leakage -inductance flyback effects . To optimize transformer efficiency, make the pulses’ voltage x time products equal for both the On and the Off pulses. In situations where the line voltage generates the drive power, a simple relax ati on os cil lat or u sing a progra mm able uni junc tion transistor can der ive its power directly from the line to provide a pulse train to the IGT gate.

The circuit shown in Figure 7 accommodates applications involving lower frequencies (a few hundred Hertz and below). The high oscillator frequency (greater than 20kHz) helps keep the pulse transformer reasonably small. The volt-

325V 10A

age-doubler circu itry impro ves the turn-on time and also provides long on-time capability. Although this design uses only a 5V supply on the primary side of a standard trigger transformer, it provides 15V gate-to-emitter voltage.

1N914

OSCILLATOR

1:2

1N914

FIGURE 7. THIS DRIVING METHOD FOR LOW-FREQUENCY

SWITCHING PROVIDES 15V TO THE IGT’S GATE

NOTES:

- Q6 = D94FR4

- D7 = 1N3913

- D13 = 1N914

R = 4.7k, C L

= 100µF, 400V

= 40µH

/2W

0.001 µF

4.7k

0.001µF

INDUCTION

MOTOR

220V

IGT

FIGURE 9A. THE POWER INVERTER’S DRIVE CIRCUIT USES SIX IGTS TO DRIVE A 2-HP MOTOR.

180

Q1 ON

15o DELAY

Q2ON

o×

OF EVERY 360o CYCLE;

φA

φB

φC

FIGURE 9B. THE TIMING DIAGRAM SHOWS THAT EACH IGT

CONDUCTS FOR 165 THE DELAY IS NECESSARY TO AVOID CROSS CONDUCTION.

FIGURE 9C. THE THREE WINDINGS’ VOLTAGES AND CUR-

RENTS ARE SHOWN. NOTE THAT ALTHOUGH COSTLY SNUBBER NETWORKS ARE ELIMINATED, FREEWHEELING DIODES ARE NEEDED; THE IGTS HAVE NO INTRINSIC OUTPUT DIODE.







   



  

Application Note 7511

Polyphase motors, controlled by solid-state, adjustable-frequency ac drives, are used extensively in pumps, conveyors, mills, machine tools and robotics applications. The specific control method could be either 6-st ep or pulse-width modulation. This section describes a 6-step drive that uses some of the previously discussed drive techniques (see page 11 , “Latch-Up: Hints, Kinks and Caveats”).

Figure 8 defines the drive’s block diagram. A 3-phase rectifier converts the 220V ac to dc; the switching regulator varies the output voltage to the IGT inverter. At the regulator’s output, a large filter capacitor provides a stiff voltage supply to the inverter.

The motor used in this example has a low slip characteristic and is therefore very efficient. You can change the motor’s speed by varying the inverter’s frequency. As the frequency increases, however, the motor’s air-gap flux diminishes, reducing developed-torque capability. You can maintain the flux at a constant level (as in a dc shunt motor) if you also vary the voltage so the V/F ratio remains constant.

Fiber-Optic Drive Eliminates Interference

In the example given, the switching regulator varies the IGT inverter’s output by controlling its dc input; the voltage-controlled oscillator (VCO) adjusts the inverter’s switching frequency, thereby varying the output frequency. The VCO also drives the 3-phase logic that provides properly timed pulsed outputs to the piezo couplers that directly drive the IGT.

Sensing the dc current in the negative rail and inhibiting the gate signa l protect the IGT from overload and shoot-through

(simultaneous conduction) conditions. If a fault continues to exist for an appreciable period, inhibiting the switching regulator causes the inv erter to shut off. The inverter’s pow er-ou tput circuit is shown in Figure 9A; the corresponding timing diagrams show resistive-load current waveforms that indicate the 3-phase power Figure 9B and waveforms of the output line voltage and cu rrent Figure 9C.

In Figure 9’s circuit, it appears that IGTs Q conduct for 180

. Howe v er, in a practical situation, it’ s ne ces-

sary to provid e some t ime de lay (typic ally 1 0

through Q6 will

to 15o×) during the positive-to-negative transition periods in the phase current. This delay allows the complementary IGTs to turn off before their opposite members turn on, thus preventing cross conduction and e ventual destruction of the IGTs.

Because of the time delay, the maximum conduction time is

of every 360o period. B ecaus e the IG Ts do n’t have an

165 integral diode, it’s necessary to connect an antiparallel diode externally to allow the freewheeling current to flow. Inductor

limits the di/dt during fault conditions; freewheeling diode

clamps the IGT’s collector supply to the dc bus .

The peak full-load line current specified by the motor manufacturer determines the maximum steady-state current that each transistor must switch. You must convert this RMSspecified current to peak values to specify the proper IGT. If the input voltage regulator had a fixed output voltage and a constant frequency, each IGT would be required to supply the starting locked-rotor current to the motor. This current could be as much as 15 times the full-load running current.

0 TO 325V 10A

FIGURE 10A. COMPONENT SELECTION IS IMPORTANT. THE IGT SELECTED CIRCUIT HANDLES 10A, 500V AT 150oC. THE ANTI-

PARALLEL DIODES HAVE A SIMILAR CURRENT RATING.

0.1

100

D(OFF)

1k 10k

0.9I

0.1I

SWITCHES ON” (1, 4, 5), (1, 3, 6), (2, 3, 6),

(2, 3, 5), (2, 4, 5)

TO LOAD

FIGURE 10B. SELECT R TO YIELD THE DESIRED TURN-OFF TIME. FINALLY, L1’S VALUE DETERMINES THE FAULT-CONDITION

ACTION TIME.

Application Note 7511

It’s impr ac tic al, however, to ra te an inverter based on lockedrotor current. You can avoid this necessity by adjusting the switching regulator’s output voltage and by providing a fixed output-current limit slightly higher than the maximum fullload current. This way, the current requirements during startup will never exceed the current capability of an efficiently sized inverter.

For example, consider a 2-hp, 3-phase induction motor specifying VL at 230V RMS and full-load current (I

2.7k

1000pF

3.3k

VCO &

TIMING

LOGIC

2.5k

NE555

A4.7k

1N914

0.001µF

470

Q 2N3903

) at 6.2A

LFL

1N914

4.7k

470

RMS. For the peak current of 8.766A, you can select IGT type D94FR4. This device has a reverse-breakdown SOA (RBSOA) of 10A, 500V for a clamped inductive load at a junction temper ature o f 1 50

C. A 400V IGT coul d als o do the job, but the 500V choice gives an additional derating safety margin. You must set the current limit at 9A to limit the inrush current during start-up. Note that thanks to the IGT’s adequate RBSOA, you don’t need turn-off snubbers.

24V

22µF

2N3903

1N914

PIEZOCOUPLER

PZT61343

1N914

2N3903

1N914

4.7k

D29E10

2N3903

24V

470

D33030 Q

DC BUS

D94FR4

φA

D94FR4

FIGURE 11A. PROVIDING PROPERLY TIMED DRIVE TO THE IGTS, THE CIRCUIT USES PIEZO COUPLING TO THE UPPER POWER

DEVICE. THE 3-TRANSISTOR DELAY CIRCUIT PROVIDES THE NEEDED 15

LAG TO THE LOWER IGT TO AVOID

CROSS CONDUCTION.

VOLTS

24V

100kHz

TIME

FIGURE 11B. THE TIMING DIAGRAM SHOWS THE 555’S 108-KHz DRIVE TO THE PIEZO DEVICE AND THE LATTER’S SLOW

RESPONSE.

Application Note 7511

Use 6-Step Drive For Speed-Invariant Torque

Figure 10A shows the inverter circuit configured for this example. Diodes D as the IGTs; consequently, they’re rated to handle peak currents of at least 8.766A. However, they only conduct for a short time (15 requirement is relati vely small.

External circuitry can control the IGT’s current fall time. Resistor R co ntr o ls t

, an inherent characteristic of the selected IGT. In this

example, a 4.7-kΩ gate-to-emitter resistor provides the appropri ate fall time. Th e choice of cu rrent-limiti ng inductor

is based on the IGT’s overload-current rating and the

action time (the sum of the sensor’s sensing and response time and the IGT’s turn-off time) in fault conditions.

through D6 carry the same peak curr ent

to 20o of 180o), so their average-current

Figure 10B; there's no way to control

You could use a set of flip flops and a multivibrator to generate the necessary drive pul se s and the corresponding 120o× delay between the three phases in Figure 10’s circuit. A voltage-controlled oscillator serves to change the inverter’s output frequency. In this circuit, IGTs Q isolated gate drive; the drive for Q

, Q3 and Q5 require

, Q4 and Q6 can be

referred to common. If you use optocouplers for isolation, you’ll need three isolated or bootstrap power supplies (in addition to the 5V and 24V po w er suppli es) t o drive th e IGTs . Another alternative is to use transformer coupling.

Conduction Prevents Shoot-Through

165

Consider, however, using Figure 11A’s novel, low-cost circuit. It uses a piezo coupler to drive the isolated IGT. As noted, the coupler needs a high-frequency square wave to induce mec hanical oscill ations in its primary side. The 555 oscillator provides th e necessary 108-kHz waveform; its output is gated ac cording to the required timi ng logic and th en applied to the pie z o c oupler’s primary. The coupler’s rectified output drives the IGT’s gate; the 4.7kW gate-to-emitter resistor provides a discharge path for C

during the IGT’s turn-

off. The circuit’s logic-timing diagram is shown in Figure 11B. The piezo coupler’s slow response time Figure 12A contrib-

utes appro ximately 2

to the 15o to 20o turn-on/turn-off delay needed to avoid shoot-through in the complementary pairs. The corresp ondin g coll ector c urre nt is s hown in F igure 1 2B.

and its associated circuitry provide the remaining delay

as follows:

FIGURE 12A. THE PIEZO COUPLER’S SLOW RESPONSE IS NOT

FIGURE 12B. THE DRIVEN IGT 'S COLLEC T OR CURRENT IS

A DISADVANTAGE IN THIS ARTICLE’S CIRCUIT. IN F ACT, IT CONTRIBUTES 2 TURN-ON/TURN-OF F DEL AY.

TRACE VERTICAL HORIZONTAL

A 5V/DIV 200µSEC/DIV B 5V/DIV 200µSEC/DIV

SHOWN

TRACE VERTICAL HORIZONTAL

A 3A/DIV 200µSEC/DIV B 5V/DIV 200µSEC/DIV

TO THE REQUIRED 15

When Q3’s base s wings negative, C1 - at this time discharged turns on Q pulse to turn the IGT on. When Q

. Once C1 is charged, Q5 turns off, allowing a drive

’s base goes to ground, Q

turns on and discharges C1, initiating the IGT’s turn-off. Figure 13 shows the motor current and corresponding line voltage under light-load Figure 12A and full-load Figure 12B conditions.

Application Note 7511

FIGURE 13A. MOTOR CURRENT AND VOLTAGE ARE SHOWN

HERE, FOR LIGHT LOADS

TRACE VERTICAL HORIZONTAL

A 1A/DIV 1mSEC/DIV B 50V/DIV 1mSEC/DIV

FIGURE 13B. MOTOR CURRENT AND VOLTAGE ARE SHOWN

HERE, FOR HEAVY LOADS.

TRACE VERTICAL HORIZONTAL

A 3A/DIV 2mSEC/DIV B 100V/DIV 2mSEC/DIV

To complete the design of the 6-step motor drive, it’s necessary to consider protection circuitry for the output IGTs. The drive receives its power from a switching supply already containing provisions for protection from line over-voltage and under-voltage and transient effects. However, you still have to guard the power switches against unwanted effects on the output lines and the possibility of noise or other extraneous signals causing gate-drive timing errors.

The best protection circuit must match the character istics of the power switch and the circuit’s bias conditions. The IGT is very rugged during turn-on and conduction, but it requires time to dissipate minority carriers when turning off high currents and voltages. An analysis of the possible malfunction condition

LINE

INPUT

RECTIFIER

AND

FILTER

SWITCHING

POWER SUPPLY

50 TO 320V DC

5V DC

ADJUST

CONTROL

AND

TIMING

ISOLATIONISOLATION

24V DC

ISOLATION

dI/dt LIMIT

UPPER 3 LOWER 3

IGT

SWITCHES

MOTOR

CHOPPER

RECYCLE

TIME

DISABLE

GATE DRIVE TURNOFF

COMPARATOR

AND LATCH

I LIMIT

10A

20A

AV = 100

FIGURE 14. THE LOWEST COST SENSOR IMAGINABLE, A PIECE OF COPPER WIRE SERVES AS THE CURRENT MONITOR IN THIS SYS-

TEM. THE CHOPPED AND AMPLIFIED VOLTAGE DROP ACROSS THE WIRE TRIGGERS A GATE-DRIVE SHUT-OFF CIRCUIT UNDER FAULT CONDITIONS.

50 TO 320V DC

24V

3.9k

5µF

25V

15V

750k

220k

2N5232

H11F3

2.2k

0.001 µF

3.3k

470pF

0.01µF

10A

20A

150

5µF 20V

H11F3 TO CONTROL CIRCUIT

Application Note 7511

2N5355

180k

2.7k

2N5306

390

470 pF

10k

C203B

0.02

µF

H11AV2 TO PZO SHUTDOWN H11AV2 TO HI-V SHUTDOWN

22k

A139M

47k

0.001 µF

50µH

DRIVE

39k

DT230F

0.2µF

TO MOTOR

2mΩ (1” #24 AWG COPPER)

POWER SUPPLY

CURRENT

SENSE AND

CHOPPER

AMPLIFIER

LATCHING

FAST

COMPARATOR

10ms

RESET

IGT POWER SWITCHES

FIGURE 15A. THIS ALL-ENCOMPASSING PROTECTION SYSTEM PROVIDES THREE INDEPENDENT SHUTDOWN FUNCTONS -

ONE EACH FOR THE UPPER AND LOWER IGTS AND THE HIGH-VOLTAGE SUPPLY.

TIMER

10µF 10V

555

47k

H11AV2

CHOPPER DRIVE

TO PIEZO

DRIVERS

DT230F

(3)

150

0.005µF

0.002 µF

180

2N5232

5.1k

150150

5.1k

2N5232

100

H11F3H11F3

0.002 µF

180

HIGH-VOLTAGE SHUTDOWN

24V

0.005 µF

2.2k 16

SG3524

CONTRO

ALL OPTOCOUPLERS GO TO PROTECTION CIRCUIT

FIGURE 15B. THIS CIRCUIT PROVIDES CHOPPER DRIVE FOR THE COPPER-WIRE

SENSOR IN FIGURE 15A.

FIGURE 15C. SHOWS THE HIGH-VOLT-

AGE SHUTDO WN C IRCUI T.

Application Note 7511

Latch-Up: Hints, Kinks and Caveats

The IGT is a rugged device, requiring no snubber network when operating within its published safe-operating-area (SOA) ratings. Within the SOA, the gate emitter voltage controls the collector current. In fact, the IGT can conduct three to four times the published maximum current if it’s in the ON state and the junction temperature is +150

C maximum.

However, if the current exceeds the rated maximum, the IGT could lose gate control and latch up during turn-off attempts. The culprit is the parasitic SCR formed by the pnpn structure shown in Figure 16. In the equivalent circuit, Q MOSFET with a normal parasitic transistor (Q emitter junction is shunted by the lo w-value resistance R

EMITTER METAL POLYSILICON GATE

N EPITAXIAL LAYER

MAIN CURRENT PATH

P+ SUBSTRATE

N+N+

MINORITY

CARRIER

INJECTION

is a power

) whose base-

Forward-Bias Latch-Up

Within the IGT’s current and junction-temperature ratings, current does not flow through Q2 under forward-biased conditions. When the current far exceeds its rated value, the current flow through R

increases and Q3’s VCE also

increases because of MOSFET channel saturation. Once Q3’s ICR1 drop exceeds Q2’s V

, Q2 turns on and

BE(ON)

more current flow bypasses the FET. The positive feedback thus established causes the device to

latch in the forward-biased mode. The value of I

at which

the IGT latches on while in forward conduction is typically three to four times the device’s maximum rated collector current. When the collector current drops below the value that provokes Q

turn-on , normal operation resu mes if chip

temperature is still within ratings. If the gate-to-emitter resistance is too low, the Q

2-Q3

parasitic SCR can cause the IGT to latch up during turn-off. During this period, R power MOSFET Q this increases Q

determines the drain-so urce d V/dt of

. A low R1 causes a rapid rise in v ol tag e -

’s VCE, increasing both R1’s value and Q2’s

gain. Because of st orage time, Q

’s collector current continues to

flow at a level that’s higher than normal for the FET bias. During rapid turn-off, a portion of this current could flow in

’s base-emitter junction, causing Q2 to conduct. This

process results in device latch-up; current distribution will probably be less uniform than in the case of forward-bias latch-up.

METAL COLLECTOR

COLLECTOR

GATE

FIGURE 16. THE IGT’S P ARASITIC SCR IS RESPONSIBLE FOR

THE DEVICE’S LATCH-UP CHARACTERISTICS.

EMITTER

For large current overloads, the current flowing through R can provoke SCR triggering. In the simplest terms, R1 represents the equivalent of a distributed resistor network, whose magnitude is a function of Q2’s VCE. During normal IGT operation, a positive gate voltage (greater than the threshold) applied betwee n Q1’s gat e and so urce turns the FET o n. The FET then turns on Q

(a pnp transistor with very low

gain), causing a small portion of the total collector current to flow through the R1 network.

To turn the IGT off, you must reduce the gate-to-emitter voltage to zero. This turns Q1 off, thus initiating the turn-off sequence within the device. Total fall time includes currentfall-time one (tF1) and current-fall-time two (tF2) components. The turn-off is a function of the gate-emitter resistance, Q3’s storage time and the value of VGE prior to turn-off. Device characteristics fix both the delay time and the fall time.

Because the gains of Q2 and Q3 increase with temperature and V

, latching current - high at +25oC - decreases as a

function of increasing junction temperature for a given gateto-emitter resistance.

How do you test an IGT’s turn-off latching characteristic? Consider the circuit in Figure 17. Q

’s base-current pulse

width is set appro xim ately 2 µsec great er than the IGT’s gatevoltage pulse width. This way, the device under test (DUT) can be switched through Q

when reverse-bias latch-up

occurs. This circuit allows you to test an IGT’s latching current nondestructively.

The results? Clamped-inductive-load testing with and

without snubbers reveals that snubbering increases current handling dramatically: With RGE = 1kΩ, a 0.02µF snubber capacitor increases current capability from 6A to 10A; with RGE = 5kΩ, a 0.09µF snubber practically doubles capacity (25A vs 13A).

Conclusions? You can double the IGT’s latching current by increasing RGE from 1kΩ to 5kΩ, and double it again with a polarized snubber using CS < 0 .1µF. The IGT is therefore useful in situations where the device must conduct currents of five to six times normal levels for short periods.

Finally, you can also use the latch ing beh a vior to y o ur adv a ntage under fault conditions. In other words, if the device latches up during turn-off under normal operation, you could arrange it so that a suitable snubber is switched electronically across the IGT.

Application Note 7511

L = 100µH

A139M

0.02µF

A139P

10µF

CLAMP

(400V MAX)

PULSE

GENERATOR

PULSE

GENERATOR

= D66EV7

= DUT D94FQ4

A114A

TRIGGER 1000pF

A114A

100

1N914

10V

D38H1

PE-63385

A114A

D44D6

DS0026x2

15V

D66EV7

R 1-10k

FIGURE 17. USE THIS LATCHING-CURRENT TESTER TO TEST IGTS NONDESTRUCTIV ELY. Q1’S BASE-DRIVE PULSE WIDTH IS GREAT-

ER THAN THAT OF THE IGT’S GATE DRIVE, SO THE IGT UNDER TEST IS SWITCHED THROUGH Q

WHEN REVERSE-BIAS

LATCH-UP OCCURS.

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FACT FACT Quiet Series

STAR*POWER is used under license

FAST FASTr FRFET GlobalOptoisolator GTO HiSeC

C ISOPLANAR LittleFET MicroFET MicroPak

MICROWIRE OPTOLOGIC

OPTOPLANAR PACMAN POP Power247 PowerTrench

QFET QS QT Optoelectronics Quiet Series

SILENT SWITCHER SMART START SPM STAR*POWER Stealth SuperSOT-3 SuperSOT-6 SuperSOT-8 SyncFET TinyLogic TruTranslation

ââ

UHC UltraFET VCX

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Advance Information

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No Identification Needed

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2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or

effectiveness.

This datasheet contains the design specifications for product development. Specifications may change in any manner without notice.

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Rev. H5

Fairchild AN-7511 User Manual

Specifications and Main Features

Frequently Asked Questions

User Manual