Datasheet ne5210 DATASHEETS (Philips)

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1

1995 Apr 26 853-1654 15170

DESCRIPTION

The NE5210 is a 7kΩ transimpedance wide band, low noise amplifier with differential outputs, particularly suitable for signal recovery in fiber-optic receivers. The part is ideally suited for many other RF applications as a general purpose gain block.

FEATURES

•Low noise: 3.5pA/√Hz

•Single 5V supply

•Large bandwidth: 280MHz

•Differential outputs

•Low input/output impedances

•High power supply rejection ratio

•High overload threshold current

•Wide dynamic range

•7kΩ differential transresistance

APPLICATIONS

•Fiber-optic receivers, analog and digital

•Current-to-voltage converters

PIN CONFIGURATION

1

2

3

4

5

6

7 8

14

13

12

11

10

9

GND

2

GND

2

NC

I

IN

NC

V

CC1

V

CC2

GND

1

GND

1

GND

1

GND

1

GND

2

OUT (–)

OUT (+)

D Package

TOP VIEW

SD00318

•Wideband gain block

•Medical and scientific instrumentation

•Sensor preamplifiers

•Single-ended to differential conversion

•Low noise RF amplifiers

•RF signal processing

ORDERING INFORMATION

DESCRIPTION TEMPERATURE RANGE ORDER CODE DWG #

14-Pin Plastic Small Outline (SO) Package 0 to +70°C NE5210D SOT108-1

ABSOLUTE MAXIMUM RATINGS

SYMBOL PARAMETER RATING UNIT

V

CC

Power supply 6 V

T

A

Operating ambient temperature range 0 to +70 °C

T

J

Operating junction temperature range -55 to +150 °C

T

STG

Storage temperature range -65 to +150 °C

P

DMAX

Power dissipation, TA=25°C (still air)

1

1.0 W

I

INMAX

Maximum input current

2

5 mA

NOTES:

1. Maximum dissipation is determined by the operating ambient temperature and the thermal resistance: θ

JA

=125°C/W.

2. The use of a pull-up resistor to VCC for the PIN diode, is recommended.

RECOMMENDED OPERATING CONDITIONS

SYMBOL PARAMETER RATING UNIT

V

CC

Supply voltage 4.5 to 5.5 V

T

A

Ambient temperature range 0 to +70 °C

T

J

Junction temperature range 0 to +90 °C

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1995 Apr 26

2

DC ELECTRICAL CHARACTERISTICS

Min and Max limits apply over operating temperature range at VCC=5V, unless otherwise specified. Typical data applies at VCC=5V and T

A

=25°C.

LIMITS

SYMBOL

PARAMETER

TEST CONDITIONS

Min Typ Max

UNIT

V

IN

Input bias voltage 0.6 0.8 0.95 V

V

O

±

Output bias voltage 2.8 3.3 3.7 V

V

OS

Output offset voltage 0 80 mV

I

CC

Supply current 21 26 32 mA

I

OMAX

Output sink/source current

1

3 4 mA

I

IN

Input current (2% linearity) Test Circuit 8, Procedure 2 ±120 ±160 µA

I

INMAX

Maximum input current overload threshold

Test Circuit 8, Procedure 4 ±160 ±240 µA

NOTES:

1. Test condition: output quiescent voltage variation is less than 100mV for 3mA load current.

AC ELECTRICAL CHARACTERISTICS

Typical data and Min/Max limits apply at VCC=5V and TA=25°C.

LIMITS

SYMBOL

PARAMETER

TEST CONDITIONS

Min Typ Max

UNIT

R

T

Transresistance

(differential output)

DC tested, RL=∞

Test Circuit 8, Procedure 1

4.9 7 10 kΩ

R

O

Output resistance

(differential output)

DC tested 16 30 42 Ω

R

T

Transresistance

(single-ended output)

DC tested, RL=∞ 2.45 3.5 5 kΩ

R

O

Output resistance

(single-ended output)

DC tested 8 15 21 Ω

f

3dB

Bandwidth (-3dB) Test Circuit 1, TA=25°C 200 280 MHz

R

IN

Input resistance 60 Ω

C

IN

Input capacitance 7.5 pF

∆R/∆V

Transresistance power supply sensitivity

VCC=5±0.5V 9.6 20 %/V

∆R/∆T

Transresistance ambient temperature sensitivity

∆TA=T

A MAX-TA MIN

0.05 0.1 %/°C

I

N

RMS noise current spectral density (referred to input)

f=10MHz, TA=25°C

Test Circuit 2

3.5 6 pA/√Hz

TA=25°C

Test Circuit 2

Integrated RMS noise current over the bandwidth (referred to input)

∆f=100MHz 37

the bandwidth (referred to input) C

S

=0

1

∆f=200MHz 56

I

T

∆f=300MHz 71

nA

∆f=100MHz 40 ∆f=200MHz 66

S

=1pF

∆f=300MHz 89

PSRR

Power supply rejection ratio

2

(V

CC1=VCC2

)

DC tested, ∆VCC=0.1V

Equivalent AC test circuit 3

20 36 dB

PSRR

Power supply rejection ratio

2

(V

CC1

)

DC tested, ∆VCC=0.1V

Equivalent AC test circuit 4

20 36 dB

PSRR

Power supply rejection ratio

2

(V

CC2

)

DC tested, ∆VCC=0.1V

Equivalent AC test circuit 5

65 dB

PSRR

Power supply rejection ratio2 (ECL configuration)

f=0.1MHz, Test Circuit 6 23 dB

CS=1pF

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1995 Apr 26

3

AC ELECTRICAL CHARACTERISTICS (Continued)

LIMITS

SYMBOL

PARAMETER

TEST CONDITIONS

Min Typ Max

UNIT

V

OMAX

Maximum output voltage swing differential

RL=∞

Test Circuit 8, Procedure 3

2.4 3.2 V

P-P

V

INMAX

Maximum input amplitude for output duty cycle of 50±5%

3

Test Circuit 7 650 mV

P-P

t

R

Rise time for 50 mV

P-P

output signal

4

Test Circuit 7 0.8 1.2 ns

NOTES:

1. Package parasitic capacitance amounts to about 0.2pF

2. PSRR is output referenced and is circuit board layout dependent at higher frequencies. For best performance use RF filter in V

CC

line.

3. Guaranteed by linearity and overload tests.

4. t

R

defined as 20-80% rise time. It is guaranteed by a -3dB bandwidth test.

TEST CIRCUITS

Test Circuit 2

Test Circuit 1

R

T



V

OUT

V

IN

R  2  S21  R R

T



V

OUT

V

IN

R  4  S21  R

SINGLE-ENDED DIFFERENTIAL

RO Z

O



1  S22 1  S22



 33 RO 2Z

O



1  S22 1  S22



 66

NETWORK ANALYZER

S-PARAMETER TEST SET

PORT 1

PORT 2

5V

33

IN DUT

OUT

50

33

GND

1

GND

2

V

CC1VCC2

ZO = 50

0.1µF

R

L

= 50

R = 1k

0.1µF

SPECTRUM ANALYZER

5V

33

IN DUT

OUT

33

GND

1

GND

2

V

CC1VCC2

0.1µF

RL = 50

0.1µF

AV = 60DB

NC

Z

O

= 50

Z

O

= 50

SD00319

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1995 Apr 26

4

TEST CIRCUITS (Continued)

Test Circuit 4

Test Circuit 3

NETWORK ANALYZER

S-PARAMETER TEST SET

PORT 1 PORT 2

V

CC2

V

CC1

GND

1

GND

2

IN

CURRENT PROBE

1mV/mA

CAL

TEST

TRANSFORMER

NH0300HB

100

33

16

5V

OUT

BAL.

0.1µF

10µF

0.1µF

50 UNBAL.

NETWORK ANALYZER

S-PARAMETER TEST SET

PORT 1 PORT 2

CURRENT PROBE

1mV/mA

CAL

TEST

TRANSFORMER

NH0300HB

100

33

16

5V

OUT

BAL.

50 UNBAL.

V

CC1

V

CC2

IN

0.1µF

10µF

0.1µF

10µF

5V

10µF

GND

1

GND

2

SD00320

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1995 Apr 26

5

TEST CIRCUITS (Continued)

NETWORK ANALYZER

S-PARAMETER TEST SET

PORT 1 PORT 2

CURRENT PROBE

1mV/mA

CAL

TEST

TRANSFORMER

NH0300HB

100

33

16

5V

OUT

BAL.

50 UNBAL.

V

CC2

V

CC1

IN

0.1µF

10µF

0.1µF

10µF

5V

Test Circuit 6

Test Circuit 5

NETWORK ANALYZER

S-PARAMETER TEST SET

PORT 1 PORT 2

CURRENT PROBE

1mV/mA

CAL

TEST

TRANSFORMER

NH0300HB

100

33

16

OUT

BAL.

50 UNBAL.

GND

2

GND

1

V

CC1

V

CC2

IN

0.1µF

10µF

GND

0.1µF

10µF

5.2V

10µF

GND

2

GND

1

SD00321

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1995 Apr 26

6

TEST CIRCUITS (Continued)

Test Circuit 7

OSCILLOSCOPE

33

1k

OUT

GND

2

GND

1

V

CC1VCC2

IN

0.1µF

PULSE GEN.

Measurement done using

differential wave forms

0.1µF

50

Z

O

= 50Ω

A

B

Z

O

= 50Ω

DUT

SD00322

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1995 Apr 26

7

TEST CIRCUITS (Continued)

GND

2

Test Circuit 8

OUT +

OUT –

GND

1

IN

DUT

IIN (µA)

5V

V

OUT

(V)

+

–

Typical Differential Output Voltage

vs Current Input

2.00

1.60

1.20

0.80

0.40

0.00

–0.40

–0.80

–1.20

–1.60

–2.00

–400 –320 –240 –160 –80 0 80 160 240 320 400

DIFFERENTIAL OUTPUT VOLTAGE (V)

CURRENT INPUT (µA)

NE5210 TEST CONDITIONS

Procedure 1

R

T

measured at 60µA

R

T

= (VO1 – VO2)/(+60µA – (–60µA))

Where: V

O1

Measured at IIN = +60µA

V

O2

Measured at IIN = –60µA

Procedure 2

Linearity = 1 – ABS((V

OA

– VOB) / (VO3 – VO4))

Where: VO3 Measured at IIN = +120µA

V

O4

Measured at IIN = –120µA

VOA RT ( 120A)  V

OB

VOB RT ( 120A)  V

OB

Procedure 3

V

OMAX

= VO7 – V

O8

Where: VO7 Measured at IIN = +260µA

V

O8

Measured at IIN = –260µA

Procedure 4

I

IN

Test Pass Conditions:

V

O7

– VO5 > 20mV and V06 – VO5 > 20mV

Where: V

O5

Measured at IIN = +160µA

V

O6

Measured at IIN = –160µA VO7 Measured at IIN = +260µA V

O8

Measured at IIN = –260µA

SD00323

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1995 Apr 26

8

TYPICAL PERFORMANCE CHARACTERISTICS

DIFFERENTIAL OUTPUT VOLTAGE (V) DIFFERENTIAL OUTPUT VOLTAGE (V)

OUTPUT VOLTAGE (V)

OUTPUT BIAS VOLTAGE (V)

AMBIENT TEMPERATURE (°C)

NE5210 Supply Current

vs Temperature

NE5210 Output Bias Voltage

vs Temperature

Output Voltage

vs Input Current

NE5210 Input Bias Voltage

vs Temperature

NE5210 Output Bias Voltage

vs Temperature

Differential Output Voltage

vs Input Current

NE5210 Output Offset Voltage

vs Temperature

NE5210 Differential Output Swing

vs Temperature

Differential Output Voltage

vs Input Current

AMBIENT TEMPERATURE (°C)

–10 10 20 30 40 50 60 70 80

32

30

28

26

24

22

20

18

0

TOTAL SUPPLY CURRENT (mA)

(I + I )

CC1

CC2

3.50

3.46

3.42

3.38

3.34

3.30 –10 10 20 30 40 50 60 70 800

OUTPUT BIAS VOLTAGE (V)

VCC = 5.0V

PIN 14

PIN 12

–10 10 20 30 40 50 60 70 80

900

0

INPUT BIAS VOLTAGE (mV)

850

800

750

700

5.5V

5.0V

4.5V

–10 10 20 30 40 50 60 70 80

20

0

OUTPUT OFFSET VOLTAGE (mV)

0

–20

–40

–60

5.5V

5.0V

4.5V

AMBIENT TEMPERATURE (°C)

–80

VOS = V

OUT12

– V

OUT14

–10 10 20 30 40 50 60 70 800

AMBIENT TEMPERATURE (°C)

4.1

3.9

3.7

3.5

3.3

3.1

2.9

2.7

PIN 14

5.5V

5.0V

4.5V

DIFFERENTIAL OUTPUT SWING (V)

–10 10 20 30 40 50 60 70 800

4.0

3.8

3.6

3.4

3.2

3.0

2.8

2.6

AMBIENT TEMPERATURE (°C)

5.5V

5.0V

4.5V

2.4

2.2

DC TESTED RL = ∞

4.5

3.0

2.5 –300.0 0 +300.0

INPUT CURRENT (µA)

+25°C

+125°C

+85°C

–55°C

+125°C

+85°C

2.0

0

–2.0

–300.0 0 +300.0

INPUT CURRENT (µA)

5.5V

5.0V

4.5V

5.5V

5.0V

4.5V

2.0

0

–2.0

–300.0 0 +300.0

INPUT CURRENT (µA)

+125°C

+85°C

+25°C

–55°C

SD00324

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1995 Apr 26

9

TYPICAL PERFORMANCE CHARACTERISTICS (Continued)

8 7 6 5 4 3 2 1 0

–1

GAIN (dB)

POPULATION (%)

Gain vs Frequency Gain vs Frequency

NE5210 Differential Transresistance

vs Temperature

Gain vs Frequency Gain vs Frequency NE5210 Bandwidth vs Temperature

Gain and Phase Shift

vs Frequency

NE5210 Typical

Bandwidth Distribution

(70 Parts from 4 Wafer Lots)

Gain and Phase Shift

vs Frequency

50

40

30

20

10

0

223 255 287 319 351 383

FREQUENCY (MHz)

PIN 12 SINGLE-ENDED

RL = 50Ω

V

CC

= 5.0V

TA = 25°C

AMBIENT TEMPERATURE (°C)

8.6

–10 10 20 30 40 50 60 70 800

DIFFERENTIAL TRANSRESISTANCE (k )

RL = ∞

8.4

8.2

8.0

7.8

7.6

7.4

Ω

5.5V

5.0V

4.5V

450

–10 10 20 30 40 50 60 70 800

BANDWIDTH (MHz)

AMBIENT TEMPERATURE (°C)

400

350

300

250

200

5.5V

5.0V

4.5V

PIN 12

PIN 12 VCC = 5V TA = 25°C

8 7 6 5 4 3 2 1 0

–1

1 10 100 1000

FREQUENCY (MHz)

GAIN (dB)

180

90

0

–90

–180

PHASE ( )

o

1 10 100 1000

FREQUENCY (MHz)

PIN 12 VCC = 5V

8 7 6 5 4 3 2 1 0

–1

GAIN (dB)

–55°C

+125°C

+85°C

25°C

+125°C

–55°C

1 10 100 1000

FREQUENCY (MHz)

PIN 12 VCC = 5V RL = 50Ω

5.0V

5.5V

4.5V

8 7 6 5 4 3 2 1 0

–1

GAIN (dB)

1 10 100 1000

FREQUENCY (MHz)

PIN 12 VCC = 5V R

L

= 50Ω

5.0V

5.5V

4.5V

1 10 100 1000

FREQUENCY (MHz)

PIN 14

8 7 6 5 4 3 2 1 0

–1

GAIN (dB)

VCC = 5V

–55°C

+125°C

25°C

85°C

PIN 14

8 7 6 5 4 3 2 1 0

–1

1 10 100 1000

FREQUENCY (MHz)

GAIN (dB)

360

270

180

90

0

PHASE ( )

o

VCC = 5V T

A

= 25°C

RL = Ω

SINGLE-ENDED

SD00325

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1995 Apr 26

10

TYPICAL PERFORMANCE CHARACTERISTICS (Continued)

FREQUENCY (MHz)

NE5210 Output Resistance

vs Temperature

NE5210 Output Resistance

vs Temperature

NE5210 Output Resistance

vs Temperature

Output Resistance

vs Frequency

NE5210 Power Supply Rejection Ratio

vs Temperature

Group Delay

Output Step Response

0 2 4 6 8 10 12 14 16 18 20

(ns)

VCC = 5V TA = 25°C

20mV/Div

VCC = 5V TA = 25°C

10

8 6 4 2 0

0.1 20 40 60 80 100 120 140 160 180 200

DELAY (ns)

40

39

38

37

36

35

34

33

–10 0 10 20 30 40 50 60 70 80

POWER SUPPLY REJECTION RATIO (dB)

AMBIENT TEMPERATURE (°C)

V

CC1

= V

CC2

= 5.0V

∆VCC = ±0.1V DC TESTED OUTPUT REFERRED

17

16

15

14

13

–10 0 10 20 30 40 50 60 70 80

AMBIENT TEMPERATURE (°C)

OUTPUT RESISTANCE ( )

Ω

PIN 14 OUTPUT REFERRED

4.5V

5.0V

5.5V

16

15

14

13

12

–10 0 10 20 30 40 50 60 70 80

OUTPUT RESISTANCE ( )

Ω

PIN 12 OUTPUT REFERRED

4.5V

5.0V

5.5V

AMBIENT TEMPERATURE (°C)

17

16

15

14

13

–10 0 10 20 30 40 50 60 70 80

OUTPUT RESISTANCE ( )

Ω

VCC = 5.0V DC TESTED

5.0V

AMBIENT TEMPERATURE (°C)

12

PIN 14 R

OUT

PIN 12 R

OUT

5.0V

80 70 60 50 40 30 20 10

0

OUTPUT RESISTANCE ( )

Ω

FREQUENCY (MHz)

0.1 1 10 100 200

PIN 12

PIN 14

VCC = 5.0V T

A

= 25°C

SD00326

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1995 Apr 26

11

THEORY OF OPERATION

Transimpedance amplifiers have been widely used as the preamplifier in fiber-optic receivers. The NE5210 is a wide bandwidth (typically 280MHz) transimpedance amplifier designed primarily for input currents requiring a large dynamic range, such as those produced by a laser diode. The maximum input current before output stage clipping occurs at typically 240µA. The NE5210 is a bipolar transimpedance amplifier which is current driven at the input and generates a differential voltage signal at the outputs. The forward transfer function is therefore a ratio of the differential output voltage to a given input current with the dimensions of ohms. The main feature of this amplifier is a wideband, low-noise input stage which is desensitized to photodiode capacitance variations. When connected to a photodiode of a few picoFarads, the frequency response will not be degraded significantly. Except for the input stage, the entire signal path is differential to provide improved power-supply rejection and ease of interface to ECL type circuitry. A block diagram of the circuit is shown in Figure 1. The input stage (A1) employs shunt-series feedback to stabilize the current gain of the amplifier. The transresistance of the amplifier from the current source to the emitter of Q

3

is approximately the value of the

feedback resistor, R

F

=3.6kΩ. The gain from the second stage (A2) and emitter followers (A3 and A4) is about two. Therefore, the differential transresistance of the entire amplifier , R

T

is

R

T



V

OUT

(diff)

I

IN

 2RF 2(3.6K)  7.2kW

The single-ended transresistance of the amplifier is typically 3.6kΩ. The simplified schematic in Figure 2 shows how an input current is

converted to a differential output voltage. The amplifier has a single input for current which is referenced to Ground 1. An input current from a laser diode, for example, will be converted into a voltage by the feedback resistor R

F

. The transistor Q1 provides most of the

open loop gain of the circuit, A

VOL

≈70. The emitter follower Q

2

minimizes loading on Q1. The transistor Q4, resistor R7, and V

B1

provide level shifting and interface with the Q15 – Q16 differential pair of the second stage which is biased with an internal reference, V

B2

. The differential outputs are derived from emitter followers Q

11

–

Q

12

which are biased by constant current sources. The collectors of

Q

11

– Q12 are bonded to an external pin, V

CC2

, in order to reduce the feedback to the input stage. The output impedance is about 17Ω single-ended. For ease of performance evaluation, a 33Ω resistor is used in series with each output to match to a 50Ω test system.

INPUT

OUTPUT +

OUTPUT –

A1 A2

A3

A4

R

F

SD00327

Figure 1. NE5210 – Block Diagram

BANDWIDTH CALCULATIONS

The input stage, shown in Figure 3, employs shunt-series feedback to stabilize the current gain of the amplifier. A simplified analysis can determine the performance of the amplifier. The equivalent input capacitance, C

IN

, in

parallel with the source, I

S

, is approximately 7.5pF, assuming that

C

S

=0 where CS is the external source capacitance.

Since the input is driven by a current source the input must have a low input resistance. The input resistance, R

IN

, is the ratio of the

incremental input voltage, V

IN

, to the corresponding input current, I

IN

and can be calculated as:

R

IN



V

IN

I

IN



R

F

1  A

VOL



3.6K 71

 51W

More exact calculations would yield a higher value of 60Ω. Thus C

IN

and RIN will form the dominant pole of the entire amplifier;

f

3dB



1

2p R

INCIN

Assuming typical values for RF = 3.6kΩ, RIN = 60Ω, CIN = 7.5pF

f

3dB



1

2p 7.5pF 60

 354MHz

INPUT

OUT–

OUT+

PHOTODIODE

VB2

+

R

1

R

3

R

12

R

13

R

5

R

4

R

7

R

14

R

15

Q

1

Q

3

Q

2

Q

4

Q

15

Q

16

Q

11

Q

12

GND

2

GND

1

V

CC2

V

CC1

R

2

SD00328

Figure 2. Transimpedance Amplifier

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1995 Apr 26

12

V

CC

V

EQ3

V

IN

I

IN

INPUT

I

F

I

B

Q1

Q2

Q3

R2

R3

R4

R

F

R1

I

C1

SD00329

Figure 3. Shunt-Series Input Stage

The operating point of Q1, Figure 2, has been optimized for the lowest current noise without introducing a second dominant pole in the pass-band. All poles associated with subsequent stages have been kept at sufficiently high enough frequencies to yield an overall single pole response. Although wider bandwidths have been achieved by using a cascode input stage configuration, the present solution has the advantage of a very uniform, highly desensitized frequency response because the Miller effect dominates over the external photodiode and stray capacitances. For example, assuming a source capacitance of 1pF, input stage voltage gain of 70, R

IN

=

60Ω then the total input capacitance, C

IN

= (1+7.5) pF which will

lead to only a 12% bandwidth reduction.

NOISE

Most of the currently installed fiber-optic systems use non-coherent transmission and detect incident optical power. Therefore, receiver noise performance becomes very important. The input stage achieves a low input referred noise current (spectral density) of

3.5pA/√Hz

. The transresistance configuration assures that the external high value bias resistors often required for photodiode biasing will not contribute to the total noise system noise. The equivalent input

RMS

noise current is strongly determined by the

quiescent current of Q

1

, the feedback resistor RF, and the bandwidth; however, it is not dependent upon the internal Miller-capacitance. The measured wideband noise was 66nA

RMS

in

a 200MHz bandwidth.

DYNAMIC RANGE CALCULATIONS

The electrical dynamic range can be defined as the ratio of maximum input current to the peak noise current:

Electrical dynamic range, D

E

, in a 200MHz bandwidth assuming

I

INMAX

= 240µA and a wideband noise of IEQ=66nA

RMS

for an

external source capacitance of C

S

= 1pF.

D

E

+ 20log

(Max. input current) (PK)

(Peak noise current) (RMS)@ 2

Ǹ

+ 20log

(240 @ 10

*6

)

( 2

Ǹ

66 10*9)

+ 68dB

In order to calculate the optical dynamic range the incident optical power must be considered.

For a given wavelength λ; (meters) Energy of one Photon =

hc

l

watt sec (Joule)

Where h=Planck’s Constant = 6.6 × 10

-34

Joule sec.

c = speed of light = 3 × 10

8

m/sec c / λ = optical frequency (Hz) No. of incident photons/sec= where P=optical incident power

No. of incident photons/sec =

P

hs

l

where P = optical incident power

No. of generated electrons/sec =

h @

P

hs

l

where η = quantum efficiency

+

no. of generated electron hole paris

no. of incident photons

N I + h @

P

hs

l

@ e Amps (Coulombsń sec.)

where e = electron charge = 1.6 × 10

-19

Coulombs

Responsivity R =

h @e

hs

l

Amp/watt

I + P @ R

Assuming a data rate of 400 Mbaud (Bandwidth, B=200MHz), the noise parameter Z may be calculated as:

1

Z +

I

EQ

qB

+

66 @ 10

*9

(1.6@ 10

*19

)(200 @ 106)

+ 2063

where Z is the ratio of

RMS

noise output to the peak response to a single hole-electron pair. Assuming 100% photodetector quantum efficiency, half mark/half space digital transmission, 850nm lightwave and using Gaussian approximation, the minimum required optical power to achieve 10

-9

BER is:

P

avMIN

+ 12

hc

l

B Z + 12 2.3@ 10

*19

200@ 1062063

+ 1139nW + * 29.4dBm

where h is Planck’s Constant, c is the speed of light, λ is the wavelength. The minimum input current to the NE5210, at this input power is:

I

avMIN

+ qP

avMIN

l

hc

+

1139@ 10

*9

@ 1.6 @ 10

*19

2.3@ 10

*19

= 792nA Choosing the maximum peak overload current of I

avMAX

=240µA, the

maximum mean optical power is:

P

avMAX

+

hcI

avMAX

l q

+

2.3@ 10

*19

1.6@ 10

*19

240 @ 10

*6

Thus the optical dynamic range, DO is: D

O

= P

avMAX

- P

avMIN

= -4.6 -(-29.4) = 24.8dB.

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1995 Apr 26

13

This represents the maximum limit attainable with the NE5210 operating at 200MHz bandwidth, with a half mark/half space digital transmission at 850nm wavelength.

APPLICATION INFORMATION

Package parasitics, particularly ground lead inductances and parasitic capacitances, can significantly degrade the frequency response. Since the NE5210 has differential outputs which can feed back signals to the input by parasitic package or board layout capacitances, both peaking and attenuating type frequency response shaping is possible. Constructing the board layout so that Ground 1 and Ground 2 have very low impedance paths has produced the best results. This was accomplished by adding a ground-plane stripe underneath the device connecting Ground 1, Pins 8–11, and Ground 2, Pins 1 and 2 on opposite ends of the SO14 package. This ground-plane stripe also provides isolation between the output return currents flowing to either V

CC2

or Ground 2 and the input photodiode currents to flowing to Ground 1. Without this ground-plane stripe and with large lead inductances on the board, the part may be unstable and oscillate near 800MHz. The easiest way to realize that the part is not functioning normally is to measure the DC voltages at the outputs. If they are not close to their

quiescent values of 3.3V (for a 5V supply), then the circuit may be oscillating. Input pin layout necessitates that the photodiode be physically very close to the input and Ground 1. Connecting Pins 3 and 5 to Ground 1 will tend to shield the input but it will also tend to increase the capacitance on the input and slightly reduce the bandwidth.

As with any high-frequency device, some precautions must be observed in order to enjoy reliable performance. The first of these is the use of a well-regulated power supply. The supply must be capable of providing varying amounts of current without significantly changing the voltage level. Proper supply bypassing requires that a good quality 0.1µF high-frequency capacitor be inserted between V

CC1

and V

CC2

, preferably a chip capacitor, as close to the package pins as possible. Also, the parallel combination of 0.1µF capacitors with 10µF tantalum capacitors from each supply, V

CC1

and V

CC2

, to the ground plane should provide adequate decoupling. Some applications may require an RF choke in series with the power supply line. Separate analog and digital ground leads must be maintained and printed circuit board ground plane should be employed whenever possible.

Figure 4 depicts a 50Mb/s TTL fiber-optic receiver using the BPF31, 850nm LED, the NE5210 and the NE5214 post amplifier.

1

2

3

4

5

6

7

8

9

10 11

12

13

14

15

16

17

18

19

20

1

2

3

4

5

6

78

9

10

11

12

13

14

LED

C

PKDET

THRESH

GND

A

FLAG

JAM

V

CCD

V

CCA

GND

D

TTL

OUT

IN

1B

IN

1A

C

AZP

C

AZN

OUT

1B

IN

8B

OUT

1A

IN

8A

R

HYST

R

PKDET

NE5214

GND

OUT

GND

OUT

V

CC

V

CC

NC

I

IN

NC

GND

NE5210

R2

220

D1

LED

C9

100pF

C7

.01µF

47µF

C1

C2

GND

+V

CC

0.1µF

R4 4k

R3 47k

V

OUT

(TTL)

L3

10µH

L2

10µH

C11

C10

.01µF

C13

C12

10µF

10

µF

C8

L1

10µH

BPF31

OPTICAL

INPUT

R1 100

C5

1.0µF

C6

.01µF

C4

10µF

C3

NOTE: The NE5210/NE5217 combination can operate at data rates in excess of 100Mb/s NRZ The capacitor C7 decreases the NE5210 bandwidth to improve overall S/N ratio in the DC–50MHz band, but does create extra high frequency noise

on the NE5210 VCC pin(s).

SD00330

Figure 4. A 50Mb/s Fiber Optic Receiver

Philips Semiconductors Product specification

NE5210Transimpedance amplifier (280MHz)

1995 Apr 26

14

GND 2

NC

INPUT

NC

GND 1

OUT (+)

GND 2

OUT (–)

1

2

3

4

5

6

7 8

9

10

11

12

13

14

VCC1

VCC 2

SD00488

ECN No.: 06027 1992 Mar 13

Figure 5. NE5210 Bonding Diagram

Die Sales Disclaimer

Due to the limitations in testing high frequency and other parameters at the die level, and the fact that die electrical characteristics may shift after packaging, die electrical parameters are not specified and die are not guaranteed to meet electrical characteristics (including temperature range) as noted in this data sheet which is intended only to specify electrical characteristics for a packaged device.

All die are 100% functional with various parametrics tested at the wafer level, at room temperature only (25°C), and are guaranteed to be 100% functional as a result of electrical testing to the point of wafer sawing only. Although the most modern processes are utilized for wafer sawing and die pick and place into waffle pack

carriers, it is impossible to guarantee 100% functionality through this process. There is no post waffle pack testing performed on individual die.

Since Philips Semiconductors has no control of third party procedures in the handling or packaging of die, Philips Semiconductors assumes no liability for device functionality or performance of the die or systems on any die sales.

Although Philips Semiconductors typically realizes a yield of 85% after assembling die into their respective packages, with care customers should achieve a similar yield. However, for the reasons stated above, Philips Semiconductors cannot guarantee this or any other yield on any die sales.