Page 1
November 2002
LM2733
0.6/1.6 MHz Boost Converters With 40V Internal FET
Switch in SOT-23
LM2733 0.6/1.6 MHz Boost Converters With 40V Internal FET Switch in SOT-23
General Description
The LM2733 switching regulators are current-mode boost
converters operating fixed frequency of 1.6 MHz (“X” option)
and 600 kHz (“Y” option).
The SOT-23 package, made possible by the minimal power
loss of the internal 1A switch, and use of small inductors and
capacitors result in the industry’s highest power density. The
40V internal switch makes these solutions perfect for boosting to voltages of 16V or greater.
These parts have a logic-level shutdown pin that can be
used to reduce quiescent current and extend battery life.
Protection is provided through cycle-by-cycle current limiting
and thermal shutdown. Internal compensation simplifies design and reduces component count.
Switch Frequency
XY
1.6 MHz 0.6 MHz
Typical Application Circuit
Features
n 40V DMOS FET switch
n 1.6 MHz (“X”), 0.6 MHz (“Y”) switching frequency
n Low R
n Switch current up to 1A
n Wide input voltage range (2.7V–14V)
n Low shutdown current (
n 5-Lead SOT-23 package
n Uses tiny capacitors and inductors
n Cycle-by-cycle current limiting
n Internally compensated
(ON) DMOS FET
DS
<
1 µA)
Applications
n White LED Current Source
n PDA’s and Palm-Top Computers
n Digital Cameras
n Portable Phones and Games
n Local Boost Regulator
20055424
20055401
© 2002 National Semiconductor Corporation DS200554 www.national.com
Page 2
Typical Application Circuit (Continued)
LM2733
Connection Diagram
20055440
Top View
5-Lead SOT-23 Package
See NS Package Number MF05A
20055402
Ordering Information
Order Number Package Type Package Drawing Supplied As Package ID
LM2733XMF
LM2733XMFX 3K Tape and Reel S52A
LM2733YMF 1K Tape and Reel S52B
LM2733YMFX 3K Tape and Reel S52B
SOT23-5 MF05A
1K Tape and Reel S52A
Pin Description
Pin Name Function
1 SW Drain of the internal FET switch.
2 GND Analog and power ground.
3 FB Feedback point that connects to external resistive divider.
4 SHDN
5V
IN
Shutdown control input. Connect to VINif this feature is not used.
Analog and power input.
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Page 3
Block Diagram
LM2733
20055403
Theory of Operation
The LM2733 is a switching converter IC that operates at a
fixed frequency (0.6 or 1.6 MHz) using current-mode control
for fast transient response over a wide input voltage range
and pulse-by-pulse current limiting. Because this is current
mode control, a 50 mΩ sense resistor in series with the
switch FET is used to provide a voltage (which is proportional to the FET current) to both the input of the pulse width
modulation (PWM) comparator and the current limit amplifier.
At the beginning of each cycle, the S-R latch turns on the
FET. As the current through the FET increases, a voltage
(proportional to this current) is summed with the ramp coming from the ramp generator and then fed into the input of the
PWM comparator. When this voltage exceeds the voltage on
the other input (coming from the Gm amplifier), the latch
resets and turns the FET off. Since the signal coming from
the Gm amplifier is derived from the feedback (which
samples the voltage at the output), the action of the PWM
comparator constantly sets the correct peak current through
the FET to keep the output volatge in regulation.
Q1 and Q2 along with R3 - R6 form a bandgap voltage
reference used by the IC to hold the output in regulation. The
currents flowing through Q1 and Q2 will be equal, and the
feedback loop will adjust the regulated output to maintain
this. Because of this, the regulated output is always maintained at a voltage level equal to the voltage at the FB node
"multiplied up" by the ratio of the output resistive divider.
The current limit comparator feeds directly into the flip-flop,
that drives the switch FET. If the FET current reaches the
limit threshold, the FET is turned off and the cycle terminated
until the next clock pulse. The current limit input terminates
the pulse regardless of the status of the output of the PWM
comparator.
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Page 4
Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
LM2733
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Storage Temperature Range −65˚C to +150˚C
Operating Junction
Temperature Range −40˚C to +125˚C
Lead Temp. (Soldering, 5 sec.) 300˚C
SW Pin Voltage −0.4V to +40V
Input Supply Voltage −0.4V to +14.5V
Shutdown Input Voltage
(Survival) −0.4V to +14.5V
θ
(SOT23-5) 265˚C/W
J-A
ESD Rating (Note 3)
Human Body Model
Machine Model
Power Dissipation (Note 2) Internally Limited
FB Pin Voltage −0.4V to +6V
Electrical Characteristics
Limits in standard typeface are for TJ= 25˚C, and limits in boldface type apply over the full operating temperature range
(−40˚C ≤ T
≤ +125˚C). Unless otherwise specified: VIN= 5V, V
J
Symbol Parameter Conditions
V
IN
I
SW
R
(ON) Switch ON Resistance ISW= 100 mA 500 650 mΩ
DS
SHDN
Input Voltage 2.7 14 V
Switch Current Limit (Note 6) 1.0 1.5 A
Shutdown Threshold Device ON 1.5
TH
Device OFF 0.50
I
SHDN
V
FB
Shutdown Pin Bias Current V
Feedback Pin Reference
=0 0
SHDN
V
=5V 0 2
SHDN
VIN=3V
Voltage
I
FB
I
Q
Feedback Pin Bias Current VFB= 1.23V 60 nA
Quiescent Current V
= 5V, Switching "X" 2.1 3.0
SHDN
V
= 5V, Switching "Y" 1.1 2
SHDN
V
= 5V, Not Switching 400 500
SHDN
V
= 0 0.024 1
SHDN
FB Voltage Line Regulation 2.7V ≤ VIN≤ 14V
= 5V, IL= 0A.
SHDN
Min
(Note 4)
Typical
(Note 5)
Max
(Note 4)
1.205 1.230 1.255 V
0.02 %/V
2kV
200V
Units
V
µA
mA
µA
F
SW
Switching Frequency “X” Option 1.15 1.6 1.85
“Y” Option 0.40 0.60 0.8
D
MAX
Maximum Duty Cycle “X” Option 87 93
“Y” Option 93 96
I
L
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the
device outside of the limits set forth under the operating ratings which specify the intended range of operating conditions.
Note 2: The maximum power dissipation which can be safely dissipated for any application is a function of the maximum junction temperature, T
the junction-to-ambient thermal resistance for the SOT-23 package, θ
at any ambient temperature for designs using this device can be calculated using the formula:
If power dissipation exceeds the maximum specified above, the internal thermal protection circuitry will protect the device by reducing the output voltage as required
to maintain a safe junction temperature.
Note 3: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. The machine model is a 200 pF capacitor discharged
directly into each pin.
Note 4: Limits are guaranteed by testing, statistical correlation, or design.
Note 5: Typical values are derived from the mean value of a large quantity of samples tested during characterization and represent the most likely expected value
of the parameter at room temperature.
Note 6: Switch current limit is dependent on duty cycle (see Typical Perrformance Characteristics). Limits shown are for duty cycles ≤ 50%.
www.national.com 4
Switch Leakage Not Switching VSW=5V 1 µA
(MAX) = 125˚C,
= 265˚C/W, and the ambient temperature, TA. The maximum allowable power dissipation
J-A
J
MHz
%
Page 5
LM2733
Typical Performance Characteristics Unless otherwise specified: T
2.2 µF, SHDN pin is tied to V
Feedback Voltage vs Temperature R
= 5V, LM2733X,L=10µH.
IN,VIN
20055406
DS
= 25˚C, C
A
(ON) vs Temperature
LM2733X Oscillator Frequency vs Temperature I Limit vs Temperature
= 4.7 µF, CIN=
OUT
20055407
20055408 20055409
LM2733X Iq vs Temperature LM2733X Efficiency vs Load Current (V
20055410
OUT
20055414
= 12V)
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Page 6
Typical Performance Characteristics Unless otherwise specified: T
2.2 µF, SHDN pin is tied to V
LM2733
LM2733X R
= 5V, LM2733X,L=10µH. (Continued)
IN,VIN
(ON) vs V
DS
IN
LM2733Y Efficiency vs Load (V
= 25˚C, C
A
= 4.7 µF, CIN=
OUT
= 15V)
OUT
LM2733Y Efficiency vs Load (V
LM2733Y Efficiency vs Load (V
20055423
= 20V) LM2733Y Efficiency vs Load (V
OUT
20055427 20055428
= 30V) LM2733Y Efficiency vs Load (V
OUT
OUT
OUT
20055435
= 25V)
= 35V)
20055429 20055430
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Page 7
LM2733
Typical Performance Characteristics Unless otherwise specified: T
2.2 µF, SHDN pin is tied to V
LM2733Y Efficiency vs Load (V
LM2733Y Frequency vs Temperature
= 5V, LM2733X,L=10µH. (Continued)
IN,VIN
= 40V) LM2733Y Iq (Active) vs Temperature
OUT
20055432
= 25˚C, C
A
= 4.7 µF, CIN=
OUT
20055442
20055443
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Page 8
Application Hints
LM2733
SELECTING THE EXTERNAL CAPACITORS
The best capacitors for use with the LM2733 are multi-layer
ceramic capacitors. They have the lowest ESR (equivalent
series resistance) and highest resonance frequency which
makes them optimum for use with high frequency switching
converters.
When selecting a ceramic capacitor, only X5R and X7R
dielectric types should be used. Other types such as Z5U
and Y5F have such severe loss of capacitance due to effects
of temperature variation and applied voltage, they may provide as little as 20% of rated capacitance in many typical
applications. Always consult capacitor manufacturer’s data
curves before selecting a capacitor. High-quality ceramic
capacitors can be obtained from Taiyo-Yuden, AVX, and
Murata.
SELECTING THE OUTPUT CAPACITOR
A single ceramic capacitor of value 4.7 µF to 10 µF will
provide sufficient output capacitance for most applications.
For output voltages below 10V, a 10 µF capacitance is
required. If larger amounts of capacitance are desired for
improved line support and transient response, Tantalum capacitors can be used in parallel with the ceramics. Aluminum
electrolytics with ultra low ESR such as Sanyo Oscon can be
used, but are usually prohibitively expensive. Typical AI electrolytic capacitors are not suitable for switching frequencies
above 500 kHz due to significant ringing and temperature
rise due to self-heating from ripple current. An output capacitor with excessive ESR can also reduce phase margin and
cause instability.
SELECTING THE INPUT CAPACITOR
An input capacitor is required to serve as an energy reservoir
for the current which must flow into the coil each time the
switch turns ON. This capacitor must have extremely low
ESR, so ceramic is the best choice. We recommend a
nominal value of 2.2 µF, but larger values can be used. Since
this capacitor reduces the amount of voltage ripple seen at
the input pin, it also reduces the amount of EMI passed back
along that line to other circuitry.
FEED-FORWARD COMPENSATION
Although internally compensated, the feed-forward capacitor
Cf is required for stability (see Basic Application Circuit).
Adding this capacitor puts a zero in the loop response of the
converter. Without it, the regulator loop can oscillate. The
recommended frequency for the zero fz should be approximately 8 kHz. Cf can be calculated using the formula:
Cf=1/(2Xπ XR1Xfz)
SELECTING DIODES
The external diode used in the typical application should be
a Schottky diode. If the switch voltage is less than 15V, a
20V diode such as the MBR0520 is recommended. If the
switch voltage is between 15V and 25V, a 30V diode such as
the MBR0530 is recommended. If the switch voltage exceeds 25V, a 40V diode such as the MBR0540 should be
used.
The MBR05XX series of diodes are designed to handle a
maximum average current of 0.5A. For applications exceeding 0.5A average but less than 1A, a Microsemi UPS5817
can be used.
LAYOUT HINTS
High frequency switching regulators require very careful layout of components in order to get stable operation and low
noise. All components must be as close as possible to the
LM2733 device. It is recommended that a 4-layer PCB be
used so that internal ground planes are available.
As an example, a recommended layout of components is
shown:
Recommended PCB Component Layout
20055422
Some additional guidelines to be observed:
1. Keep the path between L1, D1, and C2 extremely short.
Parasitic trace inductance in series with D1 and C2 will
increase noise and ringing.
2. The feedback components R1, R2 and CF must be kept
close to the FB pin of U1 to prevent noise injection on
the FB pin trace.
3. If internal ground planes are available (recommended)
use vias to connect directly to ground at pin 2 of U1, as
well as the negative sides of capacitors C1 and C2.
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the external resistors R1 and
R2 (see Basic Application Circuit). A value of approximately
12.1 kΩ is recommended for R2 to establish a divider current
of approximately 100 µA. R1 is calculated using the formula:
R1=R2X(V
/1.23 − 1)
OUT
SWITCHING FREQUENCY
The LM2733 is provided with two switching frequencies: the
“X” version is typically 1.6 MHz, while the “Y” version is
typically 600 kHz. The best frequency for a specific application must be determined based on the tradeoffs involved:
Higher switching frequency means the inductors and capacitors can be made smaller and cheaper for a given output
voltage and current. The down side is that efficiency is
slightly lower because the fixed switching losses occur more
frequently and become a larger percentage of total power
loss. EMI is typically worse at higher switching frequencies
because more EMI energy will be seen in the higher frequency spectrum where most circuits are more sensitive to
such interference.
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Page 9
Application Hints (Continued)
LM2733
Basic Application Circuit
DUTY CYCLE
The maximum duty cycle of the switching regulator determines the maximum boost ratio of output-to-input voltage
that the converter can attain in continuous mode of operation. The duty cycle for a given boost application is defined
as:
This applies for continuous mode operation.
The equation shown for calculating duty cycle incorporates
terms for the FET switch voltage and diode forward voltage.
The actual duty cycle measured in operation will also be
affected slightly by other power losses in the circuit such as
wire losses in the inductor, switching losses, and capacitor
ripple current losses from self-heating. Therefore, the actual
(effective) duty cycle measured may be slightly higher than
calculated to compensate for these power losses. A good
approximation for effctive duty cycle is :
DC (eff) = (1 - Efficiency x (V
IN/VOUT
))
Where the efficiency can be approximated from the curves
provided.
20055405
the inductor current to drop to zero during the cycle. It should
be noted that all boost converters shift over to discontinuous
operation as the output load is reduced far enough, but a
larger inductor stays “continuous” over a wider load current
range.
To better understand these tradeoffs, a typical application
circuit (5V to 12V boost with a 10 µH inductor) will be
analyzed. We will assume:
V
IN
=5V,V
OUT
= 12V, V
= 0.5V, VSW= 0.5V
DIODE
Since the frequency is 1.6 MHz (nominal), the period is
approximately 0.625 µs. The duty cycle will be 62.5%, which
means the ON time of the switch is 0.390 µs. It should be
noted that when the switch is ON, the voltage across the
inductor is approximately 4.5V.
Using the equation:
V = L (di/dt)
We can then calculate the di/dt rate of the inductor which is
found to be 0.45 A/µs during the ON time. Using these facts,
we can then show what the inductor current will look like
during operation:
INDUCTANCE VALUE
The first question we are usually asked is: “How small can I
make the inductor?” (because they are the largest sized
component and usually the most costly). The answer is not
simple and involves tradeoffs in performance. Larger inductors mean less inductor ripple current, which typically means
less output voltage ripple (for a given size of output capacitor). Larger inductors also mean more load power can be
delivered because the energy stored during each switching
cycle is:
E =L/2 X (lp)
2
Where “lp” is the peak inductor current. An important point to
observe is that the LM2733 will limit its switch current based
on peak current. This means that since lp(max) is fixed,
increasing L will increase the maximum amount of power
available to the load. Conversely, using too little inductance
may limit the amount of load current which can be drawn
from the output.
Best performance is usually obtained when the converter is
operated in “continuous” mode at the load current range of
interest, typically giving better load regulation and less output ripple. Continuous operation is defined as not allowing
10 µH Inductor Current,
20055412
5V–12V Boost (LM2733X)
During the 0.390 µs ON time, the inductor current ramps up
0.176A and ramps down an equal amount during the OFF
time. This is defined as the inductor “ripple current”. It can
also be seen that if the load current drops to about 33 mA,
the inductor current will begin touching the zero axis which
means it will be in discontinuous mode. A similar analysis
can be performed on any boost converter, to make sure the
ripple current is reasonable and continuous operation will be
maintained at the typical load current values.
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Page 10
Application Hints (Continued)
MAXIMUM SWITCH CURRENT
LM2733
The maximum FET swtch current available before the current limiter cuts in is dependent on duty cycle of the application. This is illustrated in the graphs below which show
both the typical and guaranteed values of switch current for
both the "X" and "Y" versions as a function of effective
(actual) duty cycle:
The equation shown to calculate maximum load current
takes into account the losses in the inductor or turn-OFF
switching losses of the FET and diode. For actual load
current in typical applications, we took bench data for various input and output voltages for both the "X" and "Y"
versions of the LM2733 and displayed the maximum load
current available for a typical device in graph form:
LM2733x Switch Current Limit vs Duty Cycle
20055425
LM2733y Switch Current Limit vs Duty Cycle
20055426
CALCULATING LOAD CURRENT
As shown in the figure which depicts inductor current, the
load current is related to the average inductor current by the
relation:
I
LOAD=IIND
(AVG) x (1 - DC)
Where "DC" is the duty cycle of the application. The switch
current can be found by:
I
SW=IIND
(AVG) +1⁄2(I
RIPPLE
)
Inductor ripple current is dependent on inductance, duty
cycle, input voltage and frequency:
=DCx(VIN-VSW)/(fxL)
I
RIPPLE
combining all terms, we can develop an expression which
allows the maximum available load current to be calculated:
LM2733x Max. Load Current vs V
LM2733y Max. Load Current vs V
DESIGN PARAMETERS VSWAND I
SW
20055434
IN
20055433
IN
The value of the FET "ON" voltage (referred to as VSWin the
equations) is dependent on load current. A good approximation can be obtained by multiplying the "ON Resistance" of
the FET times the average inductor current.
FET on resistance increases at V
values below 5V, since
IN
the internal N-FET has less gate voltage in this input voltage
range (see Typical performance Characteristics curves).
Above V
= 5V, the FET gate voltage is internally clamped
IN
to 5V.
The maximum peak switch current the device can deliver is
dependent on duty cycle. The minimum value is guaranteed
>
to be
1A at duty cycle below 50%. For higher duty cycles,
see Typical performance Characteristics curves.
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LM2733
Application Hints (Continued)
THERMAL CONSIDERATIONS
At higher duty cycles, the increased ON time of the FET
means the maximum output current will be determined by
power dissipation within the LM2733 FET switch. The switch
power dissipation from ON-state conduction is calculated by:
=DCxI
P
(SW)
There will be some switching losses as well, so some derating needs to be applied when calculating IC power dissipation.
MINIMUM INDUCTANCE
In some applications where the maximum load current is
relatively small, it may be advantageous to use the smallest
possible inductance value for cost and size savings. The
converter will operate in discontinuous mode in such a case.
The minimum inductance should be selected such that the
inductor (switch) current peak on each cycle does not reach
the 1A current limit maximum. To understand how to do this,
an example will be presented.
In the example, the LM2733X will be used (nominal switching frequency 1.6 MHz, minimum switching frequency
1.15 MHz). This means the maximum cycle period is the
reciprocal of the minimum frequency:
T
ON(max)
We will assume the input voltage is 5V, V
0.2V, V
= 0.3V. The duty cycle is:
DIODE
Duty Cycle = 60.3%
Therefore, the maximum switch ON time is 0.524 µs. An
inductor should be selected with enough inductance to prevent the switch current from reaching 1A in the 0.524 µs ON
time interval (see below):
(AVE)2xRDSON
IND
= 1/1.15M = 0.870 µs
OUT
= 12V, VSW=
The voltage across the inductor during ON time is 4.8V.
Minimum inductance value is found by:
V = L X dl/dt, L = V X (dt/dl) = 4.8 (0.524µ/1) = 2.5 µH
In this case, a 2.7 µH inductor could be used assuming it
provided at least that much inductance up to the 1A current
value. This same analysis can be used to find the minimum
inductance for any boost application. Using the slower
switching “Y” version requires a higher amount of minimum
inductance because of the longer switching period.
INDUCTOR SUPPLIERS
Some of the recommended suppliers of inductors for this
product are Sumida, Coilcraft, Panasonic, and Murata.
When selecting an inductor, make certain that the continuous current rating is high enough to avoid saturation at peak
currents. A suitable core type must be used to minimize core
(switching) losses, and wire power losses must be considered when selecting the current rating.
SHUTDOWN PIN OPERATION
The device is turned off by pulling the shutdown pin low. If
this function is not going to be used, the pin should be tied
directly to V
resistor must be used to V
. If the SHDN function will be needed, a pull-up
IN
(approximately 50k-100kΩ rec-
IN
ommended). The SHDN pin must not be left unterminated.
Discontinuous Design, 5V–12V Boost (LM2733X)
20055413
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Page 12
Physical Dimensions inches (millimeters) unless otherwise noted
5-Lead SOT-23 Package
Order Number LM2733XMF, LM2733XMFX, LM2733YMF or LM2733YMFX
NS Package Number MF05A
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DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL
COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:
LM2733 0.6/1.6 MHz Boost Converters With 40V Internal FET Switch in SOT-23
1. Life support devices or systems are devices or
systems which, (a) are intended for surgical implant
into the body, or (b) support or sustain life, and
whose failure to perform when properly used in
accordance with instructions for use provided in the
2. A critical component is any component of a life
support device or system whose failure to perform
can be reasonably expected to cause the failure of
the life support device or system, or to affect its
safety or effectiveness.
labeling, can be reasonably expected to result in a
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