Datasheet LM27313 Datasheet (National Semiconductor)

July 2007

LM27313

1.6 MHz Boost Converter With 30V Internal FET Switch in
SOT-23

LM27313 1.6 MHz Boost Converter With 30V Internal FET Switch in SOT-23

General Description

The LM27313 switching regulator is a current-mode boost
converter with a fixed operating frequency of 1.6 MHz.

The use of the SOT-23 package, made possible by the mini­mal losses of the 800 mA switch, and small inductors and
capacitors result in extremely high power density. The 30V
internal switch makes these solutions perfect for boosting to
voltages of 5V to 28V.

This part has a logic-level shutdown pin that can be used to
reduce quiescent current and extend battery life.

Protection is provided through cycle-by-cycle current limiting
and thermal shutdown. Internal compensation simplifies de­sign and reduces component count.

Typical Application Circuits

Features

30V DMOS FET switch

■

1.6 MHz switching frequency

■

Low RDS(ON) DMOS FET

■

Switch current up to 800 mA

■

Wide input voltage range (2.7V–14V)

■

Low shutdown current (<1 µA)

■

5-Lead SOT-23 package

■

Uses tiny capacitors and inductors

■

Cycle-by-cycle current limiting

■

Internally compensated

■

Applications

White LED Current Source

■

PDA’s and Palm-Top Computers

■

Digital Cameras

■

Portable Phones, Games and Media Players

■

GPS Devices

■

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20216857

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20216858

© 2007 National Semiconductor Corporation 202168 www.national.com

Connection Diagram

LM27313

Ordering Information

Order

Number

LM27313XMF

LM27313XMFX 3K Tape and Reel SRPB

Pin Descriptions

Pin Name Function

1 SW Drain of the internal FET switch.

2 GND Analog and power ground.

3 FB Feedback point that connects to external resistive divider to set V

4 SHDN Shutdown control input. Connect to VIN if this feature is not used.

5 V

IN

See NS Package Number MF05A

Package

Type

SOT23-5 MF05A

Analog and power input.

Top View

5-Lead SOT-23 Package

Package
Drawing

20216802

Supplied

As

1K Tape and Reel SRPB

.

OUT

Package

Marking

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LM27313

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.

Storage Temperature Range −65°C to +150°C

Lead Temp. (Soldering, 5 sec.) 300°C

Power Dissipation (Note 2) Internally Limited

FB Pin Voltage −0.4V to +6V

SW Pin Voltage −0.4V to +30V

Input Supply Voltage −0.4V to +14.5V

Shutdown Input Voltage
(Survival) −0.4V to +14.5V

ESD Rating (Note 3)
Human Body Model

±2 kV

Operating Ratings

V

IN

V

SW(MAX)

V

SHDN

Junction Temperature, T
(Note 2)

θ

(SOT23-5)

J-A

J

2.7V to 14V

0V to V

-40°C to 125°C

265°C/W

Electrical Characteristics

Unless otherwise specified: VIN = 5V, V
limits in boldface type apply over the full operating temperature range (−40°C ≤ TJ ≤ +125°C). Minimum and Maximum limits are

guaranteed through test, design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C,
and are provided for reference purposes only.

Symbol Parameter Conditions Min Typical Max Units

V

I

SW

R

DS(ON)

V

SHDN(TH)

I

SHDN

V

I

FB

I

ΔVFB/ΔV

f

SW

D

MAX

I

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
to be functional, but does not guarantee specific limits. For guaranteed specifications and conditions see the Electrical Characteristic table.

Note 2: The maximum power dissipation which can be safely dissipated for any application is a function of the maximum junction temperature, T
the junction-to-ambient thermal resistance for the SOT-23 package, θ
at any ambient temperature for designs using this device can be calculated using the formula:

Input Voltage 2.7 14 V

IN

Switch Current Limit (Note 4) 0.80 1.25 A

Switch ON Resistance

Shutdown Threshold

Shutdown Pin Bias Current

Feedback Pin Reference

FB

Voltage

Feedback Pin Bias Current

Quiescent Current

Q

FB Voltage Line Regulation

IN

Switching Frequency 1.15 1.6 1.90 MHz

Maximum Duty Cycle 80 88 %

Switch Leakage

L

= 5V, IL = 0 mA, and TJ = 25°C. Limits in standard typeface are for TJ = 25°C, and

SHDN

ISW = 100 mA

500 650

Device ON 1.5

Device OFF 0.50

V

= 0

SHDN

V

= 5V

SHDN

VIN = 3V

VFB = 1.23V

V

= 5V, Switching

SHDN

V

= 5V, Not Switching

SHDN

V

= 0

SHDN

2.7V ≤ VIN ≤ 14V

Not Switching, VSW = 5V

= 265°C/W, and the ambient temperature, TA. The maximum allowable power dissipation

J-A

0

0 2

1.205 1.230 1.255 V

60 nA

2.1 3.0 mA

400 500

0.024 1

0.02 %/V

1 µA

J(MAX)

mΩ

µA

µA

= 125°C,

30V

IN

V

If power dissipation exceeds the maximum specified above, the internal thermal protection circuitry will protect the device by reducing the output voltage as
required to maintain a safe junction temperature.

Note 3: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. Test method is per JESD22-A114.
Note 4: Switch current limit is dependent on duty cycle. Limits shown are for duty cycles ≤ 50%. See Figure 3 in Application Information – MAXIMUM SWITCH

CURRENT section.

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Typical Performance Characteristics Unless otherwise specified: V

TJ = 25°C.

LM27313

Iq VIN (Active) vs Temperature

Oscillator Frequency vs Temperature

= 5V, SHDN pin is tied to VIN,

IN

Max. Duty Cycle vs Temperature

RDS(ON) vs Temperature

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Feedback Voltage vs Temperature

20216806

Current Limit vs Temperature

20216807

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20216809

R

DS(ON)

vs V

IN

Efficiency vs Load Current (V

OUT

= 12V)

Efficiency vs Load Current (V

Efficiency vs Load Current (V

OUT

OUT

20216823

= 15V)

20216845

= 25V)

Efficiency vs Load Current (V

OUT

20216814

= 20V)

20216846

20216847

5

Block Diagram

LM27313

Theory of Operation

The LM27313 is a switching converter IC that operates at a
fixed frequency of 1.6 MHz using current-mode control for fast
transient response over a wide input voltage range and in­corporate pulse-by-pulse current limiting protection. Because
this is current mode control, a 50 mΩ sense resistor in series
with the switch FET is used to provide a voltage (which is
proportional to the FET current) to both the input of the pulse
width modulation (PWM) comparator and the current limit
amplifier.

At the beginning of each cycle, the S-R latch turns on the FET.
As the current through the FET increases, a voltage (propor­tional to this current) is summed with the ramp coming from
the ramp generator and then fed into the input of the PWM
comparator. When this voltage exceeds the voltage on the
other input (coming from the Gm amplifier), the latch resets
and turns the FET off. Since the signal coming from the Gm
amplifier is derived from the feedback (which samples the
voltage at the output), the action of the PWM comparator
constantly sets the correct peak current through the FET to
keep the output voltage in regulation.

Q1 and Q2 along with R3 - R6 form a bandgap voltage refer­ence used by the IC to hold the output in regulation. The
currents flowing through Q1 and Q2 will be equal, and the
feedback loop will adjust the regulated output to maintain this.
Because of this, the regulated output is always maintained at
a voltage level equal to the voltage at the FB node "multiplied
up" by the ratio of the output resistive divider.

The current limit comparator feeds directly into the flip-flop,
that drives the switch FET. If the FET current reaches the limit
threshold, the FET is turned off and the cycle terminated until
the next clock pulse. The current limit input terminates the
pulse regardless of the status of the output of the PWM com­parator.

20216803

Application Information

SELECTING THE EXTERNAL CAPACITORS

The LM27313 requires ceramic capacitors at the input and
output to accommodate the peak switching currents the part
needs to operate. Electrolytic capacitors have resonant fre­quencies which are below the switching frequency of the
device, and therefore can not provide the currents needed to
operate. Electrolytics may be used in parallel with the ceram­ics for bulk charge storage which will improve transient re­sponse.

When selecting a ceramic capacitor, only X5R and X7R di­electric types should be used. Other types such as Z5U and
Y5F have such severe loss of capacitance due to effects of
temperature variation and applied voltage, they may provide
as little as 20% of rated capacitance in many typical applica­tions. Always consult capacitor manufacturer’s data curves
before selecting a capacitor. High-quality ceramic capacitors
can be obtained from Taiyo-Yuden, AVX, and Murata.

SELECTING THE OUTPUT CAPACITOR

A single ceramic capacitor of value 4.7 µF to 10 µF will provide
sufficient output capacitance for most applications. For output
voltages below 10V, a 10 µF capacitance is required. If larger
amounts of capacitance are desired for improved line support
and transient response, tantalum capacitors can be used in
parallel with the ceramics. Aluminum electrolytics with ultra
low ESR such as Sanyo Oscon can be used, but are usually
prohibitively expensive. Typical AI electrolytic capacitors are
not suitable for switching frequencies above 500 kHz due to
significant ringing and temperature rise due to self-heating
from ripple current. An output capacitor with excessive ESR
can also reduce phase margin and cause instability.

SELECTING THE INPUT CAPACITOR

An input capacitor is required to serve as an energy reservoir
for the current which must flow into the inductor each time the
switch turns ON. This capacitor must have extremely low ESR
and ESL, so ceramic must be used. We recommend a nom-

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LM27313

inal value of 2.2 µF, but larger values can be used. Since this
capacitor reduces the amount of voltage ripple seen at the
input pin, it also reduces the amount of EMI passed back
along that line to other circuitry.

FEED-FORWARD COMPENSATION

Although internally compensated, the feed-forward capacitor
Cf is required for stability (see Typical Application Circuits).
Adding this capacitor puts a zero in the loop response of the
converter. Without it, the regulator loop can oscillate. The
recommended frequency for the zero fz should be approxi­mately 8 kHz. Cf can be calculated using the formula:

Cf = 1 / (2 x π x R1 x fz)

SELECTING DIODES

The external diode used in the typical application should be
a Schottky diode. If the switch voltage is less than 15V, a 20V
diode such as the MBR0520 is recommended. If the switch
voltage is between 15V and 25V, a 30V diode such as the
MBR0530 is recommended. If the switch voltage exceeds
25V, a 40V diode such as the MBR0540 should be used.

The MBR05xx series of diodes are designed to handle a max­imum average current of 500mA. For applications with load
currents to 800mA, a Microsemi UPS5817 can be used.

LAYOUT HINTS

High frequency switching regulators require very careful lay­out of components in order to get stable operation and low
noise. All components must be as close as possible to the
LM27313 device. It is recommended that a 4-layer PCB be
used so that internal ground planes are available.

As an example, a recommended layout of components is
shown:

3.

If internal ground planes are available (recommended)
use vias to connect directly to the LM27313 ground at
device pin 2, as well as the negative sides of capacitors
C1 and C2.

SETTING THE OUTPUT VOLTAGE

The output voltage is set using the external resistors R1 and
R2 (see Typical Application Circuits). A minimum value of

13.3 kΩ is recommended for R2 to establish a divider current
of approximately 92 µA. R1 is calculated using the formula:

R1 = R2 x ( (V

/ VFB) − 1 )

OUT

DUTY CYCLE

The maximum duty cycle of the switching regulator deter­mines the maximum boost ratio of output-to-input voltage that
the converter can attain in continuous mode of operation. The
duty cycle for a given boost application is defined as:

This applies for continuous mode operation.

The equation shown for calculating duty cycle incorporates
terms for the FET switch voltage and diode forward voltage.
The actual duty cycle measured in operation will also be af­fected slightly by other power losses in the circuit such as wire
losses in the inductor, switching losses, and capacitor ripple
current losses from self-heating. Therefore, the actual (effec­tive) duty cycle measured may be slightly higher than calcu­lated to compensate for these power losses. A good
approximation for effective duty cycle is :

DC (eff) = (1 - Efficiency x (VIN / V

OUT

))

Where the efficiency can be approximated from the curves
provided.

FIGURE 1. Recommended PCB Component Layout

Some additional guidelines to be observed:

1.

Keep the path between L1, D1, and C2 extremely short.
Parasitic trace inductance in series with D1 and C2 will
increase noise and ringing.

2.

The feedback components R1, R2 and CF must be kept
close to the FB pin of the LM27313 to prevent noise
injection on the high impedance FB pin.

20216822

INDUCTANCE VALUE

The first question we are usually asked is: “How small can I
make the inductor?” (because they are the largest sized com­ponent and usually the most costly). The answer is not simple
and involves trade-offs in performance. More inductance
means less inductor ripple current and less output voltage
ripple (for a given size of output capacitor). More inductance
also means more load power can be delivered because the
energy stored during each switching cycle is:

E = L/2 x (lp)

2

Where “lp” is the peak inductor current. An important point to
observe is that the LM27313 will limit its switch current based
on peak current. This means that since lp(max) is fixed, in­creasing L will increase the maximum amount of power avail­able to the load. Conversely, using too little inductance may
limit the amount of load current which can be drawn from the
output.

Best performance is usually obtained when the converter is
operated in “continuous” mode at the load current range of
interest, typically giving better load regulation and less output
ripple. Continuous operation is defined as not allowing the in­ductor current to drop to zero during the cycle. It should be
noted that all boost converters shift over to discontinuous op­eration as the output load is reduced far enough, but a larger
inductor stays “continuous” over a wider load current range.

To better understand these tradeoffs, a typical application cir­cuit (5V to 12V boost with a 10 µH inductor) will be analyzed.

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Since the LM27313 typical switching frequency is 1.6 MHz,
the typical period is equal to 1/f

0.625 µs.

LM27313

We will assume: VIN = 5V, V

0.5V. The duty cycle is:

= 12V, V

OUT

, or approximately

SW(TYP)

DIODE

= 0.5V, VSW =

Duty Cycle = ((12V + 0.5V - 5V) / (12V + 0.5V - 0.5V)) = 62.5%

The typical ON time of the switch is:

(62.5% x 0.625 µs) = 0.390 µs

It should be noted that when the switch is ON, the voltage
across the inductor is approximately 4.5V.

Using the equation:

V = L (di/dt)

We can then calculate the di/dt rate of the inductor which is
found to be 0.45 A/µs during the ON time. Using these facts,
we can then show what the inductor current will look like dur­ing operation:

FIGURE 2. 10 µH Inductor Current, 5V–12V Boost

During the 0.390 µs ON time, the inductor current ramps up

0.176A and ramps down an equal amount during the OFF
time. This is defined as the inductor “ripple current”. It can also
be seen that if the load current drops to about 33 mA, the
inductor current will begin touching the zero axis which means
it will be in discontinuous mode. A similar analysis can be
performed on any boost converter, to make sure the ripple
current is reasonable and continuous operation will be main­tained at the typical load current values.

MAXIMUM SWITCH CURRENT

The maximum FET switch current available before the current
limiter cuts in is dependent on duty cycle of the application.
This is illustrated in Figure 3 below which shows typical values
of switch current as a function of effective (actual) duty cycle:

20216812

20216825

FIGURE 3. Switch Current Limit vs Duty Cycle

CALCULATING LOAD CURRENT

As shown in the figure which depicts inductor current, the load
current is related to the average inductor current by the rela­tion:

I

LOAD

= I

IND(AVG)

x (1 - DC)

Where "DC" is the duty cycle of the application. The switch
current can be found by:

ISW = I

IND(AVG)

+ ½ (I

RIPPLE

)

Inductor ripple current is dependent on inductance, duty cy­cle, input voltage and frequency:

I

= DC x (VIN - VSW) / (fSW x L)

RIPPLE

Combining all terms, we can develop an expression which
allows the maximum available load current to be calculated:

The equation shown to calculate maximum load current takes
into account the losses in the inductor or turn-OFF switching
losses of the FET and diode. For actual load current in typical
applications, we took bench data for various input and output
voltages and displayed the maximum load current available
for a typical device in graph form:

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20216834

In this example, the LM27313 nominal switching frequency is

1.6 MHz, and the minimum switching frequency is

1.15 MHz. This means the maximum cycle period is the re­ciprocal of the minimum frequency:

T

We will assume: VIN = 5V, V
V

= 0.3V. The duty cycle is:

DIODE

= 1/1.15M = 0.870 µs

ON(max)

= 12V, VSW = 0.2V, and

OUT

Duty Cycle = ((12V + 0.3V - 5V) / (12V + 0.3V - 0.2V)) = 60.3%

Therefore, the maximum switch ON time is:

(60.3% x 0.870 µs) = 0.524 µs

An inductor should be selected with enough inductance to
prevent the switch current from reaching 800 mA in the 0.524
µs ON time interval (see below):

LM27313

FIGURE 4. Max. Load Current vs V

DESIGN PARAMETERS VSW AND I

SW

IN

The value of the FET "ON" voltage (referred to as VSW in the
equations) is dependent on load current. A good approxima­tion can be obtained by multiplying the "ON Resistance" of
the FET times the average inductor current.

FET on resistance increases at VIN values below 5V, since
the internal N-FET has less gate voltage in this input voltage
range (see Typical performance Characteristics curves).
Above VIN = 5V, the FET gate voltage is internally clamped to
5V.

The maximum peak switch current the device can deliver is
dependent on duty cycle. The minimum switch current value
(ISW) is guaranteed to be at least 800 mA at duty cycles below
50%. For higher duty cycles, see Typical performance Char­acteristics curves.

THERMAL CONSIDERATIONS

At higher duty cycles, the increased ON time of the FET
means the maximum output current will be determined by
power dissipation within the LM27313 FET switch. The switch
power dissipation from ON-state conduction is calculated by:

2

PSW = DC x I

IND(AVG)

x R

DS(ON)

There will be some switching losses as well, so some derating
needs to be applied when calculating IC power dissipation.

MINIMUM INDUCTANCE

In some applications where the maximum load current is rel­atively small, it may be advantageous to use the smallest
possible inductance value for cost and size savings. The con­verter will operate in discontinuous mode in such a case.

The minimum inductance should be selected such that the
inductor (switch) current peak on each cycle does not reach
the 800 mA current limit maximum. To understand how to do
this, an example will be presented.

20216813

FIGURE 5. Discontinuous Design, 5V–12V Boost

The voltage across the inductor during ON time is 4.8V. Min­imum inductance value is found by:

L = V x (dt/dl)

L = 4.8V x (0.524 µs / 0.8 mA) = 3.144 µH

In this case, a 3.3 µH inductor could be used, assuming it
provided at least that much inductance up to the 800 mA cur­rent value. This same analysis can be used to find the mini­mum inductance for any boost application.

INDUCTOR SUPPLIERS

Some of the recommended suppliers of inductors for this
product include, but are not limited to, Sumida, Coilcraft,
Panasonic, TDK and Murata. When selecting an inductor,
make certain that the continuous current rating is high enough
to avoid saturation at peak currents. A suitable core type must
be used to minimize core (switching) losses, and wire power
losses must be considered when selecting the current rating.

SHUTDOWN PIN OPERATION

The device is turned off by pulling the shutdown pin low. If this
function is not going to be used, the pin should be tied directly
to VIN. If the SHDN function will be needed, a pull-up resistor
must be used to VIN (50kΩ to 100 kΩ is recommended), or
the pin must be actively driven high and low. The SHDN pin
must not be left unterminated.

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Physical Dimensions inches (millimeters) unless otherwise noted

LM27313

Order Number LM27313XMF, or LM27313XMFX

5-Lead SOT-23 Package

NS Package Number MF05A

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Notes

LM27313

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Notes

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NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR
SYSTEMS WITHOUT THE EXPRESS PRIOR WRITTEN APPROVAL OF THE CHIEF EXECUTIVE OFFICER AND GENERAL
COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:

LM27313 1.6 MHz Boost Converter With 30V Internal FET Switch in SOT-23

Life support devices or systems are devices which (a) are intended for surgical implant into the body, or (b) support or sustain life and
whose failure to perform when properly used in accordance with instructions for use provided in the labeling can be reasonably expected
to result in a significant injury to the user. A critical component is any component in a life support device or system whose failure to perform
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Copyright© 2007 National Semiconductor Corporation

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