Datasheet LED7706 Datasheet (ST)

6-rows 30 mA LEDs driver with boost regulator

Features

■ Boost section

– 4.5 V to 36 V input voltage range
– Internal power MOSFET
– Internal +5 V LDO for device supply
– Up to 36 V output voltage
– Constant frequency peak current-mode

control

– 250 kHz to 1 MHz adjustable switching

frequency

– External synchronization for multi-device

application
– Pulse-skip power saving mode at light load
– Programmable soft-start
– Programmable OVP protection
– Stable with ceramic output capacitors
– Thermal shutdown

■ Backlight driver section

– Six rows with 30 mA maximum current

capability (adjustable)
– Rows disable option
– Less than 500 ns minimum dimming on-

time (1 % minimum dimming duty-cycle at

20 kHz)
– ±2 % current matching between rows
– LED failure (open and short-circuit)

detection

LED7706

for LCD panels backlight

VFQFPN-24 4x4

Description

The LED7706 consists of a high efficiency
monolithic boost converter and six controlled
current generators (rows) specifically designed to
supply LEDs arrays used in the backlighting of
LCD panels. The device can manage an output
voltage up to 36 V (i.e. 10 white LEDs per row).

The generators can be externally programmed to
sink up to 30 mA and can be dimmed via a PWM
signal (1 % dimming duty-cycle at 20 kHz can be
managed). The device allows to detect and
manage the open and shorted LED faults and to
let unused rows floating. Basic protections (output
over-voltage, internal MOSFET over-current and
thermal shutdown) are provided.

Applications

■ LCD monitors and TV panels

■ PDAs panel backlight

■ GPS panel backlight

Table 1. Device summary

Order codes Package Packaging

LED7706

VFQFPN-24 4x4 (exposed pad)

LED7706TR Tape and reel

April 2009 Rev 2 1/46

Tu b e

www.st.com

46

Contents LED7706

Contents

1 Typical application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

2 Pin settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

2.1 Connections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

2.2 Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

3 Electrical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

3.1 Maximum rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

3.2 Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

4 Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

5 Operation description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

5.1 Boost section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

5.1.1 Functional description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

5.1.2 Enable function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

5.1.3 Soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

5.1.4 Overvoltage protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14

5.1.5 Switching frequency selection and synchronization . . . . . . . . . . . . . . . 15

5.1.6 Slope compensation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

5.1.7 Boost current limit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

5.1.8 Thermal protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

5.2 Backlight driver section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19

5.2.1 Current generators . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19

5.2.2 PWM dimming . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

5.3 Fault management . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

5.3.1 FAULT pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

5.3.2 MODE pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

5.3.3 Open LED fault . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

5.3.4 Shorted LED fault . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

2/46

LED7706 Contents

6 Application information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

6.1 System stability . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

6.1.1 Loop compensation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

6.2 Thermal considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

6.3 Component selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

6.3.1 Inductor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

6.3.2 Capacitors selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29

6.3.3 Flywheel diode selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29

6.4 Design example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

6.4.1 Switching frequency setting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

6.4.2 Row current setting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

6.4.3 Inductor choice . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

6.4.4 Output capacitor choice . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

6.4.5 Input capacitor choice . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33

6.4.6 Over-voltage protection divider setting . . . . . . . . . . . . . . . . . . . . . . . . . 33

6.4.7 Compensation network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

6.4.8 Boost current limit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

6.4.9 Power dissipation estimate . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35

6.5 Layout consideration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

7 Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41

8 Package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43

9 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45

3/46

Typical application circuit LED7706

1 Typical application circuit

Figure 1. Application circuit

VIN

+5V

Internal MOS OCP

Rows current selection

Dimming

Fault

Enable

LDO5

BILIM

RILIM

SS

COMP

SGND

L

LX

VIN

AVCC

LED7706

FAULT

DIM

EN

Slope Compensation

OVSEL

SLOPE

MODE

SYNC

OVP selection

Switching Frequenc y selection

SWF

ROW1

ROW2

ROW3

ROW4

ROW5

ROW6

PGND

Faults Management Selection

Sync Output

MLCC

Up to 10 WLEDs per row

OUT

V

AM00594v1

4/46

LED7706 Pin settings

2 Pin settings

2.1 Connections

Figure 2. Pin connection (through top view)

COMP

RILIM

BILIM

FSW

MODE

AVCC

SYNC

SS

24

1

EN

DIM

FAULT

LX

19

18

OVSEL

PGND

LED7706

ROW6

ROW5

ROW4

ROW3

6

7

13

12

VIN

LDO5

SGND

SLOPE

ROW2

ROW1

AM00595v1

5/46

Pin settings LED7706

2.2 Pin description

Table 2. Pin functions

N° Pin Function

1COMP

2RILIM

3 BILIM

4FSW

5MODE

6 AVCC + 5 V analog supply. Connect to LDO5 through a simple RC filter.

7LDO5

8 VIN Input voltage. Connect to the main supply rail.

9SLOPE

10 SGND

11 ROW1 Row driver output #1.

Error amplifier output. A simple RC series between this pin and ground is
needed to compensate the loop of the boost regulator.

Output generators current limit setting. The output current of the rows can be
programmed connecting a resistor to SGND.

Boost converter current limit setting. The internal MOSFET current limit can
be programmed connecting a resistor to SGND.

Switching frequency selection and external sync input. A resistor to SGND is
used to set the desired switching frequency. The pin can also be used as
external synchronization input. See

Current generators fault management selector. It allows to detect and manage
LEDs failures. See

+ 5 V LDO output and power section supply. Bypass to SGND with a
1 μF ceramic capacitor.

Slope compensation setting. A resistor between the output of the boost
converter and this pin is needed to avoid sub-harmonic instability.

Refer to

Signal ground. Supply return for the analog circuitry and the current
generators.

Section 6.1 on page 24 for details.

Section 5.3.2 on page 21 for details.

Section 5.1.5 on page 15 for details.

12 ROW2 Row driver output #2.

13 ROW3 Row driver output #3.

14 ROW4 Row driver output #4.

15 ROW5 Row driver output #5.

16 ROW6 Row driver output #6.

17 PGND Power ground. Source of the internal Power MOSFET.

18 OVSEL

19 LX Switching node. Drain of the internal Power MOSFET.

20 DIM Dimming input. Used to externally set the brightness by using a PWM signal.

21 EN

22 FAULT

23 SYNC Synchronization output. Used as external synchronization output.

24 SS Soft-start. Connect a capacitor to SGND to set the desired soft-start duration.

6/46

Over-voltage selection. Used to set the desired OV threshold by an external
divider. See

Enable input. When low, the device is turned off. If tied high or left open, the
device is turned on and a soft-start sequence takes place.

Fault signal output. Open drain output. The pin goes low when a fault condition
is detected (see Section 5.3.1 on page 21 for details).

Section 5.1.4 on page 14 for details.

LED7706 Electrical data

3 Electrical data

3.1 Maximum rating

Table 3. Absolute maximum ratings

(1)

Symbol Parameter Value Unit

V

AVC C

V

LDO5

AVCC to SGND -0.3 to 6

LDO5 to SGND -0.3 to 6

PGND to SGND -0.3 to 0.3

V

V

VIN to PGND -0.3 to 40

IN

LX to SGND -0.3 to 40

LX

LX to PGND -0.3 to 40

RILIM, BILIM, SYNC, OVSEL, SS to SGND -0.3 to V

AVC C

+ 0.3

EN, DIM, SW, MODE, FAULT to SGND -0.3 to 6

ROWx to PGND/ SGND -0.3 to 40

SLOPE to VIN V

- 0.3 to VIN + 6

IN

SLOPE to SGND -0.3 to 40

Internal switch maximum RMS current
(flowing through LX node)

P

Power dissipation @ TA = 25 °C 2.3

TOT

2.0 A

(2)

Maximum withstanding voltage range test condition:
CDF-AEC-Q100-002- “human body model” acceptance

±1000 V

criteria: “normal performance”

1. Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the
device. Exposure to absolute maximum rated conditions for extended periods may affect device reliability.

2. Power dissipation referred to the device mounted on the demonstration board described in section 5.5

V

W

3.2 Thermal data

Table 4. Thermal data

Symbol Parameter Value Unit

R

T

Thermal resistance junction to ambient 42 °C/W

thJA

Storage temperature range -50 to 150 °C

STG

Junction operating temperature range -40 to 150 °C

T

J

7/46

Electrical characteristics LED7706

4 Electrical characteristics

VIN = 12 V; TJ = 25 °C and LDO5 connected to AVCC if not otherwise specified

Table 5. Electrical characteristics

(a)

Symbol Parameter Test condition Min. Typ. Max. Unit

Supply section

V

V

V

I

V

BST

LDO5

AVC C

IN,Q

Input voltage range 4.5 36 V

IN

Boost section output voltage 36

LDO output and IC supply
voltage

Operating quiescent current

EN High

= 0 mA

I

LDO5

= 51 kΩ, R

R

RILIM

R

SLOPE

= 680 kΩ

BILIM

4.4 5 5.5 V

= 220 kΩ,

1mA

DIM tied to SGND.

I

IN,SHDN

V

UVLO,ON

V

UVLO,OFF

Operating current in shutdown EN low 20 30 μA

LDO5 under voltage lock out
upper threshold

LDO5 under voltage lock out
lower threshold

3.3 3.6

3.8 4.0

LDO linear regulator

Line regulation 6 V

LDO dropout voltage V

LDO maximum output current

V

V

IN

LDO5

LDO5

≤ 36 V, I

IN

= 4.3 V, I

> V

UVLO,ON

< V

UVLO,OFF

LDO5

= 30 mA 30

LDO5

= 10 mA 80 120

25 40 60

20 30

≤ V

Boost section

V

mV

mA

t

ON,min

f

SW

Minimum switching on-time 200 ns

Default switching frequency FSW connected to AVCC 570 660 750

Minimum FSW sync frequency 210

FSW sync input threshold 240

FSW sync input hysteresis 20

FSW sync min. ON time 270 ns

SYNC output Duty-Cycle

SYNC output high level I

SYNC output low level I

a. Specification referred to TJ from 0 °C to +85 °C. Specification over the 0 to +85 °C TJ range are assured by

design, characterization and statistical correlation.

FSW connected to AVCC
(Internal oscillator selected)

= 10 μA

SYNC

= -10 μA 20

SYNC

8/46

V

AVC C

-20V

kHz

mVFSW sync low level 350

34 40 %

mV

LED7706 Electrical characteristics

Table 5. Electrical characteristics (continued)

Symbol Parameter Test condition Min. Typ. Max. Unit

Power switch

5

V

R

K

DSon

LX current coefficient R

B

Internal MOSFET on-resistance

OC and OV protections

= 600 kΩ 5·1056·1057·10

BILIM

280 500 mΩ

V

TH,OVP

V

TH,FRD

ΔV

OVP,FRD

Over voltage protection
reference threshold (OVSEL)

Floating channel detection
threshold

Voltage gap between OVP and
FRD thresholds

Soft-start and power management

EN, Turn-on threshold 1.6

EN, Turn-off threshold 0.8

DIM, high level threshold 1.3

DIM, low level threshold 0.8

EN, pull-up current 2.5

SS, charge current 4 5 6

SS, end-of-startup threshold 2.0 2.4 2.8

SS, reduced switching
frequency release threshold

Current generators section

T

DIM-

ON,min

K

ΔK

V

IFB

V

rowx,

FAULT

V

FAU LT,

LOW

Minimum dimming on-time 500 ns

Current generators gain 987 V

R

Current generators gain

(1)

R

accuracy

Feedback regulation voltage 400 mV

Shorted LED fault detection
threshold

FAULT pin low-level voltage I

Thermal shutdown

1.190 1.234 1.280

1.100 1.145 1.190

90 mV

0.8

MODE tied to SGND 3.4

MODE connected to AVCC 6.0

FAULT,SINK

= 4 mA 200 350 mV

V

V

μA

V

±2.0 %

V

1. I

T

SHDN

ROW

Thermal shutdown
turn-off temperature

Thermal shutdown hysteresis 30

= KR / R

, ΔI

RILIM

ROW/IROW ≈ ΔKR/KR

+ ΔR

RILIM/RRILIM

9/46

150

°C

Operation description LED7706

5 Operation description

The device can be divided into two sections: the boost section and the backlight driver
section. These sections are described in the next paragraphs. Figure 3 provides an
overview of the internal blocks of the device.

Figure 3. Simplified block diagram

VIN

LDO5

LDO5

COMP

COMP

BILIM

BILIM

SS

SS

SYNC

SYNC

FSW

FSW

AVCC

AVCC

EN

EN

MODE

MODE

FAULT

FAULT

DIM

DIM

VIN

+5V

+5V

LDO

LDO

UVLO

UVLO

Detector

Detector

UVLO

UVLO

Current Limit

Current Limit

Soft Start

Soft Start

Prot_EN

Prot_EN

Ext Sync

Ext Sync
Detector

Detector

CONTROL

CONTROL

LOGIC

LOGIC

Thermal

Thermal

Shutdown

Shutdown

SLOPE

SLOPE

Ramp

Ramp

Generator

Generator

÷2

÷2

OSC

OSC

Prot_EN

Prot_EN

Boost_EN

Boost_EN

UVLO

UVLO

CTRL6

CTRL6
CTRL5

CTRL5
CTRL4

CTRL4
CTRL3

CTRL3
CTRL2

CTRL2

OVP

OVP

1.2V

1.2V

+

+

+

+

+

+

g

m

g

m

_

_

Min Voltage

Min Voltage

Selector

Selector

CTRL1

CTRL1

ROW1

ROW1

V

V

I to V

I to V

Current Sense

Current Sense

ZCD

ZCD

+

+

Boost

Boost

Control

Control

_

_

Logic

Logic

0.7V

0.7V

Boost_EN

Boost_EN

CTRL6

CTRL6

CTRL5

CTRL5

CTRL4

CTRL4

CTRL3

CTRL3

CTRL2

CTRL2

OVP

OVP

V

ROW6

V

ROW6

V

ROW5

V

ROW5

V

ROW4

V

ROW4

V

ROW3

V

ROW3

V

ROW2

V

ROW2

4V

4V

LOGIC

LOGIC

Current

Current

Generator 1

Generator 1

Current

Current

Generator 6

Generator 6

Current

Current

Generator 5

Generator 5

Current

Current

Generator 4

Generator 4

Current

Current

Generator 3

Generator 3

Current

Current

Generator 2

Generator 2

MODE

MODE

I to V

I to V

+

+

_

_

_

_

+

+_+

V

V

TH,FLT

TH,FLT

1.172V

1.172V

LX

LX

PGND

PGND

OVSEL

OVSEL

ROW6

ROW6

ROW5

ROW5

ROW4

ROW4

ROW3

ROW3

ROW2

ROW2

ROW1

ROW1

10/46

RILIM

RILIM

SGND

SGND

AM00596v1

LED7706 Operation description

5.1 Boost section

5.1.1 Functional description

The LED7706 is a monolithic LEDs driver for the backlight of LCD panels and it consists of a
boost converter and six PWM-dimmable current generators.

The boost section is based on a constant switching frequency, peak current-mode
architecture. The boost output voltage is controlled such that the lowest row's voltage,
referred to SGND, is equal to an internal reference voltage (400 mV typ. see Figure 5). The
input voltage range is from 4.5 V up to 36 V. In addition, the LED7706 has an internal LDO
that supplies the internal circuitry of the device and is capable to deliver up to 40 mA. The
input of the LDO is the VIN pin.

The LDO5 pin is the LDO output and the supply for the power MOSFET driver at the same
time. The AVCC pin is the supply for the analog circuitry and should be connected to the
LDO output through a simple RC filter in order to improve the noise rejection.

Figure 4. AVCC filtering

VIN

VIN

LDO5

CLDO5

CLDO5

1uF

1uF

Rfilt

Rfilt
4R7

4R7

C

C

AVCC

AVCC

100nF

100nF

LDO5

AVCC

AVCC

LDO

LDO

SGND

SGND

LED7706

LED7706

AM00597v1

Two loops are involved in regulating the current sunk by the generators.

The main loop is related to the boost regulator and uses a constant frequency peak current­mode architecture to regulate the power rail that supplies the LEDs (Figure 5), while an
internal current loop regulates the same current (flowing through the LEDs) at each row
according to the set value (RILIM pin).

Figure 5. Main loop and current loop diagram

V

IN

LX

Slope

PWM

ROWx

SGND

COMP

RILIM

E/A

Error amplifier

Minimum voltage drop

selector

0.4V

11/46

AM00598v1

Operation description LED7706

A dedicated circuit automatically selects the lowest voltage drop among all the rows and
provides this voltage to the main loop that, in turn, regulates the output voltage. In fact, once
the reference generator has been detected, the error amplifier compares its voltage drop to
the internal reference voltage and varies the COMP output. The voltage at the COMP pin
determines the inductor peak current at each switching cycle. The output voltage of the
boost regulator is thus determined by the total forward voltage of the LEDs strings (see

Figure 6):

Equation 1

m

N

OUT

ROWS

=

1i

LEDS

Σ

=

mV400)V(maxV

+=

j,F

1j

where the first term represents the highest total forward voltage drop over N active rows and
the second is the voltage drop across the leading generator (400 mV typ.).

The device continues to monitor the voltage drop across all the rows and automatically
switches to the current generator having the lowest voltage drop.

Figure 6. Calculation of the output voltage of the boost regulator

Row with the highest voltage

V

IN

drop across LEDs

5.1.2 Enable function

The LED7706 is enabled by the EN pin. This pin is active high and, when forced to SGND,
the device is turned off. This pin is connected to a permanently active 2.5 μA current source;
when sudden device turn-on at power-up is required, this pin must be left floating or
connected to a delay capacitor. When turned off, the LED7706 quickly discharges the
soft-start capacitor and turns off the power MOSFET, the current generators and the LDO.
The power consumption is thus reduced to 20 μA only.

In applications where the dimming signal is used to turn on and off the device, the EN pin
can be connected to the DIM pin as shown in Figure 7.

Boost

controller

Current

generators

section

I

LED

400 mV

generator

max

Σ

Leading

V

F

V

BOOST

AM00599v1

12/46

LED7706 Operation description

Figure 7. Enable pin driven by dimming signal

DIM

DIM

BAS69

BAS69

EN

220kΩ

220kΩ

100nF

100nF

EN

LED7706

LED7706

SGND

SGND

AM00600v1

5.1.3 Soft-start

The soft-start function is required to perform a correct start-up of the system, controlling the
inrush current required to charge the output capacitor and to avoid output voltage overshoot.
The soft-start duration is set connecting an external capacitor between the SS pin and
ground. This capacitor is charged with a 5 μA (typ.) constant current, forcing the voltage on
the SS pin to ramp up. When this voltage increases from zero to nearly 1.2 V, the current
limit of the power MOSFET is proportionally released from zero to its final value. However,
because of the limited minimum on-time of the switching section, the inductor might saturate
due to current runaway. To solve this problem the switching frequency is reduced to one half
of the nominal value at the beginning of the soft-start phase. The nominal switching
frequency is restored after the SS pin voltage has crossed 0.8 V.

Figure 8. Soft-start sequence waveforms in case of floating rows

OVP

OVP

Floating ROWs detection

Floating ROWs detection

95% of

95% of
OVP

AVCC

AVCC

2.4V

2.4V

1.2V

1.2V

0.8V

0.8V

100%

100%

OVP

t

ss

t

ss

Current limit

Current limit

Output voltage

Output voltage

SS pin voltage

SS pin voltage

Protections turn active

Protections turn active

Nominal switching

Nominal switching
frequency release

frequency release

EN pin voltage

EN pin voltage

t

t

AM00601v1

During the soft-start phase the floating rows detection is also performed. In presence of one
or more floating rows, the voltage across the involved current generator drops to zero. This
voltage becomes the inverting input of the error amplifier through the minimum voltage drop

13/46

Operation description LED7706

selector (see Figure 5). As a consequence the error amplifier is unbalanced and the loop
reacts by increasing the output voltage. When it reaches the floating row detection (FRD)
threshold (95% of the OVP threshold), the floating rows are managed according to

Section 5.3 on page 21). After the SS voltage reaches a 2.4 V threshold, the start-up

(see
finishes and all the protections turn active. The soft-start capacitor C

can be calculated

SS

Ta bl e 6

according to equation 2.

Equation 2

⋅

tI

≅

C

SS

SSSS

4.2

Where I

= 5 µA and tSS is the desired soft-start duration.

SS

5.1.4 Overvoltage protection

An adjustable over-voltage protection is available. It can be set feeding the OVSEL pin with a
partition of the output voltage. The voltage of the central tap of the divider is thus compared
to a fixed 1.234 V threshold. When the voltage on the OVSEL pin exceeds the OV threshold,
the FAULT pin is tied low and the device is turned off; this condition is latched and the
LED7706 is restarted
occurs when the LDO output falls below the lower UVLO threshold and subsequently
crosses the upper UVLO threshold during the rising phase of the input voltage). Normally,
the value of the high-side resistors of the divider must be chosen as high as possible (but
lower than 1 M
(during the off phase of the dimming cycle). The R2/R1 ratio is calculated to trigger the OVP
circuitry as soon as the output voltage is 2 V higher than the maximum value for a given LED
string (see equation 3). Two additional filtering capacitors, C
improve noise rejection at the OVSEL pin, as shown in
in the 100 pF-330 pF range, while the C

Equation 3

by toggling the EN pin or by performing a Power-On Reset (the POR

Ω) to reduce the output capacitor discharge when the boost converter is off

Figure 9. The typical value for C10 is

13 value is given by equation 4.

RR

⋅=

12

OVP,OUT

V234.1

−+

10 and C13, may be required to

)V234.1V2V(

Equation 4

14/46

R

2

C2C ⋅=

1013

R

1

LED7706 Operation description

Figure 9. OVP threshold setting

V

IN

LX

OVSEL

LED7706

SGND

R

R

V

OUT

C

13

C

1

2

OUT

C

10

AM00602v1

5.1.5 Switching frequency selection and synchronization

The switching frequency of the boost converter can be set in the 250 kHz-1 MHz range by
connecting the FSW pin to ground through a resistor. Calculation of the setting resistor is
made using equation 5 and should not exceed the 100 kΩ-400 kΩ range.

Equation 5

F

SW

=

R

FSW

5.2

In addition, when the FSW pin is tied to AVCC, the LED7706 uses a default 660 kHz fixed
switching frequency, allowing to save a resistor in minimum component-count applications.

Figure 10. Multiple device synchronization

MASTER

AVCC

FSW SYNC

SYNC

FSW

LED7706

FSW

R

SGND

SLAVE

Sync Out

SYNC

LED7706

SGND

AM00603v1

The FSW pin can also be used as synchronization input, allowing the LED7706 to operate
both as master or slave device. If a clock signal with a 210 kHz minimum frequency is
applied to this pin, the device locks synchronized. The signal provided to the FSW pin must
cross the 270 mV threshold in order to be recognized. The minimum pulse width which
allows the synchronizing pulses to be detected is 270 ns. An Internal time-out allows
synchronization as long as the external clock frequency is greater than 210 kHz.

15/46

Operation description LED7706

Keeping the FSW pin voltage lower than 270 mV for more than 4.8 µs results in a stop of the
device switching activity. Normal operation is resumed as soon as FSW rises above the
mentioned threshold and the soft-start sequence is repeated.

The SYNC pin is a synchronization output and provides a 35 % (typ.) duty-cycle clock when
the LED7706 is used as master or a replica of the FSW pin when used as slave. It is used to
connect multiple devices in a daisy-chain configuration or to synchronize other switching
converters running in the system with the LED7706 (master operation). When an external
synchronization clock is applied to the FSW pin, the internal oscillator is over-driven: each
switching cycle begins at the rising edge of the clock, while the slope compensation
(

Figure 11) ramp starts at the falling edge of the same signal. Thus, to prevent sub-

harmonic instability (see

Section 5.1.6), the external synchronization clock is required to

have a 40 % maximum duty-cycle when the boost converter is working in continuous­conduction mode (CCM) in order to assure that the slope compensation is effective (starts
with duty-cycle lower than 40%)

Figure 11. External sync waveforms

FSW pin voltage (ext. sync)

Slave SYNC pin voltage

Slave LX pin voltage

5.1.6 Slope compensation

The constant frequency, peak current-mode topology has the advantage of very easy loop
compensation with output ceramic caps (reduced cost and size of the application) and fast
transient response. In addition, the intrinsic peak-current measurement simplifies the
current limit protection, avoiding undesired saturation of the inductor.

On the other side, this topology has a drawback: there is an inherent open loop instability
when operating with a duty-ratio greater than 0.5. This phenomenon is known as “Sub­Harmonic Instability” and can be avoided by adding an external ramp to the one coming
from the sensed current. This compensating technique, based on the additional ramp, is
called “slope compensation”. In
the small perturbation ΔI
and the system reverts to a stable situation.

L

270ns minimum

270mV threshold

AM00604v1

Figure 12, where the switching duty-cycle is higher than 0.5,

dies away in subsequent cycles thanks to the slope compensation

16/46

LED7706 Operation description

The SLOPE pin allows to properly set the amount of slope compensation connecting a
simple resistor R

between the SLOPE pin and the output. The compensation ramp

SLOPE

starts at 35% (typ.) of each switching period and its slope is given by the following equation:

Equation 6

⎞

VVV

−−

BEINOUT

⎟
⎟

⎠

Where KS = 5.8 ⋅10

10 s-1

⎛
⎜

KS

=

SE

⎜

R

⎝

, V

= 2 V (typ.) and SE is the slope ramp in [A/s].

BE

SLOPE

To avoid sub-harmonic instability, the compensating slope should be at least half the slope
of the inductor current during the off-phase when the duty-cycle is greater than 50%. The
value of

R

SLOPE

can be calculated according to equation 7.

Equation 7

)VVV(LK2

−−⋅⋅⋅

R

SLOPE

≤

−

INOUT

BEINOUTS

)VV(

Figure 12. Effect of slope compensation on small inductor current perturbation

(D > 0.5)

Inductor current (CCM)

Inductor current (CCM)

0.35

·T

0.35

·T

SW

SW

Programmed inductor peak current with

Programmed inductor peak current with

slope compensation (S

slope compensation (S

I

I

BOOST, PEAK

BOOST, PEAK

E)

E)

ΔI

ΔI

L

L

Inductor current

Inductor current
perturbation

perturbation

T

T

SW

SW

t

t

AM00606v1

17/46

Operation description LED7706

5.1.7 Boost current limit

The design of the external components, especially the inductor and the flywheel diode, must
be optimized in terms of size relying on the programmable peak current limit. The LED7706
improves the reliability of the final application giving the way to limit the maximum current
flowing into the critical components. A simple resistor connected between the BILIM pin and
ground sets the desired value. The voltage at the BILIM pin is internally fixed to 1.23 V and
the current limit is proportional to the current flowing through the setting resistor, according
to the following equation:

Equation 8

K

I =

PEAK,BOOST

where

B

The maximum allowed current limit is 5 A, resulting in a minimum setting resistor
R

> 120 kΩ. The maximum guaranteed RMS current in the power switch is 2 A.

BILIM

In a boost converter the RMS current through the internal MOSFET depends on both the
input and output voltages, according to equations 9a (DCM) and 9b (CCM).

B

R

BILIM

5

V106K

⋅=

The current limitation works by clamping the COMP pin voltage proportionally to R
Peak inductor current is limited to the above threshold decreased by the slope
compensation contribution.

Equation 9 a

Equation 9 b

5.1.8 Thermal protection

In order to avoid damage due to high junction temperature, a thermal shutdown protection is
implemented. When the junction temperature rises above 150 °C (typ.), the device turns off
both the control logic and the boost converter and holds the FAULT pin low. The LDO is kept
alive and normal operation is automatically resumed after the junction temperature has
been reduced by 30 °C.

.

BILIM

DV

⋅

I

rms,MOS

⎛

D

⎜

II

=

OUTrms,MOS

⎜

()

D1

−

⎝

IN

=

SW

⎛

1

⎜

+

⎜

2

12

⎝

D

3

LF

⋅

V

OUT

2

⎞
⎟

()()

⎟

LfI

⋅⋅

SWOUT

⎠

⎞

3

⎟

D1D

−
⎟

⎠

18/46

LED7706 Operation description

5.2 Backlight driver section

5.2.1 Current generators

The LED7706 is a LEDs driver with six channels (rows); each row is able to drive multiple
LEDs in series (max. 36 V) and to sink up to 30 mA maximum current, allowing to manage
different kinds of LEDs.

The LEDs current can be set by connecting an external resistor (R

) between the RILIM

RILIM

pin and ground. The voltage across the RILIM pin is internally set to 1.23 V and the rows
current is proportional to the RILIM current according to the following equation:

Equation 10

K

I =

ROWx

R

R

RILIM

Where KR = 987 V.

The graph in

Figure 13 better shows the relationship between I

ROW

and R

and helps to

RILIM

choose the correct value of the resistor to set the desired row current.

Figure 13. Row current vs R

35

30

25

20

15

RILIM

Row current vs R

RILIM

10

Row current [mA]

5

0

0 50 100 150 200 250

R

RILIM

[kΩ]

The maximum current mismatch between the rows is ± 2 % @ I

19/46

= 20 mA.

rowx

AM00605v1

Operation description LED7706

⋅

=

5.2.2 PWM dimming

The brightness control of the LEDs is performed by a pulse-width modulation of the rows
current. When a PWM signal is applied to the DIM pin, the current generators are turned on
and off mirroring the DIM pin behavior. Actually, the minimum dimming duty-cycle depends
on the dimming frequency.

The real limit to the PWM dimming is the minimum on-time that can be managed for the
current generators; this minimum on-time is approximately 500 ns.

Thus, the minimum dimming duty-cycle depends on the dimming frequency according to the
following formula:

Equation 11

fns500D

DIMmin,DIM

For example, at a dimming frequency of 20 kHz, 1% of dimming duty-cycle can be
managed.

During the off-phase of the PWM signal the boost converter is paused and the current
generators are turned off. The output voltage can be considered almost constant because of
the relatively slow discharge of the output capacitor. During the start-up sequence (see

Section 5.1.3 on page 13) the dimming duty-cycle is forced to 100% to detect floating rows

regardless of the applied dimming signal.

Figure 14. PWM dimming waveforms

20/46

LED7706 Operation description

5.3 Fault management

The main loop keeps the row having the lowest voltage drop regulated to about 400 mV.
This value slightly depends on the voltage across the remaining active rows. After the soft­start sequence, all protections turn active and the voltage across the active current
generators is monitored to detect shorted LEDs.

5.3.1 FAULT pin

The FAULT pin is an open-collector output, (with 4 mA current capability) active low, which
gives information regarding faulty conditions eventually detected. This pin can be used
either to drive a status LED or to warn the host system.

The FAULT pin status is strictly related to the MODE pin setting (see

5.3.2 MODE pin

The MODE pin is a digital input and can be connected to AVCC or SGND in order to choose
the desired fault detection and management. The LED7706 can manage a faulty condition
in two different ways, according to the application needs.
device detects and handles the internal protections related to the boost section (over­current, over-temperature and over-voltage) and to the current generators section (open and
shorted LEDs).

Table 6. Faults management summary

FAULT MODE to GND MODE to VCC

Internal MOSFET over

current

Output over voltage

Thermal shutdown

LED short shorted led

Open row(s)

FAULT pin HIGH

Power MOS turned OFF

FAULT pin LOW

Device turned OFF, latched condition

FAULT pin LOW. Device turned OFF.

Automatic restart after 30 C temperature drop.

FAULT pin LOW

Device turned OFF,

latched condition

(Vth = 3.4 V)

FAULT pin LOW

Device turned OFF at first

occurrence, latched condition

Ta bl e 6 for details).

Ta bl e 6 summarizes how the

FAULT pin LOW

Faulty row(s) disconnected

Device keeps on working with the

remaining row(s)

(Vth = 6 V)

FAULT pin HIGH Faulty row(s)

disconnected.

Device keeps on working with the

remaining row(s)

21/46

Operation description LED7706

5.3.3 Open LED fault

In case a row is not connected or a LED fails open, the device has two different behaviors
according to the MODE pin status.

Connecting the MODE pin to SGND, as soon as an open row is detected the FAULT pin is
tied low and the device is turned off. The internal logic latches this status: to restore the
normal operation, the device must be restarted by toggling the EN pin or performing a
Power-On Reset (POR occurs when the voltage at the LDO5 pin falls below the lower UVLO
threshold and subsequently rises above the upper one).

If the MODE pin is high (i.e. connected to AVCC), the LED7706 behaves in a different
manner: the open row is excluded from the control loop and the device continues to work
properly with the remaining rows. The FAULT pin is not affected. Thus, if less than six rows
are used in the application, the MODE pin must be set high.

Figure 15 shows an example of open channel detection in case of MODE connected to

AVCC.

At the point marked as “1” in
point on the output voltage is increased as long as the output voltage reaches the floating
row detection threshold (see
faulty row is disconnected and the device keeps on working only with the remaining rows.

Figure 15. Open channel detection (MODE to AVCC)

Figure 15, the row opens (row current drops to zero). From this

Section 5.1.3 on page 13). Then (point marked as “2”) the



2

121

22/46

LED7706 Operation description

5.3.4 Shorted LED fault

When a LED is shorted, the voltage across the related current generator increases of an
amount equal to the missing voltage drop of the faulty LED. Since the feedback voltage on
each active generator is constantly compared with a fault threshold V

TH,FAULT

detects the faulty condition and acts according to the MODE pin status.

, the device

If the MODE pin is low, the fault threshold is V

TH,FAULT

= 3.4 V. When the voltage across a
row is higher than this threshold, the FAULT pin is set low and the device is turned off. The
internal logic latches this status until the EN pin is toggled or a POR is performed.

In case the MODE pin is connected to AVCC, the fault threshold is set to 6 V. The LED7706
simply disconnects the rows whose voltage is higher than the threshold and the FAULT pin
is forced low. This option is also useful to avoid undesired triggering of the shorted-LED
protection simply due to the high voltage drop spread across the LEDs.

Figure 16 shows an example of shorted LED detection in case MODE is connected to GND.

At the point marked as “1” in

Figure 16 one LED fails becoming a short-circuit. The voltage

across the current generator of the channel where the failed LED is connected increases by
an amount equal to the forward voltage of the faulty LED. Since the voltage across the
current generator is above the threshold (3.4 V in this case), the device is turned off and the
fault pin is set low (point “2”).

Figure 16. Shorted LED detection (MODE to GND)



121

2

23/46

Application information LED7706

⋅

≤

6 Application information

6.1 System stability

The boost section of the LED7706 is a fixed frequency, current-mode converter. During
normal operation, a minimum voltage selection circuit compares all the voltage drops across
the active current generators and provides the minimum one to the error amplifier. The
output voltage of the error amplifier determines the inductor peak current in order to keep its
inverting input equal to the reference voltage (400 mV typ). The compensation network
consists of a simple RC series (R

The calculation of R

COMP

dynamic performance of the boost converter and is strictly related to the operating
conditions.

6.1.1 Loop compensation

The compensation network can be quickly calculated using equations 12 to 16. Once both
R
get the optimal dynamic performance from the application.

The first parameter to be fixed is the switching frequency. Normally, a high switching
frequency allows reducing the size of the inductor and positively affects the dynamic
response of the converter (wider bandwidth) but increases the switching losses. For most of
applications, the fixed value (660 kHz) represents a good trade-off between power
dissipation and dynamic response, allowing to save an external resistor at the same time. In
low-profile applications, the inductor value is often kept low to reduce the number of turns;
an inductor value in the 4.7 µH-15 µH range is a good starting choice.

COMP

and C

have been determined, a fine-tuning phase may be required in order to

COMP

and C

- C

COMP

is fundamental to achieve optimal loop stability and

COMP

) between the COMP pin and ground.

COMP

In order to avoid instability due to interaction between the DC-DC converter's loop and the
current generators' loop, the bandwidth of the boost should not exceed the bandwidth of the
current generators. A unity-gain frequency (f

) in the order of 30-40 kHz is acceptable. Also,

U

take care not to exceed the CCM-mode right half-plane zero (RHPZ).

Equation 12

F2.0f

SWU

Equation 13

OUT

2

⎞

min,IN

⎟
⎟

⎠

⋅π

⎞

⎛

V

OUT

⎟

⎜

⎟

⎜

I

OUT

⎠

⎝

L2

V

⎛
⎜

2

RM

2.0f

U

⋅≤

L2

⋅π

⎜

V

⎝

2.0

⋅=

Equation 14 a

V

V

min,IN

OUT

M =

24/46

LED7706 Application information

(

)

⋅⋅π

Equation 14b

V

OUT

R =

I

OUT

Where V

is the minimum input voltage and I

IN,min

is the overall output current.

OUT

Note that, the lower the inductor value (and the higher the switching frequency), the higher
the bandwidth can be achieved. The output capacitor is directly involved in the loop of the
boost converter and must be large enough to avoid excessive output voltage drop in case of
a sudden line transition from the maximum to the minimum input voltages.

However a more significant requirement concerns the output voltage ripple.

The output capacitor should be chosen in accordance with the following expression:

Equation 15

TII

⋅−

OFFOUTpeak,L

max,OUT

is the peak

L, peak

where ΔV

OUT, max

inductor current, T

C

>

OUT

V2

Δ⋅

is the maximum acceptable output voltage ripple, I

is the off-time of the switching cycle (for an extensive explanation see

OFF

Section 6.4.4 on page 31).

Once the output capacitor has been chosen, the R

can be calculated as:

COMP

Equation 16

Cf2

R

COMP

=

U

MgG

⋅⋅

EAM

Where GM = 2.7 S and g

Equation 16 places the loop bandwidth at f

= 375 µS

EA

. Then, the C

U

capacitor is determined to

COMP

place the frequency of the compensation zero 5 times lower than the loop bandwidth:

Equation 17

C

COMP

=

1

Rf2

⋅⋅π

COMPZ

Where fZ=fU/5.

In most of the applications an experimental approach is also very valid to compensate the
circuit. A simple technique to optimize different applications is to choose C
and to replace R

with a 10 kΩ trimmer adjusting its value to properly damp the output

COMP

COMP

= 4.7 nF

transient response. Insufficient damping will result in excessive ringing at the output and
poor phase margin.

Figure 17 (a and b) give an example of compensation adjustment for a typical application.

25/46

Application information LED7706

Figure 17. Poor phase margin (a) and properly damped (b) load transient responses

a) b)

Figure 18. Load transient response measurement set-up

VIN= 6V

+5V

6.8μH

C

IN

LX

VIN

LDO5

BILIM

RILIM

SS

COMP

SGND

AVCC

LED7706

DIM

FAULT

EN

FSW

OVSEL

SLOPE

ROW1

ROW2

ROW3

ROW4

ROW5

ROW6

PGND

SYNC

MODE

Up to 10 WLEDs per row

V

4.7μF

MLCC

BOOST

R

=

L

500Hz

V

BST

50mA

AM00607v1

26/46

LED7706 Application information

⋅+=

⋅⋅=

6.2 Thermal considerations

In order to prevent the device from exceeding the thermal shutdown threshold (150 °C), it is
important to estimate the junction temperature through the following equation:

Equation 18

PRTT

tot,DJA,thAmbJ

where TA is the ambient temperature, R
ambient and P

The R

measured on the application demonstration board (described in Section 6.5)

th,JA

is the power dissipated by the device.

D,tot

is the equivalent thermal resistance junction to

th,JA

is 42 °C/W.

The P

has several contributions, listed below.

D,tot

a) Conduction losses due to the R

of the internal power switch, equal to:\

DS(on)

Equation 19

2

DDIRP ⋅⋅⋅=

INDSoncond,D

DIM

where D is defined as:

Equation 20

V

IN

1D −=

V

OUT

and D

is the duty cycle of the PWM dimming signal.

DIM

b) Switching losses due to the power MOSFET turn on and off, calculated as:

Equation 21

)tt(

+

fr

fIVP ⋅

⋅⋅⋅=

swINOUTsw,D

D

2

DIM

where tr and tf are the power MOSFET rise time and fall time respectively.

c) Current generators losses. This contribution is strictly related to the LEDs used in

the application. Only the contribution of the leading current generator (“master”
current generator) can be predicted, regardless of the LEDs forward voltage:

Equation 22

DVIP

DIMIFBROWMaster,GEN

where I

is the current flowing through the row, whereas V

ROW

is the voltage across the

IFB

master current generator (typically 400 mV).

The voltages across the other current generators depend on the spread of the LEDs forward
voltage. The worst case for power dissipation (maximum forward voltage LEDs in the master
row, minimum forward voltage LEDs in all other rows) can be estimated as:

27/46

Application information LED7706

(

(

)

⋅⋅Δ+⋅−⋅

(

⋅−=

Equation 23

=

where n
voltage and n

d) LDO losses, due to the dissipation of the 5 V linear regulator:

Equation 24

The LED7706 is housed in a 24 leads 4x4-VFQFPN package with exposed pad that allows
good thermal performance. However it is also important to design properly the
demonstration board layout in order to assure correct heat dissipation.

Figure 19 shows a picture of the LED7706 application demonstration board taken using an

infrared camera. The chip temperature, in those application conditions, is kept below 50 °C.

Figure 19. Demonstration board thermographic analysis

is the number of active rows, ΔV

ROWs

is the number of LEDs per row.

LEDs

)

is the spread of the LEDs forward

f,LEDs

)

IVVP

LDOLDOINLDO,D

DnVV1nIP

DIMLEDsLEDs,fIFBROWsROWGEN

VIN= 12V

VIN= 12V

= 20m A

= 20m A

I

I

ROW

ROW

= 34V

= 34V

V

V

OUT

OUT

= 660kHz

= 660kHz

F

F

SW

SW

= 90%

= 90%

D

D

DIM

DIM

= 30°C

= 30°C

T

T

Amb

Amb

6.3 Component selection

6.3.1 Inductor selection

Being the LED7706 mostly dedicated to backlighting, real-estate applications dictate severe
constrain in selecting the optimal inductor. The inductor choice must take into account
different parameters like conduction losses (DCR), core losses (ferrite or iron-powder),
saturation current and magnetic-flux shielding (core shape and technology).

The switching frequency of the LED7706 can be set in the 200 kHz-1 MHz range, allowing a
wide selecting room for the inductance value. Low switching frequencies takes to high
inductance value, resulting in significant DCR and size. On the other hand, high switching
frequencies result in significant core losses. The suggested range is 4.7-22 µH, even if the

28/46

LED7706 Application information

best trade off between the different loss contributions varies from manufacturer to
manufacturer.

A 6.8 µH inductor has been experimentally found as the most suitable for applications
running at a 660 kHz switching frequency.

Table 7. Recommended inductors

Manufacturer Part number Description Size

Coilcraft LPS6235-682MLC 6.8 μH, 75 mΩ, 2.7 A 6x6 mm

Wurth 7440650068 6.8 μH, 33 mΩ, 3.6 A 10x10 mm

6.3.2 Capacitors selection

The input and output capacitors should have very low ESR (ceramic capacitors) in order to
minimize the ripple voltage. The boost converter of the LED7706 has been designed to
support ceramic capacitors. The required capacitance depends on the programmed LED
current and the minimum dimming frequency (the boost converter is off when the DIM pin is
low and the output capacitor is slowly discharged). Considering the worst case (i.e. 200 Hz
dimming frequency and 30 mA/channel), two 2.2 µF MLCCs are suitable for almost all
applications. Particular care must be taken when selecting the rated voltage and the
dielectric type of the output capacitors: 50 V rated MLCC may show a significant
capacitance drop when biased, especially in case of Y5V dielectric.

As in most of boost converters, the input capacitor is less critical, although it is necessary to
reduce the switching noise on the supply rail. The input capacitor is also important for the
internal LDO of the LED7706 and must be kept as close as possible to the chip. The rated
voltage of the input capacitor can be chosen according to the supply voltage range; a 10 µF
X5R MLCC is recommended.

Table 8. Recommended capacitors

Manufacturer Part number Description Package Notes

Taiyo Yuden UMK325BJ106KM-T Ceramic, 35V, X5R, 20 % SMD 1210 C

Murata GRM31CR71H225KA88B Ceramic, 50V, X7R, 20 % SMD 1206 C

Murata GRM31CR71H475KA88B Ceramic, 50V, X7R, 20 % SMD 1206 C

6.3.3 Flywheel diode selection

The flywheel diode must be a Schottky type to minimize the losses. This component is
subject to an average current equal to the output one and must sustain a reverse voltage
equal to the maximum output rail voltage. Considering all the channels sinking 30mA each
(i.e. 180 mA output current) and the maximum output voltage (36 V), the STP1L40M
(I

= 1 A, Vr = 40 V) diode is a good choice. Smaller diodes can be used in applications

f,ave

involving lower output voltage and/or lower output current.

IN

OUT

OUT

29/46

Application information LED7706

(

6.4 Design example

In order to help the design of an application using the LED7706, in this section a simple
step-by-step design example is provided.

A possible application could be the LED backlight in a 15” LCD panel using the LED7706.

Here below the possible application conditions are listed:

● V

● 2 strings of 48 white LEDs (20 mA) each (arranged in 6 rows, 8 LEDs per row)

● V

6.4.1 Switching frequency setting

To reduce the number of the external components, the default switching frequency is
selected (660 kHz typ.) by connecting the FSW pin to AVCC pin.

However, in case a different switching frequency is required, a resistor from FSW pin and
ground can be connected, according to the equation (5) in

6.4.2 Row current setting

= 12 ± 20 %

IN

= 3.5 V ± 200 mV

F, L ED s

Section 5.1.5.

Considering the equation 10 in Section 5.2.1, the R

Equation 25

The closest standard commercial value is 51 kΩ. The actual value of the row current will be
a little lower (19.3 mA).

6.4.3 Inductor choice

The boost section, as all DC-DC converters, can work in CCM (continuous conduction
mode) or in DCM (discontinuous conduction mode) depending on load current, input and
output voltage and other parameters, among which the inductor value.

In a boost converter it is usually preferable to work in DCM.

Once the load, the input and output voltage, and the switching frequency are fixed, the
inductor value defining the boundary between DCM and CCM operation can be calculated
as:

Equation 26

R

RILIM

K

I

ROW

L

resistor can be calculated as:

RILIM

R

0

=

B

V987

mA20

2

)

D1DR

−⋅⋅

F2

⋅

SW

Ω=== k35.49

where D is the duty-cycle defined as:

30/46

LED7706 Application information

=⋅=

=+⋅

=

(

)

μ=<

Equation 27

=

⎧

V

IN

V

OUT

=−=

⎨
⎩

1D

min,IN

max,IN

V6.9V@68.0

=

V4.14V@52.0

whereas R0 is:

Equation 28

V

R

OUT

0

I

OUT

Ω== 250

and

Equation 29

ROWOUT

mA120I6I

The output voltage in the above calculations is considered as the maximum value (LED with
the maximum forward voltage connected to the leading generator):

Equation 30

max,LEDs,Fmax,OUT

V30mV400V8V

Considering the input voltage range, the lower LB will be at the lower input voltage. Hence
the condition to assure the DCM operation becomes:

Equation 31

min,INB

H2.13VLL

An inductor value of 6.8 µH could be a suitable value, considering also a margin from the
boundary condition.

It is important to highlight that the inductor choice involves not only the value itself but the
saturation current (higher than the current limit, see

Section 6.4.4), the rated RMS current

(the compliance with the saturation current might be not enough; also the thermal
performances must be taken into account), the DCR (which affects the efficiency) and the
size (in some application might be a strict requirement).

However the DCR can’t be reduced keeping the size small. Hence a trade off between these
two requirements must be achieved according to the application.

6.4.4 Output capacitor choice

The choice of the output capacitor is mainly affected by the desired output voltage ripple.

Since the voltage across the LEDs can be considered almost constant, this ripple is
transferred across the current generators, affecting their dynamic response.

The output ripple can be estimated as (neglecting the contribution of ESR of C
in case of MLCC):

31/46

, very low

OUT

Application information LED7706

(

)

Equation 32

⋅−

TII

OUT

OFFOUTpeak,L

`

V

OUT

=Δ

⋅

C2

where I

is the inductor peak current (see Figure 20) calculated as:

L, peak

Equation 33

⎧

⋅

DV

IN

sw

=

⎨

⋅

LF

⎩

=

I

peak,L

whereas D, working in DCM, is:

Equation 34

⎧

−⋅⋅⋅

=

D

sw

R

0

)1M(MLF2

=

⎨
⎩

defining M as:

Equation 35

M

⎨

V

IN

⎩

⎧

V

OUT

=

Figure 20. Inductor current in DCM operation

=

min,IN

max,IN

=

min,IN

=

max,IN

V6.9V@A044.1

=

V4.14V@A914.0

=

min,IN

max,IN

V6.9V@488.0

=

V4.14V@285.0

V6.9V@125.3

V4.14V@083.2

I

L

I

L, peak

T

T

can be calculated as:

OFF

Equation 36

defining D2 as:

ON

T

OFF

TSW= 1/F

SW

=

⎧

=⋅=

DTT

⎨

2SWOFF

⎩

min,IN

max,IN

V6.9V@ns5.348

=

V4.14V@ns5.398

t

AM00608v1

32/46

LED7706 Application information

(

)

Equation 37

=

⎧

⋅⋅⋅

MLF2

=

D

2

SW

()

0

=

−⋅

⎨

1MR

⎩

min,IN

max,IN

V6.9V@23.0

=

V4.14V@263.0

The worst case for the output voltage ripple is when input voltage is lower (V

A simple way to select the C

In order to affect as less as possible the current generators, it would be better to fix the
maximum ripple lower than the typical voltage across the generators.

For example considering

ΔV

leading generator), the required capacitance is:

Equation 38

A margin from the calculated value should be taken into account because of the
capacitance drop due to the applied voltage when MLCCs are used.

A 4.7 µF MLCC can be a good choice for this application (two 2.2 μF MLCC in parallel can
be also a good solution).

In case a dimming duty cycle different from 100 % is used, a further contribution to the
capacitor discharge (during the off time of the dimming cycle) should be considered.

6.4.5 Input capacitor choice

The input capacitor of a boost converter is less critical than the output capacitor, due to the
fact that the inductor is in series with the input, and hence, the input current waveform is
continuous.

IN,min

value is fixing a maximum voltage ripple.

OUT

lower than 80 mV (i.e. the 20 % of the voltage across the

OUT

TII

⋅−

C

>

OUT

V2

Δ⋅

OFFOUTpeak,L

max,OUT

F02.2

μ=

= 9.6 V).

A low ESR capacitor is always recommended.

A capacitor of 10 µF is tentatively a good choice for most of the applications.

6.4.6 Over-voltage protection divider setting

The over-voltage protection (OVP) divider provides a partition of the output voltage to the
OVSEL pin. The OVP divider setting not only fixes the OVP threshold, but also the open­channel detection threshold (95 % of the OVP threshold).

The proper OVP divider setting can be calculated by the equation (3):

Equation 39

RR

⋅=

where V

OUT, MAX

the maximum V

12

is the maximum output voltage considering the worst case (all LEDs with
= V

F

= 3.7 V on the same row):

F, m a x

MAX,OUT

33/46

V234.1

−+

)V234.1V2V(

Application information LED7706

=+⋅

=

Equation 40

R1 can be chosen is in the order of hundreds of kilo-ohms to reduce the leakage current in
the resistor divider. For example, setting R
standard commercial value is R

The correct selection of the OVP divider (and, as consequence, the open channel detection
threshold) must take into account the shorted LED fault threshold (see

When the selected OVP threshold is too high, an undesired triggering of the shorted LED
detection circuitry may occur when a channel opens.

In fact, as explained in section 4.3.3, once a channel opens, the device reacts increasing the
output voltage and the voltage across the working current generators may reach the shorted
LED detection threshold before the open channel detection threshold.

6.4.7 Compensation network

For the compensation network, the suggestions provided in Section 6.1 are always valid.

In this condition, tentatively the following value of R3 and C8 (see
good choice for the loop stability:

R

= 2.4 kΩ

3

C

= 4.7 nF

8

= 22 kΩ.

2

max,FOVP,OUT

= 510 kΩ leads to R2 = 21.89 kΩ. The closest

1

V30mV400VnLEDV

Section 5.3.4).

Figure 23) are usually a

6.4.8 Boost current limit

The boost current limit is set to protect the internal power switch against excessive current.
The slope compensation may reduce the programmed current limit. Hence, to take into
account this effect, as a rule of thumb, the current limit can be set as twice as much the
maximum inductor peak current (see

I

BOOST, PEAK

Therefore, using equation (8) and choosing I

Equation 41

> 2.09 A

Section 6.4.4):

R

BILIM

BOOST, PEAK

K

I

= 2.5 A, R

B

PEAK,BOOST

Ω== k240

BILIM

will be:

34/46

LED7706 Application information

=⋅⋅

=

(

)

⋅⋅Δ+⋅

−

=++

+

=⋅+

=

6.4.9 Power dissipation estimate

As explained in Section 6.2, there are several contributions to the total power dissipation.

Neglecting the power dissipated by the LDO (surely less significant compared with the other
contributions), equation (19), (21), (22) and (23) help to estimate the overall power
dissipation.

Before starting the power dissipation estimate it is important to highlight that the following
calculations are considering the worst case (the actual value of the dissipated power would
require measurements). Therefore the power dissipation is estimated according to the
following assumptions:

1. Minimum input voltage (9.6 V), which leads to maximum input current (and also D will

have the higher value, see

2. Maximum R

of the internal power MOSFET;

DS(on)

3. LEDs in the row of the leading generator will have the maximum forward voltage,

whereas all other LEDs in the other rows will have the minimum forward voltage.

4. 100 % dimming signal duty cycle is considered.

The conduction and switching losses on the internal power switch can be calculated as:

Equation 42

Section 6.4.4);

2

INDSoncond,D

DIM

mW34DDIRP

=⋅⋅⋅=

Equation 43

+

)tt(

fIVP

swINOUTsw,D

2

DIM

fr

⋅⋅⋅=

mW111D

=⋅

where tr = tf = 15 ns

The power dissipation related to the current generators is given by:

Equation 44

mW8DVIP

DIMIFBROWMaster,GEN

Equation 45

()

⋅=

DIMLEDsLEDs,fIFBROWsROWGEN

mW360DnVV1nIP

=

Equation 46

≅

GENMaster,GENsw,Dcond,Dtot,D

mW513PPPPP

The junction temperature can be estimated by equation (18) considering TA = 25 °C:

Equation 47

C5.46PRTT

tot,DJA,thAmbJ

35/46

°

Application information LED7706

=⋅⋅

=

=++

=

In order to estimate also the efficiency, other contributions to the power dissipation must be
added to P

(which represents only the power dissipated by the device), that is:

D, t o t

Equation 48

mW5.34DIVP

where V

F, Diode

2INDiode,FDiode,DISS

= 0.4 V

Equation 49

2

RMS,IndInd,DISS

2
IN

mW2.11IDCRIDCRP

=⋅≅⋅=

where DCR = 80 mΩ (typical DCR of the recommended inductors).

Therefore the total dissipated power is:

Equation 50

Ind,DISSDiode,DISStot,DTOTAL,DISS

mW1.559PPPP

Considering the input power as the result of input voltage multiplied by the input current, the
estimated efficiency is:

Equation 51

PP

−

=η

TOT,DISSIN

P

IN

845.0

=

Note: It is important to remind that the previous calculations consider the worst case, especially for

the power dissipated on the current generators.
Statistical analysis (confirmed by bench measurements) shows that the series connection of
more LEDs on each channel leads to compensation effects.
The hypothesis 3 above mentioned is thus rather unlikely.
Therefore P

is significantly lower and the overall efficiency is typically around 90 %.

GEN

36/46

LED7706 Application information

6.5 Layout consideration

1. A careful PCB layout is important for proper operation. In this section some guidelines

are provided in order to achieve a good layout.

2. The device has two different ground pins: signal ground (SGND) and power ground

(PGND). The PGND pin handles the switching current related to the boost section; for
this reason the PCB traces should be kept as short as possible and with adequate
width.

3. The signal ground is the return for the device supply and the current generators and

can be connected to the thermal pad.

4. The heat dissipation area (adequate to the application conditions) should be placed

backside respect to the device and with the lowest thermal impedance possible (i.e.
PCB traces in the backside should be avoided). The dissipation area is thermally and
electrically connected to the thermal pad by several vias (nine vias are recommended).

5. The signal and power grounds must be connected together in a single point as close as

possible to the PGND pin to reduce ground loops.

6. The R-C components of the compensation network should be placed as close as

possible to the COMP pin in order to avoid noise issue and instability of the
compensation.

7. Noise sensitive signals (i.e. feedbacks and compensation) should be routed as short as

possible to minimize noise collection. The LED7706 pinout makes it easy to separate
power components (e.g. inductor, diode) from signal ones.

8. The LX switching node should have and adequate width for high efficiency.

9. The critical power path inductor-LX-PGND must be as short as possible by mounting

the inductor, the diode and COUT as close as possible each other.

10. The capacitors of the compensated divider connected to the OVSEL pin should be

placed as close as possible to the OVSEL pin.

11. In order to assure good performance in terms of row current accuracy/mismatch, the

PCB traces from the rows pins to the LEDs should have similar length and width.

12. The capacitors of the filter connected to LDO5 and VIN pins should be mounted as

close as possible to the mentioned pins

Figure 21 and Figure 22 shows the demonstration board layout (top view and bottom view

respectively).

37/46

Application information LED7706

Figure 21. Demonstration board layout (top view)



Figure 22. Demonstration board layout (bottom view)

Figure 23

the used components and their value.

38/46

shows the LED7706 demonstration board application circuit, whereas Ta bl e 9 lists

LED7706 Application information

Figure 23. LED7706 demonstration board schematic

AM00609v1

39/46

Application information LED7706

Table 9. LED7706 demonstration board component list

Component Description Package Part number MFR Value

C1

C2,C3

C4 SMD 1206 GRM31CR71H225KA88B N.M.

C5

C6 100 nF

C7 3.3 nF

C8 4.7 nF

C9 N.M.

C10 220 pF

C11 4.7 nF

C12 N.M.

C13 15 pF

R1

R2 16 kΩ

R3 2.4 kΩ

R4 4.7 Ω

R5 330 kΩ

R6 51 kΩ

R7

Ceramic, 35V

X5R, 20%

Ceramic, 50V

X7R, 20%

Ceramic, 25V

X5R, 20%

Chip resistor

0.1 W, 1%

SMD 1210 UMK325BJ106KM-T Taiyo Yuden 10 µF

SMD 1206 GRM31CR71H225KA88B

Murata

SMD 0603 Standard

SMD 0603 Standard

2.2 µF

1 µF

510 kΩ

180 kΩ

R8 680 kΩ

R9, R10 100 kΩ

R11 1.2 kΩ

R12 N.M.

R13 N.M.

L1 6u8, 75 mΩ, 2.7 A 6x6 mm LPS6235-682MLC Coilcraft 6.8 µF

D1 Schottky, 40 V, 1 A DO216-AA STPS1L40M ST STPS1L40M

D2 Red LED, 3 mA SMD 0603 Standard

D3 Signal Schottky SOD-523 BAS69 N.M.

U1 Integrated circuit QFN4x4 LED7706 ST LED7706

J2 PCB pad jumper

J8 Header 8 SIL 8 Standard

SW1, SW2 Jumper 3 SIL 3 Standard

SW3 Push button 6x6 mm FSM4JSMAT TYCO

40/46

Chip resistor

0.1 W, 1%

SMD 0603 Standard

LED7706 Electrical characteristics

7 Electrical characteristics

Figure 24. Efficiency versus DIM duty cycle,

V

= 6 V, 6 rows, 10 white LEDs

IN

(20 mA) in series

Figure 26. Efficiency versus DIM duty cycle,

V

= 18 V, 6 rows, 10 white LEDs

IN

(20 mA) in series

Figure 25. Efficiency versus DIM duty cycle,

VIN = 12 V, 6 rows, 10 white LEDs
(20 mA) in series

Figure 27. Efficiency versus DIM duty cycle,

VIN = 24 V, 6 rows, 10 white LEDs
(20 mA) in series

Figure 28. Soft-start waveforms (EN, SS, and

V

monitored)

OUT

Figure 29. Boost section switching signals

(LX, SYNC and inductor current
monitored), V

41/46

= 12 V, 10 LEDs

IN

Electrical characteristics LED7706

Figure 30. Dimming waveforms

(F

= 200 Hz)

Figure 32. Dimming waveforms (F

DIM

= 5 kHz) Figure 33. Dimming waveforms (F

DIM

Figure 31. Dimming waveforms (F

= 1 kHz)

DIM

= 20 kHz)

DIM

Figure 34. Dimming waveforms

(T

42/46

DIM,ON

= 10 µs)

Figure 35. Dimming waveforms with the

minimum dimming on time (500 ns)

LED7706 Package mechanical data

8 Package mechanical data

In order to meet environmental requirements, ST offers these devices in different grades of
ECOPACK
specifications, grade definitions and product status are available at: www.st.com.
ECOPACK

®

packages, depending on their level of environmental compliance. ECOPACK®

®

is an ST trademark.

43/46

Package mechanical data LED7706

Table 10. VFQFPN-24 4 mm x 4 mm mechanical data

mm

Dim.

Min. Typ. Max.

A 0.80 0.90 1.00

A1 0.00 0.02 0.05

A3 0.20

b 0.18 0.25 0.30

D 3.85 4.00 4.15

D2 2.40 2.50 2.60

E 3.85 4.00 4.15

E2 2.40 2.50 2.60

e0.50

L 0.30 0.40 0.50

ddd 0.08

Figure 36. Package dimensions

44/46

LED7706 Revision history

9 Revision history

Table 11. Document revision history

Date Revision Changes

08-Feb-2008 1 Initial release

Added: Chapter 3 on page 7, Chapter 3 on page 7 and Chapter 3 on

page 7

09-Apr-2009 2

Updated: Chapter 3 on page 7, Chapter 3 on page 7, Chapter 3 on

page 7, Ta bl e 4 , Tab l e 5 , Figure 3, Figure 4, Figure 7, Figure 23 and
Ta bl e 9

45/46

LED7706

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