Analog Devices AN563 Application Notes

AN-563

a

APPLICATION NOTE

One Technology Way • P.O. Box 9106 • Norwood, MA 02062-9106 • 781/329-4700 • World Wide Web Site: http://www.analog.com

A Tamper-Resistant Watt-Hour Energy Meter

Based on the AD7751 and Two Current Sensors

by Anthony Collins and William Koon

INTRODUCTION

This application note describes a low-cost, high-accuracy
watt-hour meter based on the AD7751. The meter
described is intended for use in single-phase, two-wire
distribution systems. However, the design can easily be
adapted to suit specific regional requirements, e.g., in
the United States power is usually distributed to resi­dential customers as single-phase, three-wire.

The AD7751 is a low-cost, single-chip solution for elec­trical energy measurement. The most distinctive feature
of the AD7751 is that it continuously monitors the phase
and neutral (return) currents. A FAULT condition occurs
if the two currents differ by more than 12.5%. Power cal­culation will be based on the larger of the two currents.
The meter calculates power correctly even if one of the
two wires does not carry any current. AD7751 provides
an effective way to combat any attempt to return the
current through earth, a very simple yet effective way of
meter tampering. The AD7751 is comprised of two
ADCs, reference circuit, and all the signal processing
necessary for the calculation of real (active) power. The
AD7751 also includes direct drive capability for electrome­chanical counters (i.e., the energy register) and has a
high-frequency pulse output for calibration and com­munications purposes.

This application note should be used in conjunction with
the AD7751 data sheet. The data sheet provides detailed
information on the functionality of the AD7751 and will
be referenced several times in this application note.

DESIGN GOALS

The International Standard IEC1036 (1996-09)—

Alternat­ing current watt-hour meters for active energy (Classes
1 and 2)

design. For readers more familiar with the ANSI C12.16
specification, see the section at the end of this application
which compares the IEC1036 and ANSI C12.16 stan­dards. This section explains the key IEC1036
specifications in terms of their ANSI equivalents.

The design greatly exceeds this basic specification for
many of the accuracy requirements, e.g., accuracy at
unity power factor and at low (PF = ±0.5) power factor. In

, was used as the primary specification for this

addition, the dynamic range performance of the meter
has been extended to 500. The IEC1036 standard speci­fies accuracy over a range of 5% Ib to I
Typical values for I
outlines the accuracy requirements for a static watt­hour meter. The current range (dynamic range) for
accuracy is specified in terms of Ib (basic current).

Table I. Accuracy Requirements

Current Value

0.05 Ib < I < 0.1 Ib 1 ±1.5% ±2.5%

0.1 Ib

0.1 Ib

0.2 Ib < I < I

NOTES

1

The current ranges for specified accuracy shown in Table I are expressed
in terms of the basic current (Ib). The basic current is defined in IEC1036
(1996-09) Section 3.5.1.1 as the value of current in accordance with
which the relevant performance of a direct connection meter is fixed.
I

is the maximum current at which accuracy is maintained.

MAX

2

Power Factor (PF) in Table I relates the phase relationship between the
fundamental (45 Hz to 65 Hz) voltage and current waveforms. PF in this
case can be simply defined as PF = cos( φ), where φ is the phase angle
between pure sinusoidal current and voltage.

3

Class index is defined in IEC1036 (1996-09) Section 3.5.5 as the limits of
the permissible percentage error. The percentage error is defined as:

Percentage Error =

The schematic in Figure 1 shows the implementation of
a simple, low-cost watt-hour meter using the AD7751.
Two current transformers (CTs) are used to provide the
current-to-voltage conversion needed by the AD7751,
and a simple divider network attenuates the line voltage.
The energy register (kWhrs) is a simple electro­mechanical counter that uses a two-phase stepper motor.
The AD7751 provides direct drive capability for this type
of counter. The AD7751 also provides a high-frequency
output at the CF pin for the meter constant (e.g.,
3200 imp/kWhr). Thus a high-frequency output is available
at the LED and opto-isolator output. This high-frequency
output is used to speed up the calibration process and
provides a means of quickly verifying meter functionality

1

< I < I

MAX

< I < 0.2 Ib 0.5 Lag ±1.5% ±2.5%

MAX

are 400% to 600% of Ib. Table I

MAX

2

PF

1 ±1.0% ±2.0%

0.8 Lead ±1.5% —

0.5 Lag ±1.0% ±2.0%

0.8 Lead ±1.0% —

energy registered by meter true energy

Percentage Error Limits
Class 1 Class 2

true energy

—see Table I.

MAX

–

100%

×

3

REV. 0

AN-563

LOAD

FG0003
1:1800

1:1800
FG0003

PHASE

NEUTRAL

240V

K1

R5

8.2

K2

K3

R6

9.1

SOLDER JUMPERS

R7
110

K4

J16 J17 J18 J19 J20 J21

R8

330

620

J5

J4

J3

J2

J1

CALIBRATION

NETWORK 2.5%

R9 R10 R11 R12 R13

1.2k 2.4k 4.7k 9.1k

CALIBRATION

NETWORK 30%

R18
18k

R17
39k

R16
75k

R15
150k

R14
300k

K3

K4

560

1.2k

2.2k

5.1k

9.1k

R24

300k

R23

R22

R21

R20

R19

Figure 1. Simple Single-Phase Watt-Hour Meter Based on the AD7751

R25

300k

J10

J9

J8

J7

J6

Z1

C17

10nF

FAULT-RESISTANT ENERGY METER
= 40A, Ib = 10A, CLASS 1 (5%Ib TO I

I

MAX

V

CLKOUT

D2

DD

P1

F1

F2

CF

REVP

FAULT

CLKIN

G0

G1

S0

S1

SCF

DGND

POWER SUPPLY

C19

+

470F
35V

8

P24

P23

P22

P20

P19

P18

Y1

P17

P16

P15

P14

P13

P12

C13
100nF

R29

820

3.579545MHz
C10

22pF

C9
22pF

G0 = 1
G1 = 0
S0 = 1
S1 = 0
SCF = 0

1

U2

7805

2,3,6,7

C12

+

C1

33nF

C2

33nF

C3

33nF

C4

33nF

C5

33nF

C7

100nF

C18

470nF

MOV1
S20K275

C11

100nF

P10

1N4744A

220F

6.3V

R1

1k

R2

1k

R3

1k

R4A

887

R4B

887

+

C6

10F

X2

R33
10

P3 P2

AVDD AC/DC DVDD

P4

V1A

AD7751

P6

V1N

P5

V1B

P7

V2N

P8

V2P

REF

IN/OUT

AGND

RESET

P9 P11 P21

R27

10k

V

DD

R26

470

1N4004

D2

)

MAX

C14

+

10F

6.3V

HP HLMP-D150

V

DD

V

DD

5V

R30

820

R28
10k

J15

J14

J13

J12

J11

C8
100nF

Z2

TO IMPULSE COUNTER/
STEPPER MOTOR

R32
20

R31
20

D1

FAULT
LED

D4

2

1

U3

PS2501-1

0R

K7

100IMP/kWhr

C16

C15

K8

CALIBRATION
LED
HP HLMP-D150

4

3200IMP/kWhr

3

JUMPERS USE

0 RESISTORS

0R

K9

K10

and accuracy in a production environment. The meter is
calibrated in a two-step process:

Step 1. With current passing through only Channel V1A's
CT, the meter is first calibrated by varying the line voltage
attenuation using the resistor network R14 to R23.

Step 2. With current passing through only Channel
V1B's CT, the small gain mismatch between the CTs in
Channel V1A and V1B is calibrated by shorting the
appropriate resistors in the resistor network R8 to R13.

DESIGN EQUATIONS

The AD7751 produces an output frequency proportional
to the time average value of the product of two voltage
signals. The input voltage signals are applied at V1 and
V2. The detailed functionality of the AD7751 is explained
in the AD7751 data sheet, see

Theory Of Operation

. The
AD7751 data sheet also provides an equation that relates
the output frequency on F1 and F2 (counter drive) to the
product of the rms signal levels at V1 and V2. This equa­tion is shown here again for convenience and will be

used to determine the correct signal scaling at V2 in
order to calibrate the meter to a fixed constant.

Frequency

5.74 1 2

V V Gain F

×× × ×

=

2

V

REF

−

14

The meter shown in Figure 1 is designed to operate at a
line voltage of 240 V and a maximum current (I
40 A. However, by correctly scaling the signals on Chan­nel 1 and Channel 2, a meter operating from any line
voltage and maximum current could be designed.

The four frequency options available on the AD7751 will
allow similar meters (i.e., direct counter drive) with an
I

of up to 120 A to be designed. The basic current for

MAX

this meter is selected as 10 A and the current range for
accuracy will be 1% Ib to I

or a dynamic range of 400

MAX

(100 mA to 40 A). The electromechanical register (kWh)
will have a constant of 100 imp/kWh, i.e., 100 impulses
from the AD7751 will be required in order to register
1 kWhr. IEC1036 Section 4.2.11 specifies that electro­magnetic registers have their lowest values numbered
in ten division, each division being subdivided into
ten parts.

–2–

(1)

) of

MAX

REV. 0

AN-563

Hence a display with five-plus-one digits is used, i.e.,
10,000s, 1,000s, 100s, 10s, 1s, 1/10s. The meter constant
(for calibration and test) is selected as 3200 imp/kWh.
The on-chip reference circuit of the AD7751 typically has
a temperature coefficient of 30 ppm/°C. However, on A
grade parts this specification is not guaranteed and may
be as high as 80 ppm /°C. At 80 ppm /°C the AD7751 error
at –20°C/+60°C would be approximately 0.65%, assum­ing a calibration at 25°C.

Current Transformer (CT) Selection

The CTs and their burden resistors should be selected to
maximize the use of the dynamic range on Channel V1A
and V1B (current channel). However there are some
important considerations when selecting the CTs and
the burden resistors for energy metering application.
Firstly, one need to select CTs that have good linearity in
both their gain and phase characteristics over the range
of current specified in the accuracy requirement. For
IEC1036, the range is between 5% Ib to I

. CT manu-

MAX

facturers often recommend the burden resistance to be
as small as possible to preserve linearity over large current
range. A burden resistance of less than 15 Ω is recom­mended. Secondly, CT introduces a phase shift between
primary and secondary current. The phase shift can con­tribute to a significant error at low-power factor. Note
that at power factor of 0.5, a phase shift as small as 0.1°
translates to 0.3% error in the energy reading. In this
design, the phase of the voltage channel (V2) is shifted
to match the phase shift introduced by the CT to elimi­nate any phase mismatch between the current and
voltage channel. This is achieved by moving the corner
frequency of the antialiasing filter in the voltage channel
input, see
and

Corrected Phase Matching between Channels

Antialias Filters

in this application note.

Design Calculations

Design Parameters:

Line Voltage = 240 V (Nominal)
I

= 40 A (Ib = 10 A)

MAX

Counter = 100 imp/kWh
Meter Constant = 3200 imp/kWh
CT Turn Ratio = 1:1800
Size of Burden Resistor (Channel 1 A) = 8.2 Ω

100 imp/hour = 100/3600 sec. = 0.027777 Hz
Meter Will Be Calibrated at Ib (10 A)
Power Dissipation at Ib = 240 V × 10 A = 2.4 kW
Frequency on F1 (and F2) at Ib = 2.4 × 0.027777 Hz
= 0.06666667 Hz

Voltage across CT at Ib (V1A) = 10 A/1800 × 8.2 Ω =

45.6 mV.

The gain setting is determined by the signal in V1 (cur­rent channel). At I

= 40 A, the rms voltage at V1 is

MAX

40 A/1800 × 8.2 Ω = 182 mV. It translates to a peak voltage
of 258 mV. From Table I of the AD7751 data sheet, it can
be seen that the gain of two provides the best utilization

of the dynamic range (±330 mV). The setting also pro­vides more than 20% headroom in the event of surge in
the current.

To select the F
data sheet,

Application

frequency for Equation 1 see the AD7751

1-4

Selecting a Frequency for an Energy Meter

. From Tables V and VI in the AD7751 data
sheet, it can be seen that the best choice of frequency for
a meter with I

= 40 A is 3.4 Hz (F2). This frequency

MAX

selection is made by the logic inputs S0 and S1, see
Table II in the AD7751 data sheet. The CF frequency
selection (meter constant) is selected by using the
logic input SCF. The two available options are 64 × F1
(6400 imp/kWh) or 32 × F1 (3200 imp/kWh). For this design,
3200 imp/kWh is selected by setting SCF logic low. With
a meter constant of 3200 imp/kWh and a maximum cur­rent of 40 A, the maximum frequency from CF is 8.53 Hz.
Many calibration benches used to verify meter accuracy
still use optical techniques. This limits the maximum fre­quency which can be reliably read to about 10 Hz. The
only remaining unknown from Equation 1 is V2 or the
signal level on Channel 2 (the voltage channel).

From Equation 1 on the previous page:

0 0666667

.

=

25

.

2

mV V Hz

××××

5 74 45 56 2 2 3 4

.. .

Hz

Where: V2 = 234.3 mV rms.

Therefore, in order to calibrate the meter, the line volt­age needs to be attenuated down to 234.3 mV.

CALIBRATING THE METER: VOLTAGE CHANNEL
CALIBRATION

From the previous section it can be seen that the meter
is simply calibrated by attenuating the line voltage down
to 234.3 mV. The line voltage attenuation is carried out
by a simple resistor divider as shown in Figure 2. The
attenuation network should allow a calibration range of
at least ±30% to allow for CT/burden resistance toler­ances and the on-chip reference tolerance of ±8%, see
the AD7751 data sheet. In addition, the topology of the
network is such that the phase matching between
Channel 1 and Channel 2 is preserved, even when the
attenuation is being adjusted, see

ing between Channels

R9

J5

J4

R8

R7

J3

R6

J2

J1

R5

in this application note.

R14

R13

R12

R11

R10

J10

J9

J8

J7

J6

Correct Phase Match-

234.3mV

R4B

R5 + R6 + ........... + R15 + R16 >> R4B

f

–3dB

C5

1/(2    R4B  C5)

R15 R16

240V

Figure 2. Attenuation Network for Calibrating the
Voltage Channel (V2)

REV. 0

–3–

AN-563

As can be seen from Figure 2, the –3 dB frequency of this
network is determined by R4B and C5. Even with all the
jumpers closed, the resistance of R15 (300 kΩ) and R16
(300 kΩ) is still much greater than R4B (887 Ω). Hence
varying the resistance of the resistor chain R5 to R14 will
have little effect on the –3 dB frequency of the network.
The network shown in Figure 2 allows the line voltage to
be attenuated and adjusted in the range 170 mV to
399 mV with a resolution of 10 bits or 223 µV. This is
achieved by using the binary weighted resister chain
R14 to R23. This will allow the meter to be accurately
calibrated using a successive approximation technique.

Starting with J1 each jumper is closed in order of
ascendance, e.g., J1, J2, J3, etc. If the calibration fre­quency on CF, i.e., 32 × 100 imp/KWh (at Ib = 10 A, CF is
expected to be 2.133 Hz) is exceeded when any jumper
is closed, it should be opened again. All jumpers are
tested, J10 being the last jumper. Note that jumper con­nections are made with soldering together the jumper
pins across the resistors in the network. This approach is
preferred over the use of trim pots, as the stability of
the latter over time and environmental conditions is
questionable.

Since the AD7751 transfer function is extremely linear, a
one-point calibration (at Ib) at unity power factor is all
that is needed to calibrate the meter. If the correct pre­cautions have been taken at the design stage no
calibration will be necessary at low-power factor (e.g.,
PF = 0.5).

CALIBRATING THE METER: MATCHING THE TWO
CURRENT SENSOR INPUTS

A calibration network consisting of six parallel resistors
is used to compensate gain variation between the two
CTs used to monitor the phase and neutral currents.
However, such mismatch is often small and needs to be
compensated with a more accurate calibration network.
In this design, six resistors are used for this purpose.
The primary burden resistors for V1B, R6, and R7, com­bined to a 8.4 Ω burden (9.1 Ω储110 Ω = 8.4 Ω). This is
about 2.5% above the nominal burden used in V1A. The
burden is reduced by connecting the jumpers from J16
to J21. This adds more resistors to be in parallel to the
burden, thus reducing the total resistance between the
two terminals of the CT. The values of R8 to R13 are cho­sen carefully so that the resulting resistance values
spread out evenly across the calibration range. Closing
all jumpers, J16 to J21, represents the lower bound for
the calibration range. In our design, the lower bound is
at approximately 7.99 Ω, or 2.5% lower than the nominal
burden of V1A.

Starting from J16, each jumper is closed in order of
ascendance, e.g., J16, J17, J18, etc. If the calibration fre­quency on CF becomes smaller than the expected value
(at Ib = 10 A, CF = 2.133 Hz) after a jumper is closed, the

jumper should be opened again. All jumpers are tested,
J21 being the last jumper.

FG0003
1:1800

J17

J18

J19

J20

R12

J21

R13R7

J16

R8

R9

R10

R6

R11

Figure 3. Calibration Network for V1B

CORRECT PHASE-MATCHING BETWEEN CHANNELS

The AD7751 is internally phase-matched over the fre­quency range 40 Hz to 1 kHz. Correct phase-matching is
important in an energy metering application because
any phase mismatch between channels will translate
into significant errors at low-power factor. This is easily
illustrated with the following example. Figure 4 shows
the voltage and current waveforms for an inductive
load. In the example shown the current lags the voltage
by 60° (PF = 0.5). Assuming pure sinusoidal conditions
the power is easily calculated as V rms × I rms × cos
(60°).

PF = 1

V.I

2

PF = 0.5

V.I COS(60)

2

CURRENT

VOLTAGE

CURRENT

INSTANTANEOUS
POWER SIGNAL

VOLTAGE

INSTANTANEOUS
POWER SIGNAL

60

INSTANTANEOUS REAL
POWER SIGNAL

INSTANTANEOUS REAL
POWER SIGNAL

Figure 4. Voltage and Current (Inductive Load)

If, however, a phase error (φe) is introduced externally to
the AD7751, e.g., in the antialias filters, the error is cal­culated as:

[cos(δ°) – cos(δ°+φ

See

Note 3 in Table I.

voltage and current and φ

)]/cos(δ°) × 100% (2)

e

Where δ is the phase angle between

is the external phase error.

e

With a phase error of 0.2°, for example, the error at
PF = 0.5 (60°) is calculated as 0.6%. As this example
demonstrates, even a very small phase error will pro­duce a large measurement error at low-power factor.

–4–

R3

V1B

C3

REV. 0

AN-563

The current sensor has an intrinsic phase shift of 0.1°. If
it is not compensated, it can introduce a significant error
at low-power factor. In this design, the phase is com­pensated by introducing a 0.1° phase shift in the voltage
channel to ensure both the current and voltage inputs
are phase matched. This is easily achieved by reduc­ing the resistance in the antialiasing filter in the
voltage channel.

Antialias Filters

As mentioned in the previous section, one possible
source of external phase errors are the antialias filters
on Channel 1 and Channel 2. The antialias filters are low­pass filters that are placed before the analog inputs of
any ADC. They are required in order to prevent a pos­sible distortion due to sampling called aliasing. Figure 5
illustrates the effects of aliasing.

ALIASING EFFECT

IMAGE

FREQUENCIES

0

–20

dB

–40

–60

1k10010

FREQUENCY – Hz

10k

100k 1M

Figure 6. RC Filter Magnitude Response

0

–20

–40

DEGREES

–60

0

2

450

FREQUENCY – kHz

900

Figure 5. Aliasing Effects

Figure 5 shows how aliasing effects could introduce
inaccuracies in an AD7751-based meter design. The
AD7751 uses two Σ-∆ ADCs to digitize the voltage and
current signals. These ADCs have a very high sampling
rate, i.e., 900 kHz. Figure 5 shows how frequency com­ponents (arrows shown in black) above half the
sampling frequency (also known as the Nyquist fre­quency), i.e., 450 kHz are imaged or folded back down
below 450 kHz (arrows shown in grey). This will happen
with all ADCs no matter what the architecture is. In the
example shown it can be seen that only frequencies
near the sampling frequency, i.e., 900 kHz, will move
into the band of interest for metering, i.e., 0 kHz–2 kHz.
This fact will allow us to use a very simple LPF (Low­Pass Filter) to attenuate these high frequencies (near
900 kHz) and so prevent distortion in the band of interest.

The simplest form of LPF is the simple RC filter. This is a
single-pole filter with a roll-off or attenuation of –20 dB/dec.

CHOOSING THE FILTER –3 dB FREQUENCY

As well as having a magnitude response, all filters
also have a phase response. The magnitude and
phase response of a simple RC filter (R = 1 kΩ, C = 33 nF)
are shown in Figures 6 and 7. From Figure 6 it is seen
that the attenuation at 900 kHz for this simple LPF is
greater than 40 dBs. This is enough attenuation to ensure
no ill effects due to aliasing.

–80

–100

1k10010

FREQUENCY – Hz

10k

100k 1M

Figure 7. RC Filter Phase Response

As explained in the last section, the phase response can
introduce significant errors if the phase response of the
LPFs on both Channel 1 and Channel 2 are not matched.
Phase mismatch can easily occur due to poor compo­nent tolerances in the LPF. The lower the –3 dB
frequency in the LPF (antialias filter) the more pro­nounced these errors will be at the fundamental
frequency component or the line frequency. Even with
the corner frequency set at 4.8 kHz (R = 1 kΩ, C = 33 nF)
the phase errors due to poor component tolerances can
be significant. Figure 8 illustrates the point. In Figure 8,
the phase response for the simple LPF is shown at 50 Hz
for R = 1 kΩ ± 10%, C = 33 nF ± 10%. Remember a phase
shift of 0.1°–0.2° can cause measurement errors of 0.6%
at low-power factor. This design uses resistors of 1% tol­erance and capacitors of 10% tolerance for the antialias
filters to reduce the possible problems due to phase
mismatch. Alternatively the corner frequency of the
antialias filter could be pushed out to 10 kHz–15 Hz.
However, the corner frequency should not be made too
high. This could allow enough high-frequency compo­nents to be aliased and cause accuracy problems in a
noisy environment.

REV. 0

–5–

AN-563

–0.4

–0.5

–0.6

DEGREES

–0.7

–0.8

45

FREQUENCY – Hz

(50Hz, –0.481)
(R = 900, C = 29.7nF)

(50Hz, –0.594)
(R = 1k, C = 33nF)

(50Hz, –0.718)
(R = 1.1k, C = 36.3nF)

50

55

Figure 8. Phase Shift at 50 Hz Due to Component
Tolerances

Note this is also why precautions were taken with the
design of the calibration network on Channel 2 (voltage
channel). Calibrating the meter by varying the resis­tance of the attenuation network will not vary the –3 dB
frequency and hence the phase response of the network
on Channel 2, see

nel Calibration

Calibrating the Meter: Voltage Chan-

. Shown in Figure 9 is a plot of phase
lag at 50 Hz when the resistance of the calibration net­work is varied from 600 kΩ (J1–J10 closed) to 1.2 MΩ
(J1–J10 open).

when no load is connected. IEC 1036 (1996–09) Section

4.6.4 specifies the start-up current as being not more
than 0.4% Ib at PF = 1. For this design the start current is
calculated at 7.14 mA or 0.07% Ib, see

old

, AD7751 data sheet.

No Load Thresh-

POWER SUPPLY DESIGN

This design uses a simple low-cost power supply based
on a capacitor divider network, i.e., C18 and C19. Most of
the line voltage is dropped across C18, a 470 nF, 250 V
metalized polyester film capacitor. The impedance of
C18 dictates the effective VA rating of the supply. How­ever the size of C18 is constrained by the power
consumption specification in IEC1036. The total power
consumption in the voltage circuit including power sup­ply is specified in Section 4.4.1.1 of IEC1036 (1996–9).
The total power consumption in each phase is 2 W and
10 VA under nominal conditions. The nominal VA rating
of the supply in this design is 8.5 VA. The total power
dissipation is approximately 0.59 W. Figure 10 shows
the basic power supply design.

240V

C18

R26

V1

D2

+

C19

D3

U2

7805

5V

V2

V

DD

–0.591

–0.592

–0.593

DEGREES

–0.594

–0.595

J1–J10 OPEN
(50Hz, –0.59348)

49.9 50.0
FREQUENCY – Hz

J1–J10 CLOSED
(50Hz, –0.59299)

50.1

Figure 9. Phase Shift Due to Calibration

For the resistor network used for matching the CTs in
V1A and V1B, the calibration network is based on alter­ing the burden resistance. High-precision components
are used for the RC filters in V1A and V1B. Any mis­match will be caused by the component tolerance.
Under normal operation, this mismatch is well within
the 12.5% threshold needed to cause AD7751 to indicate
a FAULT condition and perform current channel switching.

Figure 10. Power Supply

The plots shown in Figures 11, 12, 13, and 14 show the
PSU performance under heavy load (50 A) with the line
voltage varied from 180 V to 250 V. By far the biggest
load on the power supply is the current required to drive
the stepper motor which has a coil impedance of about
400 Ω. This is clearly seen by looking at V1 (voltage on
C19) in the plots below. Figure 11 shows the current
drawn from the supply. Refer to Figure 10 when review­ing the simulation plots below.

15

10

VOLTS

5

V1
(C19)

V2
(V

DD

C19 VOLTAGE DROP DUE TO

)

STEPPER MOTOR DRIVE

NO LOAD THRESHOLD

The AD7751 has on-chip anticreep functionality. The
AD7751 will not produce a pulse on CF, F1, or F2 if the
output frequency falls below a certain level. This feature
ensures that the energy meter will not register energy

0

0

2

4

68

TIME – s

Figure 11. Power Supply Voltage Output at 220 V and
50 A Load

–6–

10

REV. 0