Analog Devices AN559 Application Notes

AN-559

a

APPLICATION NOTE

One Technology Way • P.O. Box 9106 • Norwood, MA 02062-9106 • 781/329-4700 • World Wide Web Site: http://www.analog.com

A Low Cost Watt-Hour Energy Meter Based on the AD7755

By Anthony Collins

INTRODUCTION

This application note describes a low-cost, high-accuracy
watt-hour meter based on the AD7755. The meter
described is intended for use in single phase, two­wire distribution systems. However the design can easily
be adapted to suit specific regional requirements, e.g., in
the United States power is usually distributed to residential
customers as single-phase, three-wire.

The AD7755 is a low-cost, single-chip solution for elec­trical energy measurement. The AD7755 is comprised of
two ADCs, reference circuit, and all the signal process­ing necessary for the calculation of real (active) power.
The AD7755 also includes direct drive capability for elec­tromechanical counters (i.e., the energy register), and
has a high-frequency pulse output for calibration and
communications purposes.

This application note should be used in conjunction with
the AD7755 data sheet. The data sheet provides detailed
information on the functionality of the AD7755 and will be
referenced several times in this application note.

DESIGN GOALS

The international Standard IEC1036 (1996-09) –

Alternat­ing Current Watt-Hour Meters for Active Energy (Classes
1 and 2)

, was used as the primary specification for this
design. For readers more familiar with the ANSI C12.16
specification, see the section at the end of this applica­tion note, which compares the IEC1036 and ANSI C12.16
standards. This section explains the key IEC1036 specifi­cations in terms of their ANSI equivalents.

The design greatly exceeds this basic specification for
many of the accuracy requirements, e.g., accuracy at unity­power factor and at low (PF = ±0.5) power factor. In addi­tion, the dynamic range performance of the meter has
been extended to 500. The IEC1036 standard specifies
accuracy over a range of 5% Ib to I
values for I

are 400% to 600% of Ib. Table I outlines the

MAX

—see Table I. Typical

MAX

accuracy requirements for a static watt-hour meter. The
current range (dynamic range) for accuracy is specified in
terms of Ib (basic current).

Current Value

0.05 Ib < I < 0.1 Ib 1 ±1.5% ±2.5%

< I < I

0.1 Ib

0.1 Ib

< I < 0.2 Ib 0.5 Lag ±1.5% ±2.5%

< I < I

0.2 Ib

NOTES

1

The current ranges for specified accuracy shown in Table I are expressed
in terms of the basic current (Ib). The basic current is defined in IEC1036
(1996–09) section 3.5.1.1 as the value of current in accordance with
which the relevant performance of a direct connection meter is fixed.
I

is the maximum current at which accuracy is maintained.

MAX

2

Power Factor (PF) in Table I relates the phase relationship between the
fundamental (45 Hz to 65 Hz) voltage and current waveforms. PF in this
case can be simply defined as PF = cos( φ), where φ is the phase angle
between pure sinusoidal current and voltage.

3

Class index is defined in IEC1036 (1996–09) section 3.5.5 as the limits of
the permissible percentage error. The percentage error is defined as:

Percentage Error

The schematic in Figure 1 shows the implementation of
a simple, low-cost watt-hour meter using the AD7755. A
shunt is used to provide the current-to-voltage conver­sion needed by the AD7755 and a simple divider
network attenuates the line voltage. The energy register
(kWh) is a simple electromechanical counter that uses a
two-phase stepper motor. The AD7755 provides direct
drive capability for this type of counter. The AD7755 also
provides a high-frequency output at the CF pin for the
meter constant (3200 imp/kWh). Thus a high-frequency
output is available at the LED and opto-isolator output.
This high-frequency output is used to speed up the
calibration process and provides a means of quickly
verifying meter functionality and accuracy in a production
environment. The meter is calibrated by varying the line
voltage attenuation using the resistor network R5 to R14.

Table I. Accuracy Requirements

3

1

MAX

Percentage Error Limits
Class 1 Class 2

PF

2

1 ±1.0% ±2.0%

0.8 Lead ±1.5%

MAX

0.5 Lag ±1.0% ±2.0%

0.8 Lead ±1.0%

Energy

=×

Registered by Meter – True Energy

True Energy

100%

REV. A

AN-559

NEUTRAL

LOAD

PHASE

V

U1

REVP

NC

CLKOUT

CLKIN

SCF

DGND

POWER SUPPLY

8

U2

7805

2,3,6,7

DD

C12

C13

P1

P24

F1

P23

F2

P22

CF

P20

P19

P18

Y1

P17

G0

P16

G1

P15

S0

P14

S1

P13

P12

NC = NO CONNECT

V

DD

1

5V

+

V

C9

C8

Z2

DD

R17

J15

J14

J13

J12

J11

C7

TO IMPULSE COUNTER/
STEPPER MOTOR

R20

R19

R18

CALIBRATION

D1

LED

1

2

U3

PS2501-1

JUMPERS USE
0 RESISTOR

C15

C14

4

3

K5

K6

100 imp/kWhr

K7

3200 imp/kWhr

K8

CALIBRATION
NETWORK

R9

R14

R8

R13

R7

R12

R6

R11

R5

R10

K3

K4

Z3

Z4

J10

J9

C19

J8

J7

J6

R15 R16

AD780

Z1

C16

V

U4

4

K1

–

350

+

K2

JUMPERS USE
0 RESISTOR

J5

J4

J3

J2

J1

DD

2

6

+

C11

R1

R2

R3

R4

+

C5 C6

EXTERNAL
REFERENCE
(OPTIONAL)

C17

MOV1

C10

P3 P2

AVDD AC/DC DVDD

P4

NC

P5

V1P

C1

P6

V1N

C2

P7

V2N

C3

P8

V2P

C4

P10

REF

IN/OUT

RESET

P9 P11 P21
R23

V

DD

R21 D2

D3

C18

R22

AD7755

AGND

+

220V

Figure 1. Simple Single-Phase Watt-Hour Meter Based on the AD7755

DESIGN EQUATIONS

The AD7755 produces an output frequency that is pro­portional to the time average value of the product of two
voltage signals. The input voltage signals are applied at
V1 and V2. The detailed functionality of the AD7755 is
explained in the AD7755 data sheet, Theory Of Opera­tion section. The AD7755 data sheet also provides an
equation that relates the output frequency on F1 and F2
(counter drive) to the product of the rms signal levels
at V1 and V2. This equation is shown here again for
convenience and will be used to determine the correct
signal scaling at V2 in order to calibrate the meter to a
fixed constant.

Frequency

=

×× × ×8.06 1

2

2

V

REF

1–4

(1)

V V Gain F

The meter shown in Figure 1 is designed to operate at
a line voltage of 220 V and a maximum current (I

MAX

) of

40 A. However, by correctly scaling the signals on
Channel 1 and Channel 2, a meter operating of any line
voltage and maximum current could be designed.

The four frequency options available on the AD7755 will
allow similar meters (i.e., direct counter drive) with an
I

of up to 120 A to be designed. The basic current (Ib)

MAX

for this meter is selected as 5 A and the current range for
accuracy will be 2% Ib to I

, or a dynamic range of 400

MAX

(100 mA to 40 A). The electromechanical register (kWh)
will have a constant of 100 imp/kWh, i.e., 100 impulses
from the AD7755 will be required in order to register
1 kWh. IEC1036 section 4.2.11 specifies that electromag­netic registers have their lowest values numbered in ten
division, each division being subdivided into ten parts.
Hence a display with a five plus one digits is used, i.e.,
10,000s, 1,000s, 100s, 10s, 1s, 1/10s. The meter constant
(for calibration and test) is selected as 3200 imp/kWh.

–2–

REV. A

AN-559

Design Calculations

Design parameters:

Line voltage = 220 V (nominal)
I

= 40 A (Ib = 5 A)

MAX

Counter = 100 imp/kWh
Meter constant = 3200 imp/kWh
Shunt size = 350 µΩ
100 imp/hour = 100/3600 sec = 0.027777 Hz
Meter will be calibrated at Ib (5A)
Power dissipation at Ib = 220 V × 5 A = 1.1 kW
Frequency on F1 (and F2) at Ib = 1.1 × 0.027777 Hz
= 0.0305555 Hz

Voltage across shunt (V1) at Ib = 5 A × 350 µΩ = 1.75 mV.

Figure 2. Final Implementation of the AD7755 Meter

AD7755 Reference

The schematic in Figure 1 also shows an optional reference
circuit. The on-chip reference circuit of the AD7755 has a
temperature coefficient of typically 30 ppm/°C. However,
on A Grade parts this specification is not guaranteed
and may be as high as 80 ppm/°C. At 80 ppm/°C the
AD7755 error at –20°C/+60°C could be as high as 0.65%,
assuming a calibration at 25°C.

Shunt Selection

The shunt size (350 µΩ) is selected to maximize the use
of the dynamic range on Channel V1 (current channel).
However there are some important considerations when
selecting a shunt for an energy metering application.
First, minimize the power dissipation in the shunt. The
maximum rated current for this design is 40 A, therefore,
the maximum power dissipated in the shunt is (40 A)
350 µΩ = 560 mW. IEC1036 calls for a maximum power
dissipation of 2 W (including power supply). Secondly,
the higher power dissipation may make it difficult to
manage the thermal issues. Although the shunt is
manufactured from Manganin material, which is an
alloy with a low temperature coefficient of resistance,
high temperatures may cause significant error at heavy
loads. A third consideration is the ability of the meter to
resist attempts to tamper by shorting the phase circuit.
With a very low value of shunt resistance the effects of
externally shorting the shunt are very much minimized.
Therefore, the shunt should always be made as small as
possible, but this must be offset against the signal range
on V1 (0 mV–20 mV rms with a gain of 16). If the shunt is
made too small it will not be possible to meet the
IEC1036 accuracy requirements at light loads. A shunt
value of 350 µΩ was considered a good compromise for
this design.

2

×

To select the F
data sheet, Selecting a Frequency for an Energy Meter
Application section. From Tables V and VI in the AD7755
data sheet it can be seen that the best choice of fre­quency for a meter with I
frequency selection is made by the logic inputs S0 and
S1—see Table II in the AD7755 data sheet. The CF fre­quency selection (meter constant) is selected by using
the logic input SCF. The two available options are 64 
F1(6400 imp/kWh) or 32 × F1(3200 imp/kWh). For this
design, 3200 imp/kWh is selected by setting SCF logic
low. With a meter constant of 3200 imp/kWh and a
maximum current of 40 A, the maximum frequency
from CF is 7.82 Hz. Many calibration benches used to
verify meter accuracy still use optical techniques. This
limits the maximum frequency that can be reliably read
to about 10 Hz. The only remaining unknown from
equation 1 is V2 or the signal level on Channel 2 (the
voltage channel).

From Equation 1 on the previous page:

0 030555

.

Therefore, in order to calibrate the meter the line volt­age needs to be attenuated down to 248.9 mV.

CALIBRATING THE METER

From the previous section it can be seen that the meter
is simply calibrated by attenuating the line voltage down
to 248.9 mV. The line voltage attenuation is carried out
by a simple resistor divider as shown in Figure 3. The
attenuation network should allow a calibration range of
at least ±30% to allow for shunt tolerances and the on-chip
reference tolerance of ±8%—see AD7755 data sheet.
In addition, the topology of the network is such that the
phase-matching between Channel 1 and Channel 2 is
preserved, even when the attenuation is being adjusted
(see Correct Phase Matching Between Channels section).

frequency for Equation 1 see the AD7755

1–4

= 40 A is 3.4 Hz (F2). This

MAX

8 06 1 75 2 16 3 4

Hz

.. .

=

V2 = 248.9 mV rms

mV V Hz

××××

25

.

2

REV. A

–3–

AN-559

248.9mV

R9

J5

J4

J3

J2

J1

R14

R8

R13

R7

R12

R6

R11

R5

R10

J10

J9

J8

J7

J6

R15 R16

R4 C4

R5 + R6 + ............. + R15 + R16 >> R4

1/(2..R4.C4)

f

–3dB

220V

Figure 3. Attenuation Network

As can be seen from Figure 3, the –3 dB frequency of this
network is determined by R4 and C4. Even with all the
jumpers closed, the resistance of R15 (330 kΩ) and R16
(330 kΩ) is still much greater than R4 (1 kΩ). Hence vary­ing the resistance of the resistor chain R5 to R14 will
have little effect on the –3 dB frequency of the network.
The network shown in Figure 3 allows the line voltage
to be attenuated and adjusted in the range 175 mV to
333 mV with a resolution of 10 bits or 154 µV. This is
achieved by using the binary weighted resistor chain R5
to R14. This will allow the meter to be accurately cali­brated using a successive approximation technique.
Starting with J1, each jumper is closed in order of
ascendance, e.g., J1, J2, J3, etc. If the calibration fre­quency on CF, i.e., 32 × 100 imp/hr (0.9777 Hz), is
exceeded when any jumper is closed, it should be
opened again. All jumpers are tested, J10 being the last
jumper. Note jumper connections are made with 0 Ω
fixed resistors which are soldered into place. This approach
is preferred over the use of trim pots, as the stability of
the latter over time and environmental conditions is
questionable.

Since the AD7755 transfer function is extremely linear a
one-point calibration (Ib) at unity power factor, is all that
is needed to calibrate the meter. If the correct precau­tions have been taken at the design stage, no calibration
will be necessary at low-power factor (PF = 0.5). The
next section discusses phase matching for correct calcu­lation of energy at low-power factor.

If, however, a phase error (φ

) is introduced externally to

e

the AD7755, e.g., in the antialias filters, the error is cal­culated as:

[cos(δ°) – cos(δ°+ φ

)]/cos(δ°) × 100% (2)

e

See Note 3 on Table I. Where δ is the phase angle between
voltage and current and φ

is the external phase error.

e

With a phase error of 0.2°, for example, the error at
PF = 0.5 (60°) is calculated as 0.6%. As this example
demonstrates, even a very small phase error will pro­duce a large measurement error at low power factor.

PF = 1

V.I

PF = 0.5

V.I COS(60)

2

2

CURRENT

VOLTAGE

VOLTAGE

CURRENT

INSTANTANEOUS
POWER SIGNAL

INSTANTANEOUS
POWER SIGNAL

60

INSTANTANEOUS REAL
POWER SIGNAL

INSTANTANEOUS REAL
POWER SIGNAL

Figure 4. Voltage and Current (Inductive Load)

ANTIALIAS FILTERS

As mentioned in the previous section, one possible
source of external phase errors are the antialias filters
on Channel 1 and Channel 2. The antialias filters are low­pass filters that are placed before the analog inputs of any
ADC. They are required to prevent a possible distortion
due to sampling called aliasing. Figure 5 illustrates the
effects of aliasing.

CORRECT PHASE MATCHING BETWEEN CHANNELS

The AD7755 is internally phase-matched over the fre­quency range 40 Hz to 1 kHz. Correct phase matching is
important in an energy metering application because
any phase mismatch between channels will translate
into significant measurement error at low-power fac­tor. This is easily illustrated with the following example.
Figure 4 shows the voltage and current waveforms for
an inductive load. In the example shown, the current
lags the voltage by 60° (PF = –0.5). Assuming pure sinu­soidal conditions, the power is easily calculated as
V rms × I rms × cos (60°).

–4–

0

IMAGE

FREQUENCIES

2

Figure 5. Aliasing Effects

450

FREQUENCY – kHz

900

REV. A

AN-559

Figure 5 shows how aliasing effects could introduce inac­curacies in an AD7755-based meter design. The AD7755
uses two Σ-∆ ADCs to digitize the voltage and current
signals. These ADCs have a very high sampling rate, i.e.,
900 kHz. Figure 5 shows how frequency components
(arrows shown in black) above half the sampling fre­quency (also know as the Nyquist frequency), i.e., 450 kHz
is imaged or folded back down below 450 kHz (arrows
shown dashed). This will happen with all ADCs no mat­ter what the architecture. In the example shown it can be
seen that only frequencies near the sampling frequency,
i.e., 900 kHz, will move into the band of interest for meter­ing, i.e., 0 kHz–2 kHz. This fact will allow us to use a very
simple LPF (Low-Pass Filter) to attenuate these high fre­quencies (near 900 kHz) and so prevent distortion in the
band of interest.

The simplest form of LPF is the simple RC filter. This is
a single-pole filter with a roll-off or attenuation of
–20 dBs/dec.

Choosing the Filter –3 dB Frequency

As well as having a magnitude response, all filters also
have a phase response. The magnitude and phase
response of a simple RC filter (R = 1 kΩ, C = 33 nF) are
shown in Figures 6 and 7. From Figure 6 it is seen that
the attenuation at 900 kHz for this simple LPF is greater
than 40 dBs. This is enough attenuation to ensure no ill
effects due to aliasing.

0

–20

dB

–40

0

–20

–40

DEGREES

–60

–80

–100

1k10010

FREQUENCY – Hz

10k

100k 1M

Figure 7. RC Filter Phase Response

As explained in the last section, the phase response can
introduce significant errors if the phase response of the
LPFs on both Channel 1 and Channel 2 are not matched.
Phase mismatch can easily occur due to poor component
tolerances in the LPF. The lower the –3 dB frequency in the
LPF (antialias filter) the more pronounced these errors
will be at the fundamental frequency component or the
line frequency. Even with the corner frequency set at

4.8 kHz (R = 1 kΩ, C = 33 nF) the phase errors due to poor
component tolerances can be significant. Figure 8 illus­trates the point. In Figure 8, the phase response for the
simple LPF is shown at 50 Hz for R = 1 kΩ ± 10%, C = 33 nF
± 10%. Remember a phase shift of 0.2° can causes mea­surement errors of 0.6% at low-power factor. This
design uses resistors of 1% tolerance and capacitors of
10% tolerance for the antialias filters to reduce the pos­sible problems due to phase mismatch. Alternatively the
corner frequency of the antialias filter could be pushed
out to 10 kHz–15 Hz. However, the corner frequency
should not be made too high, as this could allow enough
high-frequency components to be aliased and so cause
accuracy problems in a noisy environment.

REV. A

–60

1k10010

FREQUENCY – Hz

10k

100k 1M

Figure 6. RC Filter Magnitude Response

–5–

–0.4

–0.5

–0.6

DEGREES

–0.7

–0.8

45

FREQUENCY – Hz

(50Hz, –0.481)
(R = 900, C = 29.7nF)

(50Hz, –0.594)
(R = 1k, C = 33nF)

(50Hz, –0.718)
(R = 1.1k, C = 36.3nF)

50

55

Figure 8. Phase Shift at 50 Hz Due to Component
Tolerances

AN-559

L

SH2

L

GND

R

GND

GND

R

SH2

R

SH1

L

SH1

L

W2

R2

R1

L

W1

V1N

V1P

C2

C1

IN

OUT

PHASE

V1N

V1P

SHUNT 330

GND

FREQUENCY – Hz

10

–100

10k

100k 1M

1k

100

–50dB

–40dB

–30dB

–20dB

–10dB

0dB

–80

–60

–40

–20

–0

PHASE

MAGNITUDE

Note that this is also why precautions were taken with
the design of the calibration network on Channel 2 (volt­age channel). Calibrating the meter by varying the
resistance of the attenuation network will not vary the
–3 dB frequency and hence the phase response of the
network on Channel 2—see Calibrating the Meter sec­tion. Shown in Figure 9 is a plot of phase lag at 50 Hz
when the resistance of the calibration network is varied
from 660 kΩ (J1–J10 closed) to 1.26 MΩ (J1–J10 open).

–0.591

–0.592

–0.593

DEGREES

–0.594

–0.595

J1–J10 OPEN
(50Hz, –0.59348)

49.9 50.0
FREQUENCY – Hz

Figure 9. Phase Shift Due to Calibration

COMPENSATING FOR PARASITIC SHUNT INDUCTANCE

When used at low frequencies a shunt can be considered a
purely resistive element with no significant reactive ele­ments. However, under certain situations even a small
amount of stray inductance can cause undesirable effects
when a shunt is used in a practical data acquisition sys­tem. The problem is very noticeable when the resistance
of the shunt is very low, in the order of 200 µΩ. Shown
below is an equivalent circuit for the shunt used in the
AD7755 reference design. There are three connections
to the shunt. One pair of connections provides the cur­rent sense inputs (V1P and V1N) and the third connection
is the ground reference for the system.

J1–J10 CLOSED
(50Hz, –0.59308)

50.1

Figure 10. Equivalent Circuit for the Shunt

Canceling the Effects of the Parasitic Shunt Inductance

The effect of the parasitic shunt inductance is shown in
Figure 11. The plot shows the phase and magnitude
response of the antialias filter network with and without
(dashed) a parasitic inductance of 2 nH. As can be seen
from the plot, both the gain and phase response of the
network are affected. The attenuation at 1 MHz is now
only about –15 dB, which could cause some repeatabil­ity and accuracy problems in a noisy environment. More
importantly, a phase mismatch may now exist between
the current and voltage channels. Assuming the network
on Channel 2 has been designed to match the ideal
phase response of Channel 1, there now exists a phase
mismatch of 0.1° at 50 Hz. Note that 0.1° will cause a

0.3% measurement error at PF = ±0.5. See Equation 2
(Correct Phase Matching Between Channels).

The shunt resistance is shown as R

(350 µΩ). R

SH1

the resistance between the V1N input terminal and the
system ground reference point. The main parasitic ele­ments (inductance) are shown as L

SH1

and L
10 also shows how the shunt is connected to the AD7755
inputs (V1P and V1N) through the antialias filters. The
function of the antialias filters is explained in the previ­ous section and their ideal magnitude and phase
responses are shown in Figures 6 and 7.

. Figure

SH2

SH2

is

Figure 11. Effect of Parasitic Shunt Inductance on the
Antialias Network

The problem is caused by the addition of a zero into the
antialias network. Using the simple model for the shunt
shown in Figure 10, the location of the zero is given as
R

SH1/LSH1

radians.

One way to cancel the effects of this additional zero in
the network is to add an additional pole at (or close to)
the same location. The addition of an extra RC on each
analog input of Channel 1 will achieve the additional

–6–

REV. A

AN-559

FREQUENCY – Hz

10

–100

10k 100k 1M

1k

100

–50dB

–40dB

–30dB

–20dB

–10dB

0dB

–80

–60

–40

–20

–0

MAGNITUDE

PHASE

pole required. The new antialias network for Channel 1
is shown in Figure 12. To simplify the calculation and
demonstrate the principle, the Rs and Cs of the network
are assumed to have the same value.

j

POLE #1

POLE #2

3
2

RR

5

11

–



RC



4

RC

C



H (s) =

3

11

–

2

RC

ZERO #1

–(RSH/LSH)

2R2C2

S

+ S3RC + 1

+

1

5



4

RC

–

C

Figure 12. Shunt Inductance Compensation Network

Figure 12 also gives an expression for the location of the
poles of this compensation network. The purpose of
Pole #1 is to cancel the effects of the zero due to the
shunt inductance. Pole #2 will perform the function of
the antialias filters as described in the Antialias Filters
section. The following illustrates a sample calculation
for a shunt of 330 µΩ with a parasitic inductance of 2 nH.

The location of the pole #1 is given as:



321541

−× + ×



RC



For

R

= 330 µΩ,

SH

1

R

is calculated as approximately 480 Ω (use 470 Ω).

L

SH

= 2 nH,

1




RC



C

= 33 nF

R

1

SH

=

L

1

SH

The location of Pole #1 is 165,000 rads or 26.26 kHz.

This places the location of Pole # 2 at:



321

−× + ×



RC541RC





=

3.838 kHz




To ensure phase-matching between Channel 1 and
Channel 2, the pole at Channel 2 must also be positioned
at this location. With C = 33 nF, the new value of resistance
for the antialias filters on Channel 2 is approximately

1.23 kΩ (use 1.2 kΩ).

Figure 13 shows the effect of the compensation network
on the phase and magnitude response of the antialias
network in Channel 1. The dashed line shows the
response of Channel 2 using practical values for the

newly calculated component values, i.e., 1.2 kΩ and
33 nF. The solid line shows the response of Channel 1
with the parasitic shunt inductance included. Notice
phase and magnitude responses match very closely
with the ideal response—shown as a dashed line. This is
the objective of the compensation network.

Figure 13. Antialias Network Phase and Magnitude
Response after Compensation

The method of compensation works well when the pole
due to shunt inductance is less than 25 kHz or so. If zero
is at a much higher frequency its effects may simply be
eliminated by placing an extra RC on Channel 1 with a
pole that is a decade greater than that of the antialias
filter, e.g., 100 Ω and 33 nF.

Care should be taken when selecting a shunt to ensure
its parasitic inductance is small. This is especially true of
shunts with small values of resistance, e.g., <200 µΩ. Note
that the smaller the shunt resistance, the lower the
zero frequency for a given parasitic inductance (Zero =
R

SH1/LSH1

).

POWER SUPPLY DESIGN

This design uses a simple low-cost power supply based
on a capacitor divider network, i.e., C17 and C18. Most of
the line voltage is dropped across C17, a 470 nF 250 V
metalized polyester film capacitor. The impedance of
C17 dictates the effective VA rating of the supply. How­ever the size of C17 is constrained by the power
consumption specification in IEC1036. The total power
consumption in the voltage circuit, including power sup­ply, is specified in section 4.4.1.1 of IEC1036 (1996-9).
The total power consumption in each phase is 2 W and
10 VA under nominal conditions. The nominal VA rating of
the supply in this design is 7 VA. The total power dissi­pation is approximately 0.5 W. Together with the power
dissipated in the shunt at 40 A load, the total power con­sumption of the meter is 1.06 W. Figure 14 shows the
basic power supply design.

REV. A

–7–