Analog Devices ADT14 Datasheet

Quad Setpoint, Programmable

VOLTAGE

REFERENCE

AND

SENSOR

CURRENT

MIRROR

WINDOW

COMPARATORS

HYSTERESIS
VOLTAGE

TEMPERATURE
OUTPUT

HYSTERESIS

SETPOINT
OUTPUT 1

SETPOINT
OUTPUT 2

SETPOINT
OUTPUT 3

SETPOINT
OUTPUT 4

VPTAT

2.5V

V

REF

SET 1

SET 2

SET 3

SET 4

GND

V+

ADT14

a

Temperature Monitor and Controller

ADT14

FEATURES
Four Programmable Temperature Setpoints
Programmable Thermal Hysteresis
Accuracy 638C Typ from –408C to +1258C
Temperature Output Scale Factor = 5 mV/K
Resistor Programmable Temperature Setpoints
5 mA Open-Collector Setpoint Outputs
Internal 2.5 V Reference
600 mA Max Quiescent Current at +5 V

APPLICATIONS
Power Supply Monitor and Control System
Multiple Fan Controller System
Workstation Thermal Management System

GENERAL DESCRIPTION

The ADT14 is a temperature sensor and controller that generates
an output voltage proportional to temperature and provides four
temperature trip points. The four trip points, or temperature
setpoints, and their hysteresis are determined by voltage levels set
by the user. An on-chip voltage reference provides an easy method
for setting the temperature trip points.

The ADT14 consists of a bandgap voltage reference combined
with four matched comparators. The reference provides both a
temperature-stable 2.5 V output, and a voltage proportional to
absolute temperature (VPTAT) which has a precise temperature
coefficient of 5 mV/K = 5 mV/(°C +273.15). The VPTAT out­put is nominally 1.49 V at +25°C. The comparators determine
whether the VPTAT output is above the voltages set up by
external resistive dividers (temperature trip points) and generate
an open-collector output signal when one of their respective
thresholds has been exceeded.

Hysteresis is programmed by a user-selected voltage at the hys­teresis pin. This voltage adjusts the hysteresis current which is
used to generate a hysteresis offset voltage. The comparator’s
noninverting inputs are connected in parallel, which guarantees
that there is no hysteresis overlap and eliminates erratic transi­tions between adjacent trip zones.

Using a proprietary thin-film resistor process in conjunction
with production laser trimming, a temperature accuracy of ±3°C
at 25°C is guaranteed. The open-collector outputs are capable
of sinking 5 mA, and provide TTL/CMOS logic compatibility
with an external pull-up resistor. Operating from a single 5 V
supply, the quiescent current is 600 µA max.

The ADT14 is available in the 16-lead epoxy DIP and SO
(small outline) packages.

REV. 0

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] World Wide Web Site: http://www.analog.com
Fax: [phone] © Analog Devices, Inc., 1997

FUNCTIONAL BLOCK DIAGRAM

PIN CONFIGURATIONS

DIP & SO

OUTPUT 1

SETPOINT 1

NC
NC

VPTAT

SETPOINT 2

OUTPUT 2

1
2
3

ADT14

4

TOP VIEW

5

(Not to Scale)

6
7
8

NC = NO CONNECT

OUTPUT 4

16

SETPOINT 4

15

2.5V REFERENCE

14

V+

13

NCGROUND

12

HYSTERESIS

11

SETPOINT 3

10

OUTPUT 3

9

ADT14G–SPECIFICATIONS

(V+ = 5 V, –408C ≤ TA ≤ +1258C unless otherwise noted)

Parameter Symbol Conditions Min Typ Max Units

ACCURACY

Accuracy (Setpoints 1, 2, 3 and 4) T
Accuracy (Setpoints 1, 2, 3 and 4) T

= +25°C ±2 ±3 °C

A

= –40°C to +125°C ±3 ±5 °C

A

VPTAT Output Scale Factor +4.9 +5 +5.1 mV/K
Setpoint Hysteresis Levels T
Power Supply Rejection Ratio PSRR 4.5 V ≤ V

= +25°C 0.65, 1.5, 5 °C

A

≤ 13 V 0.1 0.25 °C/V

S

Linearity 0.5 °C

SETPOINT INPUTS

Offset Voltage V

OS

Offset Voltage Mismatch T
Output Voltage Drift TCV
Input Bias Current I

B

OS

T

= +25°C1mV

A

= +25°C 0.5 mV

A

3 µV/°C
25 100 nA

REFERENCE OUTPUT

Line Regulation 4.5 V ≤ V
Load Regulation I
Output Voltage V
Output Voltage V
Output Drift TCV

REF
REF

REF

SOURCE

T

= +25°C, No Load 2.49 2.5 2.51 V

A

No Load, –40°C ≤ TA ≤ +125°C 2.48 2.5 2.52 V

≤ 13 V ±0.1 %/V

S

= 10 µA to 500 µA ±0.25 %/mA

±30 ppm/°C

OPEN-COLLECTOR OUTPUTS

Output Low Voltage V

Output Low Voltage V
Output Leakage Current I
Fall Time t

OL

OL
OH
HL

I

= 1.6 mA,

SINK

–40°C ≤ T
I

SINK

≤ +85°C 0.25 0.4 V

A

= 5 mA 0.6 V

V+ = 12 V 1 100 µA
See Figure 1 40 ns

POWER SUPPLY

Supply Range V+ 4.5 5.5 V
Supply Current I
Supply Current I

Specifications subject to change without notice.

SY
SY

Unloaded at +5 V 400 600 µA
Unloaded at +12 V 450 µA

ABSOLUTE MAXIMUM RATINGS*

Maximum Supply Voltage . . . . . . . . . . . . . . –0.3 V to +15 V

Maximum Setpoint Input Voltage . . –0.3 V to ([V+] +0.3 V)

Maximum Reference Output Current . . . . . . . . . . . . . .1 mA

Maximum Open-Collector Output Current . . . . . . . . . 15 mA

Maximum Open-Collector Output Voltage . . . . . . . . . +15 V

Operating Temperature Range . . . . . . . . . –55°C to +150°C

Dice Junction Temperature . . . . . . . . . . . . . . . . . . . . +175°C

Storage Temperature Range . . . . . . . . . . . –65°C to +160°C

Lead Temperature (Soldering, 60 sec) . . . . . . . . . . . +300°C

*CAUTION

1. Stresses above those listed under Absolute Maximum
Ratings may cause permanent damage to the device. This is
a stress rating only; functional operation at or above this
specification is not implied. Exposure to the above maximum
rating conditions for extended periods may affect device
reliability.

2. Digital inputs are protected; however, permanent damage
may occur on unprotected units from high-energy electro­static fields. Keep units in conductive foam or packaging at all
times until ready to use. Use proper antistatic handling
procedures.

3. Remove power before inserting or removing units from their
sockets.

ORDERING GUIDE

Temperature Package Package

Model/Grade Range

1

Description Options

ADT14GP XIND Plastic DIP N-16
ADT14GS XIND SOIC R-16A

NOTES

1

XIND = 2408C to 11258C.

2

N = Plastic DIP; R = Small Outline.

THERMAL PACKAGE CHARACTERISTICS

Package Type u

JA

u

JC

Units

16-Lead Plastic DIP 103 34 8C/W
16-Lead SOIC 81 27 8C/W

V+

1kΩ

20pF

Figure 1. Test Load

2

–2–

REV. 0

ADT14

WAFER TEST LIMITS

(V+ = +5 V, GND = 0 V, TA = +258C unless otherwise noted)

Parameter Symbol Conditions Min Typ Max Units

INPUTS SET HIGH, SET LOW

Input Bias Current I

B

70 nA

OUTPUT VPTAT

Temperature Accuracy T

OUTPUT V

Nominal Value V

REF

REF

= +25°C, No Load 1.5 °C

A

T

= +25°C, No Load 2.490 2.510 V

A

Line Regulation 4.5 V ≤ V ≤ 13.2 V ±0.08 %/V
Load Regulation 10 µA ≤ IV

≤ 500 µA ±0.25 %/mA

REF

OPEN-COLLECTOR OUTPUTS OVER, UNDER

Output Low Voltage V
Output Leakage Current I

OL

OH

I

= 1.6 mA 0.4 V

SINK

100 µA

POWER SUPPLY

Supply Range V+ 4.5 5.5 V
Supply Current I

NOTE
Electrical tests are performed at wafer probe to the limits shown. Due to variations in assembly and nominal yield loss, yield after packaging is not guaranteed for
standard product dice. Consult factory to negotiate specifications based on lot qualification through sample lot assembly and testing.

SY

Unloaded 600 µA

DICE CHARACTERISTICS

Die Size 0.069 × 0.080 inch, 5520 sq. mils

(1.75 × 2.03 mm, 3.55 sq. mm)

Transistor Count: 130

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the ADT14 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.

REV. 0

–3–

WARNING!

ESD SENSITIVE DEVICE

ADT14

–Typical Performance Characteristics

160

140

120

100

80

60

TIME CONSTANT – sec

40

20

0

0 700100 200 300 400 500 600

TRANSITION FROM +100°C STIRRED OIL BATH
TO FORCED +25°C AIR

V+ = +5V, NO LOAD

SOIC SOLDERED TO 0.6" 0.5" Cu PCB

PDIP IN SOLDERTAIL SOCKET
SOLDERED TO 0.8"

AIR VELOCITY – FPM

0.6" Cu PCB

Figure 2. Thermal Time Constant in Forced Air

100

90

80
70

60
50

40
30

CHANGE IN TEMPERATURE – %

20
10

0

0305 10152025

SOIC SOLDERED TO 0.6" 0.5" Cu PCB

PDIP IN SOLDERTAIL SOCKET
SOLDERED TO 0.8"

V+ = +5V, NO LOAD

0.6" Cu PCB

TIME – sec

40µs

V

VPTAT

IN

1.5V

5V

100

90

0V

10

0%

0V

Figure 5. Start-Up Response

5

4

3
2

1
0

1
2

ACCURACY ERROR – °C

3
4

5

50 15025 25 50 75 125

V+ = +5V, NO LOAD

TEMPERATURE – °C

500mV

MAX LIMIT

MIN LIMIT

1000

Figure 3. Thermal Response in Stirred Oil Bath

5.0

START-UP VOLTAGE DEFINED AS OUTPUT VALUE
BEING WITHIN

4.5

4.0

3.5

START-UP VOLTAGE – V

3.0
75 17525 25 75 125

5 C OF OUTPUT AT +5V

NO LOAD

TEMPERATURE – C

Figure 4. Start-Up Voltage vs. Temperature

Figure 6. Accuracy Error vs. Temperature, Stirred Oil Bath

600

500

400

300

200

SUPPLY CURRENT – A

100

0

0162 4 6 8 10 12 14

TA = +25°C, NO LOAD

SUPPLY VOLTAGE – V

Figure 7. Supply Current vs. Supply Voltage

–4–

REV. 0

ADT14

500

475

450

425

400

375

SUPPLY CURRENT – µA

350

325

300

75 17525 25 75 125

TEMPERATURE – °C

V+ = +5V, NO LOAD

Figure 8. Supply Current vs. Temperature

40

35

30

25

20

VOL = +1V, V+ = +5V

2.52

2.515

2.51

2.505

2.5

2.495

2.49

REFERENCE VOLTAGE – V

2.485

2.48
75 17525 25 75 125

TEMPERATURE – °C

MAX LIMIT

V+ = +5V, NO LOAD

MIN LIMIT

Figure 11. Reference Voltage vs. Temperature

0.1
V+ = +4.5V TO +13V

NO LOAD

0.075

0.05

15

10

5

OPEN-COLLECTOR SINK CURRENT – mA

0

75 17525 25 75 125

TEMPERATURE – °C

Figure 9. Open-Collector Output Sink Current vs.
Temperature

700

V+ = +5V, NO LOAD

600

500

400

300

200

100

OPEN-COLLECTOR OUTPUT VOLTAGE – mV

0

75 17525 25 75 125

I

= 5mA

LOAD

I

= 1mA

LOAD

TEMPERATURE – °C

0.025

POWER SUPPLY REJECTION – C/V

0

25

50

0 25 50 75 100 125

TEMPERATURE – °C

150

Figure 12. VPTAT Power Supply Rejection vs. Temperature

500µs

20mV

V

2.5V

LOAD

V

REF

100

90

0V

10

0%

Figure 10. Open-Collector Output Voltage vs. Temperature

REV. 0

–5–

Figure 13. V

= 5 k

R

LOAD

Under Load Switching (0 µA–500 µA),

REF

Ω

ADT14

APPLICATIONS INFORMATION
Programming the ADT14

To program any of the four temperature trip points, simply set
the voltage at the setpoint pin equal to the voltage expected at
the VPTAT output at the desired trip temperature. The on­board 2.5 V precision reference provides a temperature stable
output that is used to establish the setpoint voltages. Figure 14
illustrates the basic temperature setpoint configuration. This
parallel resistor ladder configuration uses six resistors to set the
four setpoint voltages.

I

L

I

L1

R1

R2 R5

R3 R6

V

I

L2

R4

REF

SET 4

ADT14

SET 3

SET 2

SET 1

GND

HYS

Figure 14. ADT14 Setpoint Programming

For SET4,

T4 = 100°C
V(SET4) = (T4 + 273.15)(5 mV/K)

For SET1,

T1 = 75°C
V(SET1) = (T1 + 273.15)(5 mV/K)

Choose I

R1 = (V
R2 = (V[SET4] – V[SET1])/I
R3 = V(SET1)/I

= I

= 25 µA,

L1

L2

– V[SET4])/I

REF

L1

L1

L1

For SET3,

T3 = 50°C
V(SET3) = (T3 + 273.15)(5 mV/K)

For SET2,

T2 = –10°C
V(SET2) = (T2 + 273.15)(5 mV/K)

Since I

= I

= 25 µA,

L1

R4 = (V
R5 = (V[SET3] – V[SET2])/I
R6 = V(SET2)/I

L2

– V[SET3])/I

REF

L2

L2

L2

For the example shown in Figure 14 and the related equations,
Setpoint 4 was chosen as highest temperature trip point, fol­lowed in order by Setpoint 1, Setpoint 3 and Setpoint 2. The
user may choose to alter this order at will as all the setpoint
inputs and trip point outputs are identical in operation. The
minimum recommended parallel ladder resistance is 12.5 kΩ.
This maintains a V
errors, choose 50 µA < I

load current less than 200 µA. To avoid

REF

< 200 µA.

L

For those applications where setpoints do not need to be
changed, a single ladder of five resistors can be used. Figure 15
illustrates a single resistor ladder configuration.

L

I

LR1

< 200 µA

L

L
L
L

50

A < IL < 200 A

T4 > T1 > T3 > T2

SET4

SET1

SET3

SET2

50 µA < I

R1 = (V

– V[SET4])/I

REF

R2 = (V[SET4] – V[SET1])/I
R3 = (V[SET1] – V[SET3])/I
R4 = (V[SET3] – V[SET2])/I
R5 = V(SET2)/I

L

V

REF

R2

R3

R4

R5

Figure 15. Single Resistor Ladder

Temperature Hysteresis

Temperature hysteresis is the number of degrees below the
original setpoint temperature that must be sensed by the
ADT14 before the setpoint comparator will be reset and the
open-collector output disabled. Figure 16 shows the hysteresis
profile. Hysteresis is programmed by the user by setting a speci­fied voltage at the hysteresis pin (See Table I).

HYSTERESIS

HI

OUTPUT VOLTAGE

LO

T

SET2

T

SET3

TEMPERATURE

T

SET1

T

SET4

Figure 16. ADT14 Hysteresis Profile

Table I. Common Hysteresis Values and Voltages

Hysteresis Connect Pin 11 to

0.65°C +2.5 V Reference

1.5°C Leave Open
5°C Ground

The voltage on Pin 11 programs the internal hysteresis current,
which is mirrored and fed to a buffer controlled by an analog
switch. After a temperature setpoint has been exceeded and a
comparator tripped, the buffer is enabled, allowing hysteresis
current to flow. This current generates a hysteresis offset voltage

–6–

REV. 0

ADT14

across an internal 1 kΩ resistor at the comparator input. The
comparator output remains on until the voltage at the compara­tor input, now equal to VPTAT plus the hysteresis offset, has
returned to the setpoint voltage. At this point the comparator
turns off, the open-collector output is deactivated, and the hys­teresis current buffer is disabled.

While Table I shows simple pin-programmable hysteresis val­ues, the user may design for intermediate values of hysteresis
between 0.65°C and 5°C using a resistive divider. To set the
hysteresis in the range 0.65°C < T
connected from the HYS pin (Pin 11) to V

< 1.5°C, a resistor is

HYS

(Pin 14). The

REF

value of the resistor is given by the following equation:

R1=

For various values of T

57.5

1. 5 − Hysteresis

, the table below shows the values for

HYS

–57.5

R1:

Table II

T

(8C) R1 (kV)

HYS

0.7 14.4

0.8 24.6

0.9 38.3

1.0 57.5

1.1 86.3

1.2 134.2

1.3 230

1.4 517.5

To set hysteresis in the range of 1.5°C < T

< 5°C, a resistor

HYS

is connected from the HYS pin (Pin 11) to the ADT14’s com­mon (Pin 5). The value of the resistor is given by:

R2 =

For various values of T

201.1

Hysteresis –1.5

, the table below shows values for R2:

HYS

–57.5

Table III

T

(8C) R2 (kV)

HYS

1.6 1953

1.75 746.9

2.0 344.7

2.25 210.6

2.5 143.6

2.75 103.4

3.0 76.6

3.25 57.4

3.50 43.1

3.75 31.8

4.0 22.9

4.25 15.6

4.5 9.5

4.75 4.4

The schematics for the two circuits are shown below. As the
resistor values are approximate, it is a good idea to add a poten­tiometer for fine adjustments.

HYS

P1

HYS

< 1.5°C

ADT14

1.5°C < T

HYS

< 5°C

HYS

R2

P2

R1

V

REF

ADT14

0.65°C < T

Figure 17. Hysteresis Connections for Miscellaneous
Values

Understanding Error Sources

The accuracy of the VPTAT sensor output is well characterized
and specified, however preserving this accuracy in a heating or
cooling control system requires some attention to minimizing
potential error sources. The internal sources of setpoint pro­gramming error include the initial tolerances and temperature
drifts of the reference voltage V

, the setpoint comparator

REF

input offset voltage and bias current, and the hysteresis current
scale factor. When evaluating setpoint programming errors,
remember that any V

error contribution at the comparator

REF

inputs is reduced by the resistor divider ratios. Each comparator’s
input bias current drops to less than 1 nA (typ) when the com­parator is tripped. This change accounts for some setpoint volt­age error, equal to the change in bias current multiplied by the
effective setpoint divider ladder resistance to ground.

The thermal mass of the ADT14 package and the degree of
thermal coupling to the surrounding circuitry are the largest
factors in determining the rate of thermal settling, which ulti­mately determines the rate at which the desired temperature
measurement accuracy may be reached (see Figure 2). Thus,
one must allow sufficient time for the device to reach the final
temperature. The typical thermal time constant for the plastic
SOIC package is approximately 70 seconds in still air. There­fore, to reach the final temperature accuracy within 1%, a set­tling time of five time constants, or six minutes, is necessary.

External error sources to consider are the accuracy of the exter­nal programming resistors, ground voltage errors, and thermal
gradients. The accuracy of the external programming resistors
directly impacts the resulting setpoint accuracy. Thus, in fixed­temperature applications the user should select resistor toler­ances appropriate to the desired programming accuracy. Resistor
temperature drift must also be taken into account. This effect
can be minimized by selecting high quality components, and by
keeping all components in close thermal proximity. Applications
requiring high measurement accuracy require great attention to
detail regarding thermal gradients. Careful circuit board layout,
component placement, and protection from stray air currents
are necessary to minimize common thermal error sources. The
user should also take care to keep the bottom of the setpoint
programming divider ladder as close as possible to GND (Pin 5)
to minimize errors due to IR voltage drops and coupling of
external noise sources. In any case, a 0.1 µF capacitor for power
supply bypassing is always recommended at the chip.

REV. 0

–7–

ADT14

Safety Considerations

In heating and cooling system design, designers should antici­pate potential system fault conditions which may result in sig­nificant safety hazards which are outside the control of, and
cannot be corrected by, the ADT14 based circuit. Governmen­tal and industrial regulations regarding safety requirements and
standards for such designs should be observed where applicable.

Self-Heating Effects

In some applications the user should consider the effects of self­heating due to the power dissipated by the open-collector out­puts, which are capable of sinking 5 mA each continuously.
Under full load, the ADT14 open-collector output device is
dissipating,

P

=0.6 V ×0.005 A ×4 =12 mW

DISS

which, in the small outline package, accounts for a temperature
increase due to self-heating of

∆T = P

DISS×θJA

=0.012 W ×

81°C

W

= 0.97°C

This will directly affect the accuracy of the ADT14 and will, for
example, cause the device to switch the heating output off 0.97
degrees early. Alternatively, bonding the same package to a
moderate heatsink limits the self-heating effect to approximately

∆T = P

DISS×θJC

=0.012 W ×

which is a much more tolerable error in most systems. The V

27°C

W

= 0.32°C

REF

and VPTAT outputs are also capable of delivering sufficient
current to contribute heating effects and should not be ignored.

Buffering the Voltage Reference

The reference output V

is used to generate the temperature

REF

setpoint programming voltages for the ADT14. The onboard
V

output buffer is capable of 500 µA output drive into as

REF

much as a 50 pF load. Exceeding this load will affect the accu­racy of the reference voltage, will increase thermal errors due to
internal heat generation, and may induce oscillations. External
buffering of V

with a low drift voltage follower will ensure

REF

optimal reference accuracy if a large load current is required.
Amplifiers that offer low drift, low power consumption, and low
cost appropriate to this application include the OP295 and
members of the OP90, OP97, OP177 families, and others shown
in the following applications circuits.

With excellent drift and noise characteristics, V

offers a good

REF

voltage reference for data acquisition and transducer excitation
applications as well.

Preserving Accuracy Over Wide Temperature Range Operation

The ADT14 is unique in offering both a wide-range tempera­ture sensor and the associated detection circuitry needed to
implement a complete thermostatic control function in one
monolithic device. The voltage reference, setpoint comparators,
and output buffer amplifiers have been carefully compensated to
maintain accuracy over the specified temperature ranges in this
application. Since the ADT14 is both sensor and control circuit,
in many applications the external components used to program
and interface the device are subjected to the same temperature
extremes. Thus, it is necessary to place components in close
thermal proximity to minimize temperature differentials, and to

account for thermal drift errors where appropriate, such as
resistor matching temperature coefficients, amplifier error drift,
and the like. Circuit design with the ADT14 requires a slightly
different perspective regarding the thermal behavior of elec­tronic components.

Thermal Response Time

The time required for a temperature sensor to settle to a speci­fied accuracy is a function of the thermal mass of the sensor,
and the thermal conductivity between the sensor and the object
being sensed. Thermal mass is often considered equivalent to
capacitance. Thermal resistance is commonly specified in units
of degrees per watt of power transferred across the thermal joint.
Figure 3 illustrates the typical response to a step change in am­bient temperature for PDIP and SOIC packages. Thus, the time
required for the ADT14 to settle to the desired accuracy is
dependent on the package selected, the thermal contact estab­lished in the particular application, and the equivalent thermal
conductivity of the heat source. For most applications, the set­tling time is probably best determined empirically.

Switching Loads with the Open-Collector Outputs

In many temperature sensing and control applications some type
of switching is required. Whether it’s to turn on a heater when
the temperature goes below a minimum value or to turn off a
motor that is overheating, the open-collector outputs can be
used. For the majority of applications, the switches used need to
handle large currents on the order of 1 amp and above. Because
the ADT14 is accurately measuring temperature, the open­collector outputs should handle less than 5 mA of current to
minimize self-heating. Clearly, the trip point outputs should not
drive the equipment directly. Instead, an external switching
device is required to handle the large currents. Some examples
of these are power MOSFETs, thyristors, IGBTs, and Darlingtons.
Figures 18a–18d show a variety of circuits where the ADT14
controls a switch. The main consideration in these circuits is the
current required to activate the switch.

Power FETs are popular for handling a variety of high current
DC loads. Figure 18b shows the ADT14 driving a P-channel
MOSFET transistor for a simple heater circuit. When the out­put transistor turns on, the gate of the MOSFET is pulled down
to approximately 0.6 V, turning it on. For most MOSFETs a
gate-to-source voltage, or V

, on the order of –2 V to –5 V is

GS

sufficient to turn on the device.
Isolated Gate Bipolar Transistors (IGBT) combine many of the

benefits of power MOSFETs with bipolar transistors, and are
used for a variety of high power applications. Because IGBTs
have a gate similar to MOSFETs, turning the devices on and off
is relatively simple as shown in Figure 18c. The turn-on voltage
for the IGBT shown (IRGB40S) is between 3.0 and 5.5 volts.
This part has a continuous collector current rating of 50 A and a
maximum collector-to-emitter voltage of 600 V, enabling it to
work in very demanding applications.

The last class of high power devices discussed here are thyris­tors, which include SCRs and triacs. Triacs are a useful alter­native to relays for switching ac line voltages. The 2N6073A
shown in Figure 18d is rated to handle 4 A (rms). The opto­isolated MOC3021 triac shown features excellent electrical
isolation from the noisy AC line and complete control over the
high power triac with only a few additional components.

–8–

REV. 0

ADT14

1

2

3

NC

4

NC

5

6

7

8

NC = NO CONNECT

ADT14

TEMP

SENSOR

& VOLTAGE

REFERENCE

HYSTERESIS

GENERATOR

Figure 18a. Reed Relay Drive

NC

NC

16

12V

15

V

REF

14

13

12

11

10

9

1

2

3

4

5

6

7

2604-12-311 COTO

IN4001 OR EQUIVALENT

ADT14

MOTOR
SHUTDOWN

TEMP

SENSOR

& VOLTAGE

REFERENCE

HYSTERESIS

GENERATOR

1

2

3

NC

4

NC

5

6

7

8

NC = NO CONNECT

16

15

V

REF

14

13

12

11

10

16

15

V

REF

ADT14

TEMP

SENSOR

& VOLTAGE

REFERENCE

HYSTERESIS
GENERATOR

14

13

12

11

10

9

A7812

(24V ONLY)

INOUT

2.4kΩ (24V)

1.2kΩ (6V)
5%

IRFR9022 OR
EQUIVALENT

Figure 18b. Driving a P-Channel MOSFET

V+

4.7kΩ

4.7kΩ

24V/6V

HEATING
ELEMENT

8

NC = NO CONNECT

1

2

3

NC

4

NC

5

6

7

8

NC = NO CONNECT

9

Figure 18c. Driving an IGBT

16

V+ = +5V

15

V

ADT14

TEMP

SENSOR

& VOLTAGE

REFERENCE

HYSTERESIS
GENERATOR

REF

14

13

12

11

10

9

500Ω

1
2

MOC3011

3

Figure 18d. Controlling the 2N6073A Triac

MOTION
CONTROL

1RGBC40S2N1711

LOAD

150Ω

6
5
4

2N6073A

AC

REV. 0

–9–

ADT14

High Current Switching

As mentioned earlier, internal dissipation due to large loads on
the ADT14 outputs will cause some temperature error due to
self-heating. External transistors buffer the load from the ADT14,
so that virtually no power is dissipated in the internal transistors
and minimal self-heating occurs. Figures 19a–19c show a few
examples using external transistors. The simplest case uses a
single transistor on the output to invert the output signal as
shown in Figure 19a. When the open-collector of the ADT14
turns ON and pulls the output down, the external transistor Q1’s
base will be pulled low, turning off the transistor. Another transis­tor can be added to reinvert the signal as shown in Figure 19b.

16

15

V

V+

REF

14

13

12

11

10

9

4.7kΩ

I

L

Q1

NC

NC

1

2

ADT14

3

4

5

6

7

8

TEMP

SENSOR

& VOLTAGE

REFERENCE

HYSTERESIS
GENERATOR

Now, when the output of the ADT14 is pulled down, the first
transistor, Q1, turns off and its collector goes high, which turns
on Q2, pulling its collector low. Thus, the output taken from the
collector of Q2 is identical to the output of the ADT14. By pick­ing a transistor that can accommodate large amounts of current,
many high power devices can be switched.

An example of a higher power transistor is a standard Darlington
configuration as shown in Figure 19c. The part chosen, TIP-110,
can handle 2 A continuous, which is more than enough to control
many high power relays. In fact the Darlington itself can be used
as the switch, similar to MOSFETs and IGBTs.

16

15

V+

V

REF

14

13

4.7kΩ

4.7kΩ

Q1
2N1711

I

L

Q2
2N1711

12

11

10

9

NC

NC

1

2

ADT14

3

4

5

6

7

8

TEMP

SENSOR

& VOLTAGE

REFERENCE

HYSTERESIS

GENERATOR

NC = NO CONNECT

Figure 19a. An External Resistor Minimizes Self-Heating

1

2

3

NC

4

NC

5

6

7

8

NC = NO CONNECT

ADT14

TEMP

SENSOR

& VOLTAGE

REFERENCE

HYSTERESIS
GENERATOR

16

15

V

14

13

12

11

10

9

Figure 19c. Darlington Transistor Can Handle Large Currents

NC = NO CONNECT

Figure 19b. Second Transistor Maintains Polarity of
ADT14 Output

REF

4.7kΩ

V+

2N1711

+12V

4.7kΩ

TIP110

I

L

RELAY

–10–

REV. 0

ADT14

ADT14

VPTAT

6

7

2

3

4

V+

GND

V

REF

OP90

R2

39.2kΩ

R1

243kΩ

R3

100kΩ

R6

100Ω

R5

100kΩ

2N1711

14

5

13

6

+5V TO +13.2V

R

L

4-20mA

Buffering the Temperature Output Pin

The VPTAT sensor output is a low impedance dc output volt­age with a 5 mV/K temperature coefficient, and is useful in a
number of measurement and control applications. In many
applications, this voltage may need to be transmitted to a central
location for processing. The unbuffered VPTAT voltage output
is capable of 500 µA drive into 50 pF (max). As mentioned in
the discussion regarding buffering circuits for the V

output, it

REF

is useful to consider external amplifiers for interfacing VPTAT
to external circuitry to ensure accuracy, and to minimize load­ing, which could create dissipation-induced temperature sensing

the OP90s is 20 µA max, totaling much less than 4 mA. Although
not shown, the open collector outputs and temperature setting
pins can be connected to do any local control of switching.

The current is proportional to the voltage on the VPTAT out­put, and is calibrated to 4 mA at a temperature of –40°C, to
20 mA for +85°C. The main equation governing the operation
of this circuit gives the current as a function of VPTAT:

I

OUT



1

=



R6



VPTAT ×R5

R2





×R3

V

REF

−





R3+R1





×1+

R5
R2






errors. An excellent general-purpose buffer circuit using the
OP177, which is capable of driving over 10 mA and will remain
stable under capacitive loads of up to 0.1 µF, is shown in Figure

20. Other interface circuits are shown below.

10kΩ

0.1 F

V+

ADT14

VPTAT

V–

OP177

100Ω

V

OUT

Figure 20. Buffer VPTAT to Handle Difficult Loads

Differential Transmitter

In noisy industrial environments, it is difficult to send an accu­rate analog signal over a significant distance. However, by send­ing the signal differentially on a wire pair, these errors can be
significantly reduced. Since the noise will be picked up equally
on both wires, a receiver with high common-mode input rejec­tion can be used very effectively to cancel out the noise at the
receiving end. Figure 21 shows two amplifiers being used to
send the signal differentially, and an excellent differential re­ceiver, the AMP03, (SSM2141 or SSM2143 are two other
options), which features a common-mode rejection ratio of
95 dB at dc and very low input and drift errors.

4.9kΩ

ADT14

VPTAT

10kΩ

V+

V–

Figure 21. Send the Signal Differentially for Noise
Immunity

4 mA to 20 mA Current Loop

Another very common method of transmitting a signal over long

1/2 OP297

10kΩ

1/2 OP297

50Ω

50Ω

V+

AMP03 OR
SSM2143

V–

V

OUT

distances is to use a 4 mA-to-20 mA loop (see Figure 22). An
advantage of using a 4 mA-to-20 mA loop is that the accuracy of
a current loop is not compromised by voltage drops across the
line. One requirement of 4 mA-to-20 mA circuits is that the
remote end must receive all of its power from the loop, meaning

To determine the resistor values in this circuit, first note that
V

REF

over R2 must give a variation of I
VPTAT varies from 1.165 V at –40°C to 1.79 V at +85°C. The
absolute value of the resistors is not important, only the ratio.
For convenience, 100 kΩ is chosen for R5. Once R2 is calcu­lated, the value of R3 and R1 is determined by substituting
4 mA for I
values are shown in the circuit. The OP90 is chosen for this
circuit because of its ability to operate on a single supply and its
high accuracy. For initial accuracy, a 10 kΩ trim potentiometer
can be included in series with R3, and the value of R3 lowered
to 95 kΩ. The potentiometer should be adjusted to produce an
output current of 12.3 mA at 25°C.

Temperature-to-Frequency Converter

Another common method of transmitting analog information is
to convert a voltage to the frequency domain. This is easily done
with any of the available low cost monolithic Voltage-to-Fre­quency Converters (VFCs) that feature an open-collector digital
output. A digital signal is immune to noise and voltage drops
because the only important information is the frequency. As
long as the conversions between temperature and frequency are
accurately performed, the temperature data can be accurately
transmitted.

A simple circuit to do this combines the ADT14 with an AD654
VFC and is shown in Figure 23. The AD654 outputs a square
wave that is proportional to the dc input voltage according to
the following equation:

that the circuit must consume less than 4 mA. Operating from
+5 V, the quiescent current of the ADT14 is 600 µA max, and

Figure 22. 4 mA to 20 mA Current Loop

remains constant over temperature. Thus the ratio of R5

from 4 mA to 20 mA as

OUT

and 1.165 V for VPTAT and solving. The final

OUT

V

F

=

OUT

10 (R1+ R2) C

IN

T

REV. 0

–11–

ADT14

By connecting the VPTAT output to the input of the AD654,
the 5 mV/K, temperature coefficient gives a sensitivity of
25 Hz/°C, centered around 7.5 kHz at 25°C. The trimming
resistor, R2, is needed to calibrate the absolute accuracy of the
AD654. For more information on the converter, consult the
AD654 data sheet. An AD650 frequency-to-voltage converter
can be used to accurately convert the frequency back to a dc
voltage on the receiving end.

ADT14

VPTAT

1.8kΩ

500Ω

R1

R2

V+

867

4

3

C1

0.1

OSC

AD654

52

F

V+

5kΩ

1

F

OUT

Figure 23. Temperature-to-Frequency Converter

Isolation Amplifier

In many industrial applications the sensor is located in an envi­ronment that needs to be electrically isolated from the central
processing area. Figure 24 shows a simple circuit that uses an
8-pin opto-isolator (IL300XC) that can operate across a 5,000
V barrier. IC1 (an OP290 single-supply amplifier) is used to
drive the LED connected between Pins 1 to 2. The feedback
actually comes from the photodiode connected to Pins 3 to 4.
The OP290 drives the LED such that there is enough current
generated in the photodiode to exactly equal the current derived

from the VPTAT voltage across the 470 kΩ resistor. On the
receiving end, an OP90 converts the current from the second
photodiode to a voltage through its feedback resistor R2. Note
that the other amplifier in the dual OP290 is used to buffer the

2.5 V reference voltage of the ADT14 for an accurate, low drift
LED bias level without affecting the programmed hysteresis
current. A REF43 (a precision 2.5 V reference) provides an
accurate bias level at the receiving end.

To understand this circuit, it helps to examine the overall equa­tion for the output voltage. First, the current (I1) in the photo­diode is set by:

2.5V –VPTAT

I1=

470 kΩ

Note that the IL300XC has a gain of 0.73 (typical) with a min
and max of 0.693 and 0.769 respectively. Since this is less than

1.0, R2 must be larger than R1 to achieve overall unity gain. To
show this the full equation is:

V

=2.5V – I 2R2 = 2.5V –0.7

OUT



2.5V – VPTAT



470 kΩ





×644 kΩ=VPTAT





A trim is included for R2 to correct for the initial gain accuracy
of the IL300XC. To perform this trim, simply adjust for an
output equal to VPTAT at any particular temperature. For
example, at room temperature, VPTAT = 1.49 V, so adjust R2
until V

= 1.49 V as well. Both the REF43 and the OP90

OUT

operate from a single supply, and contribute no significant error
due to drift.

In order to avoid the accuracy trim, and to reduce board space,
complete isolation amplifiers, such as the high accuracy AD202,
are available.

V

ADT14

VPTAT

REF

OP290

R1

470kΩ

IL300XC

IN9148

1

2

3

I

1

4

ISOLATION

BARRIER

V+

OP290

100Ω

60µF

Figure 24. Isolation Amplifier

V+

REF43

V+

OP90

680pF

100kΩ

2
4

1.16V TO 1.7V

6

2.5V

I

2

604kΩ

–12–

REV. 0

Out-of-Range Warning

6

7

2

3

4

OP177

+15V

105kΩ 4.22kΩ

10pF

–15V

100kΩ

100kΩ

4.12kΩ 487Ω

V

OUT

(10mV/°C)

(V

OUT

= 0.0V

@ T = 0.0°C)

ADT14

V

REF

VPTAT

14

6

In Figure 25, connecting an open-collector output of the
ADT14 and an inverted open-collector output together into a
wired-OR configuration, a temperature “out-of-range” warning
signal is generated. This can be useful in sensitive equipment
calibrated to work over a limited temperature range.

ADT14

1

2

3

NC

4

NC

5

6

7

8

NC = NO CONNECT

ADT14

TEMP

SENSOR

& VOLTAGE

REFERENCE

HYSTERESIS
GENERATOR

16

V

REF

15

14

13

12

NC

11

10

9

47kΩ

2N1711

V+

LED

200Ω

Figure 25. Out-of-Range Warning

Translating 5 mV/K to 10 mV/8C

A useful circuit is shown in Figure 26 that translates the
VPTAT output voltage, which is calibrated in Kelvins, into an
output that can be read directly in degrees Celsius on a voltme­ter display. To accomplish this, an external amplifier is config­ured as a differential amplifier. The resistors are scaled so the
V

voltage will exactly cancel the VPTAT voltage at 0.0°C.

REF

Figure 26. Translating 5 mV/K to 10 mV/8C

However, the gain from VPTAT to the output is two, so that
5 mV/K becomes 10 mV/°C. Thus, for a temperature of +80°C,
the output voltage is 800 mV. Circuit errors will be due prima­rily to the inaccuracies of the resistor values. Using 1% resistors
the observed error was less than 10 mV, or 1°C. The 10 pF
feedback capacitor helps to cancel the effects of stray capaci­tance that could cause oscillations. For improved accuracy, an
adjustment potentiometer can be added in series with either
100 kΩ resistor.

Translating VPTAT to the Fahrenheit Scale

A similar circuit to the one shown in Figure 26 can be used to
translate VPTAT into an output that can be read directly in
degrees Fahrenheit, with a scaling of 10 mV/°F. Only unity gain
or less is available from the first stage differentiating circuit, so
the second amplifier provides a gain of two to complete the
conversion to the Fahrenheit scale. Using the circuit in Figure
27, a temperature of 0.0°F gives an output of 0.00 V. At room
temperature (77°F) the output voltage is 770 mV. A –40°C to
+85°C operating range translates into –40°F to +85°F. The
errors are essentially the same as for the circuit in Figure 26.

REV. 0

V

ADT14

VPTAT

REF

10pF

90.9kΩ 1kΩ

+15V

100kΩ

14

6.49kΩ

6

121Ω

100kΩ

2

1/2 OP297

3

7

4

–15V

100kΩ

6

6

1/2 OP297

5

100kΩ

7

Figure 27. Translating 5 mV/K to 10 mV/8F

–13–

ADT14

USER

INPUTS

0.1mF

0.1mF

P0.0 TO
P0.7

OUT

CONT

+5V

V

CC

TLC555

GND

RS

DIS

THR

TRIG

P3.2
P3.1
P3.0
P3.3

P2.0
P2.1
P2.2
P2.3
P2.4P1.0/INT2

TO 2ND
AD8403

CS

CLK
SDI

SHDN

NC

2.5V

REF

SET1

SET2

SET3

AD840380CL51mC

V

GND

DD

+5V

10mF

ADT14

SET4

HYS
TRIP1
TRIP2
TRIP3
TRIP4

V

GND

DD

+5V

0.1mF

NC = NO CONNECT

Figure 28. Typical Microcontroller Setup Configuration

Intelligent Temperature Control System—Interfacing the
8051 mC with the ADT14

Connecting the 8051, or any modern microcontroller, with the
ADT14 yields a powerful temperature control tool. For ex­ample, by using digitally controlled resistors (AD8402/3), a user
may automatically set the temperature trip points via computer
or touch pad. The ADT14 outputs are fed back to the controller
allowing it to sense whether or not a setpoint has been exceeded.
The higher temperature trip-point data can then be used to turn
on a fan to either a high or low setting. In addition, the lower
temperature trip-point data may be used to turn on a heater to
either a high or low setting.

The AD8403 is shown in a single ladder resistor configuration
with an additional fix resistor to set the voltages of all four
setpoints. The microcontroller program that runs the AD8403
can also be set to ensure that no out-of-bound conditions occur.
Finally, the AD8403 also has shutdown pin which, when acti­vated, will lower power consumption to 5 µA max (0.01 µA
typical), while still retaining memory of the potentiometer set­tings.

Power-Down in Battery Powered Applications

Battery powered applications require low power consumption
components. The ADT14 is just such a device. The low supply
current (< 600 µA) of the ADT14 makes it possible to power
the device from any suitable logic output pin. Figure 29 is a
schematic of an 'HC04 CMOS inverter driving the ADT14. When
the SLEEP pin is high, the ADT14 is powered down. Figure 5
shows a typical start-up response of an ADT14. Most tempera­ture measurements vary slowly over time, which allows designs

that consume very little power over very long periods of time by
using a duty-cycling technique. The basic idea of the duty­cycling technique is to simply measure the temperature at se­lected time intervals and leave the device powered down the rest
of the time. Instead of a CMOS inverter, any other logic pin
with sufficient drive capability (such as the output of a micro­processor) can be used to power on and off the ADT14.

'HC04

SLEEP

V+

ADT14

Figure 29. Using a CMOS Inverter to Power Down the
ADT14

Combining Trip Point Outputs to Yield 25%, 50%, 75%, and
100% Power Outputs

It is often desirable to power a device at varying levels based
upon ambient temperature. For example, suppose a liquid is
below a desired temperature, the lowest trip point temperature.
In this case, full power should be delivered to the heating ele­ment, causing the temperature to rise as rapidly as possible.
When the lowest trip-point is finally crossed, a reduction in
power should occur, slowing the rate of increase in the heating
process. This process continues until all trip points are crossed
and the current to the heating element is reduced to zero.

–14–

REV. 0

ADT14

Figure 30 illustrates the system described in the previous para­graph, where each trip-point output corresponds to a 25% re­duction in power when tripped. This is accomplished by the
dual rail-to-rail precision OP284 op amp configured as an in­verting summer in the first stage, and as a V/I converter in the
second stage.

Suppose, however, that the heating element at a reduced power
level is unable to raise the ambient temperature to the desired

ADT14

+5V

V

CC

TRIP4 (0%)

TRIP3 (25%)

TRIP2 (50%)

TRIP1 (75%)

2.5V

REF

setting. This issue can be addressed by using a microcontroller
as shown in Figure 28. The microcontroller allows the time for
the given volume of material to heat up from a low trip-point
temperature to the next highest temperature trip-point to be
measured. If this time is too long, the microcontroller could add
additional power to the heater by altering a setpoint level.

10kΩ

10kΩ

10kΩ

10kΩ 2.49kΩ

+5V

1/2 OP284

1/2 OP284

ELEMENT

+5V

Q1

HEATING

Figure 30. Combination Trip-Point Output

REV. 0

–15–

ADT14

0.210 (5.33)
MAX

0.160 (4.06)

0.115 (2.93)

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm).

16-Lead Plastic DIP

(N-16)

0.840 (21.34)

0.745 (18.92)

16

18

PIN 1

0.022 (0.558)

0.014 (0.356)

0.100
(2.54)

BSC

9

0.280 (7.11)

0.240 (6.10)

0.060 (1.52)

0.015 (0.38)

0.130
(3.30)
MIN

0.070 (1.77)

0.045 (1.15)

SEATING
PLANE

16-Lead Narrow-Body SOIC

(R-16A)

0.3937 (10.00)

0.3859 (9.80)

0.325 (8.26)

0.300 (7.62)

0.015 (0.381)

0.008 (0.204)

0.195 (4.95)

0.115 (2.93)

C3067–12–5/97PRINTED IN U.S.A.

0.1574 (4.00)

0.1497 (3.80)

0.0098 (0.25)

0.0040 (0.10)

SEATING

PLANE

16 9

PIN 1

0.0500

0.0192 (0.49)

(1.27)

0.0138 (0.35)

BSC

0.2440 (6.20)

81

0.2284 (5.80)

0.0688 (1.75)

0.0532 (1.35)

0.0099 (0.25)

0.0075 (0.19)

0.0196 (0.50)

0.0099 (0.25)

8°
0°

0.0500 (1.27)

0.0160 (0.41)

x 45°

–16–

REV. 0