Titan 2000 schematic

Elektor Electronics 2/99

It could be argued

that most of the out-

put amplifiers pub-

lished in this maga-

zine lack power.

Although this is a

debatable point, it

was felt that a true

heavyweight output

amplifier would make

a welcome change for

many constructors.

The Titan 2000 can

produce 300 watts

into 8 Ω, 500 watts

into 4 Ω, and

800 watts into 2 Ω.

For those who believe

that music power is a

reputable quantity, the

amplifier can deliver

2000 watts of this

magical power into

4 Ω.

58

Design by T. Giesberts

Titan 2000

High-power hi-fi and

public-address amplifier

Brief parameters

Sine-wave power output 300 W into 8 Ω; 500 W into 4 Ω; 800 W into 2 Ω
Music power* 2000 W into 4 Ω
Harmonic distortion <0.005%
Slew limiting 85 V µs

–1

Open-loop bandwidth 55 kHz
Power bandwidth 1.5 Hz – 220 kHz

*See text about the validity of this meaningless quantity.

AUDIO & HI-FI

Contents

INTRODUCTION

Amplifier output has been a cause of
argument for as long as there have
been audio power amplifiers. For
domestic use, a power rating of
2× 50 W is more than sufficient. With
the volume control at maximum and
the use of correctly matched good­quality loudspeakers, this will provide

a sound pressure level (SPL) equiva­lent to that of a grand piano being
played forte in the same room.

However, not all amplifiers are
intended for domestic use: many are
destined for discos, small music halls
and other large rooms. But even here,
what power is really required? Since
doubling the amplifier output increases
the SPL by a barely audible 3 dB, it was
felt that 300 watts sine wave power into
8Ωwould appeal to many .

‘ PROGRAMMABLE’

POWER OUTPUT

The amplifier has been designed in
such a manner that its output is ‘pro­grammable’ as it were. With a sine
wave input, it delivers an average
power of 300 W into an 8Ω load,
which should meet the requirements
of all but the power drunk. Compared

with the output of 50 W from a
domestic audio amplifier, this
gives an increase in SPL of

7.5 dB. If even higher outputs are
needed, the load impedance may be
lowered to 4 Ω, which will give an
increase in SPL of 10 dB compared
with a 50 W output.

Although music power is a depre­catory term, since it does not really
give the true power rating of an ampli­fier, readers may note that the Titan
2000 can deliver 2 kW of this magical
power into 4 Ω. (True power is average

power , that is, the product of the r.m.s. volt-

age across the loudspeaker and the r.m.s.
current flowing into the speaker. The term
music power is generally meaningless,
because to some manufacturers it means the
product of the peak voltage and peak cur­rent; to others it means merely double the
true power; and to yet others, even more
disreputable, it means quadrupling the true
power).

However, power is not the only cri­terion of an amplifier. Low distortion,
good slew limiting, and an extended
power bandwidth, as possessed by the
Titan 2000, are also hallmarks of a good
amplifier.

Power bandwidth denotes the fre­quency range over which the power
falls to not less than half its maximum
value. This is much more telling than
the frequency response, which is usu­ally measured at a much lower output
level.

Slew limiting is the maximum input
voltage change that can occur in one

59

Elektor Electronics 2/99

1

Figure 1. Simplified block dia­gram of the Titan 2000. The aux­iliary power supply, protection
networks and thermal control
are discrete circuits built on dis­crete PCBs.

Contents

regulator

± 78V

T43...T52

voltage amplifier

T1...T10

T15...T26

input stages &

cascode amplifiers

± 85V

auxiliary

power supply

2x 15V

current amplifier

T27...T34
T35...T42

drivers & output stages

offset control

main power

supply

± 70V

heat sink

sensor

fan

protection

circuits

UinU

thermal

control

out

990001 - 12

0

60

Elektor Electronics 2/99

Figure 2. Although the circuit diagram gives the impres­sion of a highly complex design, the amplifier is, in
essence, fairly straightforward.

2

Contents

R56

85V

BF871

BD711

BC550

BF245A BC639

R62

R57

15Ω

BF872

BD712

BC560

BC640

BF256C

R26

R25

R24

D4

R18

R13

T47

330Ω

15k

1V

2R

1R

K1

2SC5359

1M

C24

2µ2 63V

1N4004

T24...T26=MJE340

R17

-39V

T51

2x

15n

T50

D11

220n

100V

470µ

2SA1987

2SC5171

MJE340

BD139

1µ

150Ω

R72

BC

640

T48

30V

2SA1930

MJE350

BD140

D14

2

5

7

1

C19

T18 T19 T20

C23

C22

D6

C11

R20

C5

R15

C18

-78V

C39

15k

BF

245A

1W3

0W5

12V

3

IC1

6

68Ω

0V83 0V83

68Ω

68Ω

270Ω

1V45 1V45

1k00

2µ2

63V

R67

I

+5V

2k2

C42

R78

C

EB

5V

C45

100n

C44

100n

C43

100n

T31

T30

BE

E

B

C

BE

C

S

GD

70V

C

T29...T31=2SC5171

T21...T23=MJE340

3k3

R22

T11

2mA1

63V

T46

BC

639

T45

R58

270Ω

0V36

T29

T23

T22

T21

1W

T13

53V-53V

15V

39V

12k

R64

22Ω

R61

C31

15n

22Ω

R60

C29

220n

C28

470µ 100V

C17

100p

68Ω

T17

68Ω

T16

68Ω

T15

C21

100p

C20

100p

5V6

0W5

C8

100n

270Ω

150Ω

R16

C4

2n2

1k00

C16

100p

+78V

D10

1N4004

C32

2µ2

BD712

15k

R63

2x

BF

245A

T43

D8

30V

1W3

R76

10Ω

R41

38mV32mV

10Ω

R40

10Ω

R39

560Ω

R36

BF256C

D5

R19

T9

P2

250Ω

D1

8V4

5k6

R59

39V

1n

R77

100Ω

6

8

IC2

2

T35...T38=2SC5359

C9

10k

C10

BF871

R5

330Ω

C6

1V71V7

C34

C33

P4

T44

BF256A

D9

C30

100Ω

5

7

6N136

3

2Ω2

R79

100Ω

R74

33Ω

R75

0Ω22

25V

100V

1W3

63V

R48

0Ω22

R47

0Ω22

R46

0Ω22

R45

BD

139

T27

220Ω

R33

22k

R31

1n

C3

T3

5k

22Ω

R12

53mV53mV

22Ω

R4

T7

BF

245A

JP2

D3

1N4148

T38

T37

T36

T35

15V

1W3

100n

1W

100µ

T5

220µ

25V

470µ

220n

39V

47µ

MUTE

D18D19

L1

Re4 Re3 Re2

LS+ LS+

0Ω22

R52

20mV20mV

0Ω22

R51

0Ω22

R50

0Ω22

R49

150Ω

R38

R30RC14

C15

100n

2V24

R34

500Ω

470Ω

P3

470Ω

R11

470Ω

JP1

R10

45mV45mV

T1

R2
C1

Re1

390Ω

R9

T2

562Ω

2µ2

5k

P1

22Ω1

R8

22Ω

R14

R6

V23042-A2003-B101

LS1

T39...T42=2SA1987

T39 T40 T41 T42

C

T28

T4

22Ω

T6

T8

BF

245A

47k

R3

C2

1n

1M

R1

P-IN

T27...T42 on common heatsink

LS- LS-

10Ω

R44

10Ω

R43

10Ω

R42

560Ω

R37

BD

140

220Ω

R35

22k

R32

15V

1W3

D7

C12

100n

10k

1W

R21

C13

25V

100µ

T10

470Ω

R7

C7

C41

C40

P5

5k6

R68

39V

D12

C37

220µ

D2

25V

470µ

220n

47µ

C48

C47

C46

BF872

100V

5k

BF256A

1W3

63V

100n

100n

100n

T32 T33 T34

T14

15V

-30V

T49

-39V

70V

P-LS

R53

C26

D16

T32...T34=2SA1930

15V

T24 T25 T26

1W

BF256C

3k3

R23

35V 35V

T12

12k

R73

22Ω

R70

C38

22Ω

R69

C36

C35

D15

4

100p

R29

R28

R27

100p

100p

5V6

100n

2n2

100p

D13

BD711

270Ω

0V36

E

C

B

990001 - 11

E

C

B

B

C

E

B

C

E

C27

2µ2 63V

OP90G

D17

1N4004

15V

4M7

R54

C25

68n

4M7

R55

0W5

T1, T4, T5, T15...T17 = BC560C

T2, T3, T6, T18...T20 = BC550C

1N4004

T52

330Ω

R71

15k

R66

1V

15Ω

R65

85V

microsecond, and to which the ampli­fier can respond.

DESIGN
CONSIDERATIONS

The Titan 2000 is based on the ‘com­pact power amplifier’ published in the
May 1997 issue of this magazine. That
was a typical domestic amplifier with a
power output of 50 W into 8 Ω or 85 W
into 4 Ω. The special property of this
fully balanced design was the use of
current feedback instead of voltage
feedback, which resulted in a fast­responding amplifier with a large
open-loop bandwidth. The amplifier
performed well both as regards instru­ment test and measurements and lis­tening tests. However, to serve as a
basis for the Titan 2000, its output cur­rent and drive voltage range had to be
increased substantially.

For a start, the supply voltage has to
be more than doubled, which means
that transistors with a higher power
rating have to be used in the power
supply . The higher supply voltage also
results in larger potential drops across a
number of components, and this
means that dissipation problems may
arise.

The large output current required
for the Titan 2000 makes a complete
redesign of the current amplifier used
in the ‘compact power amplifier’
unavoidable, since that uses insulated­gate bipolar transistors (IGBTs).
Although these are excellent devices,
the large spread of their gate-emitter
voltage makes their use in parallel net-

works next to impossible. To obtain the
requisite output power, the use of par­allel networks of symmetrical pairs of
transistors is inevitable.

In view of the foregoing, bipolar
transistors are used in the current
amplifier of the Titan 2000. However,
these cannot be driven as readily as
IGBTs, which means that current drive
instead of voltage drive is used. This
entails a substantial upgrading of the
driver stages and the preceding cas­code amplifiers (which also consist of a
couple of parallel-connected transis­tors). The good news is that the power
transistors in the Titan 2000 are consid­erably less expensive than IGBTs: an
important factor when eight of these
devices are used.

Finally, the protection circuits have
been enhanced in view of the higher
voltages and currents. The circuits pro­tecting against direct voltages and
short-circuits are supplemented by
networks protecting against overload
and (too) high temperatures. The latter
is coupled to a proportional fan con­trol.

In short, a large part of the Titan
2000 is a virtually new design rather
than a modified one.

BRIEF DESCRIPTION

The block diagram of the Titan 2000 is
shown in Figure 1. The voltage ampli­fier consists of input stages T

1–T10

, and

cascode amplifiers/pre-drivers T

15–T26

.
The current amplifier is formed by dri­ver transistors T

27–T34

, and output

transistors T35–T42.

The offset control stage prevents
any direct voltage appearing at the
output of the amplifier.

The loudspeaker is linked to the
amplifier by three heavy-duty relays.

The current amplifier operates from
a ±70 V supply, which is provided by
two 50 V mains transformers. To enable
the voltage amplifier to drive the cur­rent amplifier to its full extent, it needs
a slightly higher supply voltage to
compensate for the inevitable losses
caused by inevitable voltage drops.
This is accomplished by superimposing
a ±15 V potential from an external
auxiliary supply on to the main ±70 V
supply and dropping the resulting
voltage to ±78 V with the aid of regu­lator T

43–T52

.

The combined protection circuits
constantly compare the input and out­put voltage of the amplifier: any devi­ation from the nominal values leads to
the output relays disconnecting the
loudspeaker and the input relay
decoupling the input signal.

The thermal protection circuit mon­itors the temperature of the heat sink
and, if necessary, switches on a fan. If,
with the fan operating, the tempera­ture approaches the maximum per­missible limit, the output relays are
deenergized and disconnect the loud­speaker.

CIRCUIT DESCRIPTION

The circuit diagram of the Titan 2000 is
shown in Figure 2. In spite of the large
number of components, the basic cir-

61

Elektor Electronics 2/99

Contents

cuit is straightforward.

As already noted in
the previous para­graph, transistors
T

1–T10

form the input

amplifier, T

11

and T12are buffers, T

13

and T14are current sources, T15–T

26

form the cascode amplifier/pre-driver
stage, T

27–T34

are the driver transistors

in the current amplifier , T

35–T42

are the

output transistors, and T

43–T52

form a

sophisticated supply voltage regulator.

Input amplifier

Strictly speaking, the input amplifier is
formed by transistors T

3–T4

. Cascode

stages T

9–T10

serve merely to enable
the input section handling the high
voltages. These voltages are limited by
zener diodes D

5

and D7, which are part
of the potential divider that also sets
the operating points of T

21–T26

. In
view of the requisite stability, the cur­rent through the zener diodes is held
constant by current sources T

13

and
T

14

. Resistors R22and R23limit the
potential across, and thus the dissipa­tion in, these field-effect transistors.

Otherwise, the input section is vir­tually identical to that of the ‘compact
power amplifier’. The drop across the
emitter resistors of buffers T

1

and T

2

determines the drop across the emitter
resistors of T

3

and T4, and conse­quently the setting of the operating
point of the overall input section. To
eliminate the influence of temperature
variations, T

1

is thermally coupled to

T

3

and T2to T4.

Since the operating point of buffers

T

1

and T2is critical, current sources T

5

and T6have been added. The reference
for these current sources is provided
by light-emitting diodes (LEDs) D

1

and

D

2

. The current through these diodes

is determined by current sources T

7

and T8. In view of the
requisite stability,
diode D

1

is thermally

coupled to T

5

and D

2

to T6.

Any imbalance of the input stages is
compensated by making the current
through T

5

equal to that through T

6

with potentiometer P2.

Cascode amplifiers/pre-drivers

The large output current of the Titan
2000 necessitates a proportionally
large pre-drive voltage, which is pro­vided by three parallel-connected cas­code amplifiers, T

15–T26

. The current
through these amplifiers is arranged at
10–15 mA, but the current feedback
used may cause this level to be appre­ciably higher. This is the reason that the
transistors used in the T

21–T26

posi­tions are types that can handle cur­rents of up to 50 mA when their collec­tor-emitter voltage is 150V.

The input section is linked to the

cascode amplifiers by buffers T

11

and

T

12

, which results in a lowering of the
input impedance. The arrangement
also enables an increase in the values
of R

13

and R15, which results in a 3 dB
increase in amplification of the input
section.

The function of resistors R

19

and

R

21

is threefold: they limit the dissipa­tion of the buffers; they obviate the
need of an additional voltage to set the
operating point of the buffers; they
limit the maximum current through
the buffers, and thus the cascode
amplifiers, to a safe value.

The open-loop amplification of the
Titan 2000 is determined solely by
those of the input section and cascode
amplifiers. The amplification of the
input section depends on the ratios
R

13

:(R12+R8) and R15:(R14+R8) and,

with values as specified is ×10 (i.e., a

gain of 20 dB).

The amplification of the cascode
amplifiers is determined largely by the
ratio of parallel-connected resistors R

31

and R32and the parallel network of
R

24–R26

. With values as specified, the
amplification is about ×850 (remem-
ber, this is a push-pull design), so that
the overall amplification of input sec­tion plus cascode amplifiers is ×8500 (a
gain of close to 80 dB).

Current amplifier

Since one of the design requirements is
that the amplifier is to work with loads
down to 1.5 Ω, the output stages con­sist of four parallel-connected pairs of
transistors, T

35–T38

and T39–T42. These
transistors have a highly linear transfer
characteristic and provide a direct-cur­rent amplification that remains virtu­ally constant for currents up to 7 A.

Like the output transistors, the dri­ver stages need to remain within their
safe operating area (SOA), which
necessitates a threefold parallel net­work. The transistors used in the dri­ver stages are fast types
(f

T

=200 MHz).

Setting the bias voltage for the req­uisite quiescent current is accom­plished by balanced transistors T

27

and

T

28

. These transistors are mounted on
the same heat sink as the output tran­sistors and driver transistors to ensure
good thermal coupling and current
control. Of course, the current rises
during full drive conditions, but drops
again to its nominal level when the
amplifier cools off. The quiescent cur­rent is set to 200 mA with potentiome­ter P

3

.

Owing to the large output current,
the connection between amplifier out­put and loudspeaker is not arranged
via a single relay, but via three. Two of
these, Re

3

–Re4, are controlled in syn­chrony by the protection circuits.
When they are deenergized, their dis­abling action is delayed slightly to give
the contacts of the third relay , Re

2

, time
to open, which is of importance in a
fault situation.

Input relay Re

1

is switched off in

synchrony with Re

2

to ensure that

there is no input signal by the time Re

3

and Re4are deenergized.

Optoisolator IC

2

serves as sensor for
the current protection circuits. The
light-emitting diode in it monitors the
voltage across R

48–R52

via potential

divider R

74–R75

, so that the positive as
well as the negative output currents
are guarded. The use of an optoisola­tor prevents earth loops and obviates
compensation of the ±70 V common­mode voltage. The +5 V supply for the
optoisolator is derived from the pro­tection circuits.

Feedback

The feedback loop runs from the out-

62

Elektor Electronics 2/99

3

Figure 3. Circuit dia­gram of the requisite
auxiliary power supply.

Contents

K1

12V / 1VA5

70V

Tr1

D3

D4

4x 1N4007

K2

D1

C3

100n

160mA T

R1

1M

D2

C1

470µ
100V

K3

F1

85V

70V

4x 1N4007

Tr2

D5

D6

12V / 1VA5

C4

100n

R2

1M

K4

F2

160mA T

990001 - 13

C2

470µ
100V

D7

D8

85V

put of the power stages to the junction
of T

3

and T4via resistors R10and R11.
This is current feedback because the
current through T

3

and T4depends on

the potential across R

8

, which is deter­mined largely by the current through
R

10

and R11. The overall voltage ampli­fication of the output amplifier is deter­mined by the ratio R

8

:(R10+R11).

Compensation

Capacitors C3–C5and resistors R16, R

17

form part of the compensation net­work required for stable operation.

Low-pass filter R

2–C2

at the input is
essential to prevent fast, that is, high­frequency, signals causing distortion.
This filter is also indispensable for sta­bility’s sake.

Coupling capacitor C

1

is needed
because the available offset compensa­tion network merely redresses the bias
current of the input buffers and is not
intended to block any direct voltages at
the input.

Relay Re

1

at the input enables the
input signal to be ‘switched off’. It
forms part of the overall protection
and in particular safeguards the input
section against overdrive. The overall
protection circuit will be discussed in
detail next month.

Network R

9-P1

is intended specifi­cally for adjusting the common-mode
suppression when two amplifiers are
used in a bridge arrangement. It is
needed for only one of these ampli­fiers, and may be interconnected or
disabled by jumper JP

1

as needed.

Offset compensation is provided by

integrator IC

1

, which ensures that if
there is any direct voltage at the output
of the amplifier, the operating point of
T

1-T2

is is shifted as needed to keep the
output at earth potential. The opera­tional amplifier (op amp) used draws
only a tiny current (20 µA) and has a
very small input offset (450 µV).

Supply voltage for IC

1

is taken from
the ±15 V line for the input section via
diodes D

16

and D17. This arrangement
ensures that the supply to the IC is
retained for a short while after the
main supply is switched off so that any
interference is smoothed out.

Diodes D

14

and D15safeguard the

input of IC

1

against (too) high input

voltages in fault conditions.

The values of resistors R

54

and R

55

arrange the level of the compensating
current at not more than 1 µA, which is
sufficient to nullify the difference
between the base currents of T

1

and T2.

Regulation

Although current feedback has many
advantages, it also has a serious draw­back: poor supply voltage suppression.
This makes it essential for the supply
voltage for the voltage amplifier to be
regulated. In view of the requisite high
symmetrical potential and the fact that
the unregulated voltage that serves as
input voltage can vary substantially
under the influence of the amplifier
load, two discrete low-drop regulators,
T

43–T47

and T48–T52are used.

As mentioned before, owing to

inevitable losses through potential
drops, the supply voltage for the input
section and cascode amplifiers needs to
be higher than the main ±70 V line.
Furthermore, the input voltage to the
regulators must be higher than the
wanted output voltage to ensure effec­tive regulation.

Fortunately, the current drawn by
the voltage amplifier is fairly low
(about 70 mA) so that the input voltage
to the regulators can be increased with
a simple auxiliary supply as shown in
Figure 3. This consists of two small
mains transformers,two bridge recti­fiers, D

1–D4

and D5–D8, and the neces-

sary reservoir and buffer capacitors.

The ±15 V output is linked in series
with the ±70 V line to give an unregu­lated voltage of ±85 V.

The 39 V reference is provided by
zener diode D

9

. This means that the
regulator needs to amplify the refer­ence voltage ×2 to obtain the requisite
output voltage.

The zener diode is powered by cur-

rent source T

43

, to ensure a stable ref­erence, which is additionally buffered
by C

30

.

Differential amplifier T

45-T46

, whose
operating point is set by current source
T

44

, compares the output voltage with
the reference via potential divider
R

63-R64-P4

. This shows that the output

voltage level can be set with P

4

.

Transistor T

47

is the output stage of
the regulator. The output voltage
remains stable down to 0.2 V below the
input voltage.

63

Elektor Electronics 2/99

Current-feedback

In an amplifier using voltage feedback (Figure a), the differential voltage at its inputs is multiplied by the open-loop
amplification. The feedback loop forces the output voltage to a level that, divided by network R

1-R2

, is equal to the

input voltage.

Whereas an amplifier with voltage feedback has high-impedance inputs, an amplifier with current feedback (Figure
b) has an high-impedance and a low-impedance input. Its input stage consists of a buffer with unitary gain between
the inverting and non-inverting inputs. Essentially, the inverting input is the low-impedance input. The buffer is fol­lowed by an impedance matching stage that converts the output current of the buffer into a directly proportional out­put voltage.

The current feedback loop operates as follows. When the potential at the non-inverting input rises, the inverting input
will also rise, resulting in the buffer current flowing through resistor R

1

. This current, magnified by the impedance
matching stage, will cause the output
voltage of the amplifier to rise until the
output current flowing through resistor R

2

is equal to the buffer current through R1.
The correct quiescent output voltage can
be sustained by a very small buffer cur­rent. The closed-loop amplification of the
circuit is determined by the ratio
(1+R

2

):R1.

A interesting property of an amplifier
with current feedback is that the closed­loop bandwidth is all but independent of
the closed-loop amplification, whereas
that of an amplifier with voltage feedback
becomes smaller in inverse proportion to
the closed-loop amplification – a relation
known as the gain-bandwidth product.

Contents

U

in

A(s)

ab

U

out

U

in

R2

R1

R2

A

= 1 +

v

R1

R1

R(s)

A

=1

v

I

R2

A

= 1 +

v

R2
R1

990001 - 14

U

out

65

Elektor Electronics 2/99

Resistor R57and diode D8protect T

43

against high voltage during switch-on,
while D

10

prevents current flowing
through the regulator in the wrong
direction.

Capacitors C

31

and C32enhance the

rate of operation of the regulator.

Network R

56-C28-C29

provides
additional smoothing and r.f. decou­pling of the ±85 V lines.

NEXT MONTH

Next month’s second and concluding
instalment of this article will describe
details of the protection circuits, the fan
control, and the construction of the
amplifier. The instalment will also
include detailed specifications and per­formance characteristics.

[990001-1]

Contents

Elektor Electronics 3/99

SIX FUNCTIONS

The integrated protection network con­sists of six sub-circuits:

• power-on delay

• transformer voltage sensor

• temperature sensor

• current sensor

• direct-current sensor

• overdrive sensor
The power-on delay ensures that the

relays in the amplifier are energized
50–100 milliseconds after the supply
has been switched on to prevent
switch-on clicks.

The transformer voltage sensor
reacts to the cessation of the secondary
voltage of the mains transformers to
prevent switch-off clicks and crackles.

The temperature sensor responds to
excessive heat sink temperatures, but it
should be noted that this works only in

This second of four

parts deals primarily

with the protection

network incorporated

in the amplifier. This

indispensable net-

work safeguards the

amplifier and the

loudspeakers con-

nected to it against all

kinds of error that

may arise. The net-

work is an indepen-

dent entity with its

own power supply.

32

Design by T. Giesberts

Titan 2000

Part 2: protection network

INTRODUCTION

As mentioned briefly in Part 1, exten­sive and thorough protection is a must
in an amplifier of this nature. It may
well be asked why this is so: is there
such a likelihood of mishaps arising?
Or is the amplifier so vulnerable? On
the contrary: extended tests on the
prototype have shown that the Titan
2000 is a very stable and reliable piece
of equipment. In fact, unusual means
had to be used to actuate the protec­tion circuits during these tests, since
not any standard test prompted the
amplifier into an error situation.

The extensive protection is neces­sary because by far the largest number
of mishaps occur owing to actions by
the user, not because of any shortcom­ings in the amplifier. For example, the
most robust and reliable amplifier can
not always cope with extremely high
overdrive or overload conditions.

Correction. In last month’s first part
of this article, it was stated erro­neously that the article consists of
two parts, whereas in fact it will be
described in four parts.

AUDIO & HI-FI

Contents

conjunction with the fan drive, which
is reverted to later in this article.

The current sensor monitors the
output current, while the direct-cur­rent and overdrive sensors form a
combined circuit that monitors differ­ences between the input and output
signals, and reacts to excessive direct­current levels or distortion. This circuit
is the most important and ‘intelligent’,
but also the most complex of the six.

All sensors, when actuated, react in
the same way: they cause the output
relays and the mute relay at the input
of the amplifier to be deenergized
immediately. This action causes the

input signal and the
output load to be dis­connected from the
amplifier. After the fault
causing the sensor
action has been
removed or remedied, the relevant
protection circuit is disabled, where­upon the amplifier relays are reener­gized after a short delay.

When the protection network is
actuated, a red LED lights to indicate
an error. When the fault has been
removed or remedied, the red LED
remains on, but a yellow LED flashes
to indicate that the amplifier will be

reenabled shortly. The red LED then
goes out, shortly followed by the yel­low, whereupon a green LED lights to
indicate that all is well.

COMMON SECTION
AND POWER

- ON DELAY

The circuit of the integrated protection
network, including the +5 V and ±12 V
power supplies, is shown in Figure 4.

33

Elektor Electronics 3/99

V

V

4

Figure 4. The protection network con­sists of six sensor circuits each of which
causes the input and output relays of
the amplifier to be deenergized when a
fault occurs.

Contents

LSP

input

C1

100n

R1

R5

680Ω

30µH

2x 100n
250V

100k

IC1

P2

C2

C16

100n

8

4

500Ω

R2

1k05

1n

16

IC3

P1
250Ω

8

C12

100n

C13

100n

R6

820k

K4

P3
500k

12V

12V
12V

12V

C17

100n

C14

100n

C15

100n

D1

2x

BAT82

D2

D3

2x

BAS45A

D4

16

IC4

8

F1

50mA T

Tr1

3

2

5V

2x 15V

8VA

IC2

IC1a

R7

11

6

1M

12V

12V

C3

100n

IC1 = OP249GP
IC2 = LM319

5

IC1b

6

R4

10k0

R3

10k0

1

R14

2M2

R13

470k

C7

470n
R15

1k

D6

EARLY

B80C1500

C26

47n

temp

I

11
12
13

IC2a

IC2b

3
4
5
6
7
8
9

5V

R12

C6

2k2

100n

12

3

7

8

4

5

12

13

5V

7
5
4
6
14
13
15
1
2
3

C8

100n

9

1

12V

R8

47k

4

R9

C4

100n

7

C5

100n

11
10

9

12

12V

RCX
RX
CX

CT=0

IC3

470Ω

R10

470Ω

R11

47k

CTR14

!G

D5

+

5

9

10

1N4148

CT

74HC4060

IC4

74HC175

1D

C1

R

5

4

K1

IC5

4N35

R16

4k7

126

D7

R19

47k

R24

47k

47k

R25

R20
2M7

C9

4µ7
63V

R23

4k7

2k2

R22
R21

2k2

T1

BC

R27

4k7

4k7

R26

T3

BC

547B

547B

T2

T4

R28

BD
140

BD
140

3k9

D9

ON

1N
4148

5V

D8

ERROR

R17

1k

2
3
7
6
10
11
15
14

R18
47k

Vre

12V

JP1

Vre Ext.

K2

K3

int
ext

2R

1R

mute

IC9

12V

+5V

5V

+5V

+12

+12V

4µ7

C18

4µ7
63V

C19

4µ7
63V

C22

63V

+12V

+5V

D13

R35

3k3

+5V

+12V

POWER

50V

50V

D10

D11

C10

10µ
63V

2x

1N4007

R32

15k

R31

15k

R30

3k3

IC6

126

4N35

R29

47k

5

4

5V

C11

47µ 25V

BC

547B

R34

T6

100k

R33

D12

100k

1N4001

T5

2x

7805

R36

C23

22Ω

47µ

C20

1000µ
25V

C21

470µ
25V

25V

IC7

7812

IC8

C24

47n

B1

C25

47n

7912

–12V

–12V

–12V

12V

–12

990001 - 2 - 11