Datasheet UC3848N, UC3848DWTR, UC3848DW, UC2848DW, UC2848N Datasheet (Texas Instruments)
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UC1848
UC2848
UC3848
4/96
FEATURES
• Practical Primary Side Control of
Isolated Power Supplies with DC
Control of Secondary Side Current
• Accurate Programmable Maximum
Duty Cycle Clamp
• Maximum Volt-Second Product Clamp
to Prevent Core Saturation
• Practical Operation Up to 1MHz
• High Current (2A Pk) Totem Pole
Output Driver
• Wide Bandwidth (8MHz) Current Error
Amplifier
• Under Voltage Lockout Monitors VCC,
VIN and VREF
• Output Active Low During UVLO
• Low Startup Current (500µA)
• Precision 5V Reference (1%)
The UC3848 family of PWM control ICs makes primary
side average current mode control practical for isolated
switching converters. Average current mode control insures that both cycle by cycle peak switch current and
maximum average inductor current are well defined and
will not run away in a short circuit situation. The UC3848
can be used to control a wide variety of converter topologies.
In addition to the basic functions required for pulse width
modulation, the UC3848 implements a patented technique of sensing secondary current in an isolated buck
derived converter from the primary side. A current waveform synthesizer monitors switch current and simulates
the inductor current down slope so that the complete current waveform can be constructed on the primary side
without actual secondary side measurement. This information on the primary side allows for full DC control of
output current.
The UC3848 circuitry includes a precision reference, a
wide bandwidth error amplifier for average current control, an oscillator to generate the system clock, latching
PWM comparator and logic circuits, and a high current
output driver. The current error amplifier easily interfaces
with an optoisolator from a secondary side voltage sensing circuit.
A full featured undervoltage lockout (UVLO) circuit is contained in the UC3848. UVLO monitors the supply voltage
to the controller (VCC), the reference voltage (VREF),
and the input line voltage (VIN). All three must be good
before soft start commences. If either VCC or VIN is low,
the supply current required by the chip is only 500µA and
the output is actively held low.
Two on board protection features set controlled limits to
prevent transformer core saturation. Input voltage is monitored and pulse width is constrained to limit the maximum volt-second product applied to the transformer. A
unique patented technique limits maximum duty cycle
within 3% of a user programmed value.
These two features allow for more optimal use of transformers and switches, resulting in reduced system size
and cost.
Patents embodied in the UC3848 belong to Lambda
Electronics Incorporated and are licensed for use in applications employing these devices.
DESCRIPTION
BLOCK DIAGRAM
Average Current Mode PWM Controller
UDG-93003-1
2
UC1848
UC2848
UC3848
Supply Voltage (Pin 15). . . . . . . . . . . . . . . . . . . . . . . . . . . . 22V
Output Current, Source or Sink (Pin 14)
DC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0.5A
Pulse (0.5 s) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2A
Power Ground to Ground (Pin 1 to Pin 13) . . . . . . . . . . . ± 0.2V
Analog Input Voltages
(Pins 3, 4, 7, 8, 12, 16). . . . . . . . . . . . . . . . . . . . . –0.3 to 7V
Analog Input Currents, Source or Sink
(Pins 3, 4, 7, 8, 11, 12, 16) . . . . . . . . . . . . . . . . . . . . . . 1mA
Analog Output Currents, Source or Sink (Pins 5 & 10) . . . 5mA
Power Dissipation at TA = 60°C. . . . . . . . . . . . . . . . . . . . . . 1W
Storage Temperature Range. . . . . . . . . . . . . . .−65°C to +150°C
Lead Temperature (Soldering 10 seconds) . . . . . . . . . . +300°C
Notes: All voltages are with respect to ground (DIL and SOIC
Pin 1). Currents are positive into the specified terminal. Pin
numbers refer to the 16 pin DIL and SOIC packages. Consult
Packaging Section of Databook for thermal limitations and
considerations of packages.
ABSOLUTE MAXIMUM RATINGS
ELECTRICAL CHARACTERISTICS:
Unless otherwise stated, all specifications are over the junction temperature range
of −55°C to +125°C for the UC1848, −40°C to +85°C for the UC2848, and 0°C to +70°C for the UC3848. Test conditions are: VCC
= 12V, CT = 400pF, CI = 100pF, IOFF = 100µA, CDC = 100nF, Cvs = 100pF, and Ivs = 400µA, TA = TJ.
PARAMETER TEST CONDITIONS MIN TYP MAX UNITS
Real Time Current Waveform Synthesizer
Ion Amplifier
Offset Voltage 0.95 1 1.05 V
Slew Rate (Note 1) 20 25 V/µs
lib -2 -20 µA
IOFF Current Mirror
Input Voltage 0.95 1 1.05 V
Current Gain 0.9 1 1.1 A/A
Current Error Amplifier
A
VOL 60 100 dB
Vio 12V ≤ VCC 20V, 0V VCM 5V 10 mV
lib -0.5 -3 µA
Voh IO = −200µA 3 3.3 V
Vol I
O = 200µA 0.3 0.6 V
Source Current VO = 1V 1.4 1.6 2.0 mA
GBW Product f = 200kHz 5 8 MHz
Slew Rate (Note 1) 810 V/µs
PACKAGE PIN FUNCTION
FUNCTION PIN
N/C 1
GND 2
VREF 3
NI 4
INV 5
N/C 6
CAO 7
CT 8
VS 9
DMAX 10
N/C 11
CDC 12
CI 13
IOFF 14
ION 15
N/C 16
PGND 17
OUT 18
VCC 19
UV 20
PLCC-20 & LCC-20
(Top View)
Q & L Packages
CONNECTION DIAGRAMS
DIL-16, SOIC-16 (Top View)
J, N, or DW Packages
3
UC1848
UC2848
UC3848
ELECTRICAL CHARACTERISTICS:
Unless otherwise stated, all specifications are over the junction temperature range
of −55°C to +125°C for the UC1848, −40°C to +85°C for the UC2848, and 0°C to +70°C for the UC3848. Test conditions are: VCC
= 12V, CT = 400pF, CI = 100pF, IOFF = 100µA, CDC = 100nF, Cvs = 100pF, and Ivs = 400µA, TA = TJ.
PARAMETER TEST CONDITIONS MIN TYP MAX UNITS
Oscillator
Frequency T
A = 25°C 240 250 260 kHz
235 265 kHz
Ramp Amplitude 1.5 1.65 1.8 V
Duty Cycle Clamp
Max Duty Cycle V(D
MAX) = 0.75 • VREF 73.5 76.5 79.5 %
Volt Second Clamp
Max On Time 900 1100 ns
VCC Comparator
Turn-on Threshold 13 14 V
Turn-off Threshold 910 V
Hysteresis 2.5 3 3.5 V
UV Comparator
Turn-on Threshold 4.1 4.35 4.6 V
R
HYSTERESIS Vuv = 4.2V 77 90 103 kΩ
Reference
VREF T
A = 25°C 4.95 5 5.05 V
0 < IO < 10mA, 12 < VCC < 20 4.93 5.07 V
Line Regulation 12 < VCC < 20V 4 15 mV
Load Regulation 0 < IO < 10mA 3 15 mV
Short Circuit Current V
REF = 0V 30 50 70 mA
Output Stage
Rise & Fall Time (Note 1) Cl = 1nF 20 45 ns
Output Low Saturation I
O = 20mA 0.25 0.4 V
IO = 200mA 1.2 2.2 V
Output High Saturation IO = -200mA 2.0 3.0 V
UVLO Output Low Saturation IO = 20mA 0.8 1.2 V
I
CC
ISTART VCC = 12V 0.2 0.4 mA
ICC (pre-start) VCC = 15V, V(UV) = 0 0.5 1 mA
ICC (run) 22 26 mA
Note 1: Guaranteed by design.
Under Voltage Lockout
The Under Voltage Lockout block diagram is shown in
Fig 1. The VCC comparator monitors chip supply voltage.
Hysteretic thresholds are set at 13V and 10V to facilitate
off-line applications. If the VCC comparator is low, ICC is
low (<500µA) and the output is low.
The UV comparator monitors input line voltage (V
IN
). A
pair of resistors divides the input line to UV. Hysteretic in-
put line thresholds are programmed by Rv1 and Rv2.
The thresholds are
VIN(on) = 4.35V • (1 + Rv1/Rv2′) and
VIN(off) = 4.35V • (1 + Rv1/Rv2) where
Rv2′= Rv2||90k.
The resulting hysteresis is
V
IN
(hys) = 4.35V • Rv1 / 90k.
When the UV comparator is low, ICCis low (500µA) and
the output is low.
APPLICATION INFORMATION
4
UC1848
UC2848
UC3848
Figure 1: Under voltage lockout.
UDG-93004
Figure 2: Oscillator frequency.
10
100
1000
10000
10 100 1000 10000
C (pF)
FREQUENCY (kHz)
Oscillator Frequency as a Function of CT
Frequency Decrease as a Function of RT
RT = Open
UDG-93006
UDG-93005
When both the UV and VCC comparators are high, the
internal bias circuitry for the rest of the chip is activated.
The CDC pin (see discussion on Maximum Duty Cycle
Control and Soft Start) and the Output are held low until
VREF exceeds the 4.5V threshold of the VREF comparator. When VREF is good, control of the output driver
is transferred to the PWM circuitry and CDC is allowed to
charge.
If any of the three UVLO comparators go low, the UVLO
latch is set, the output is held low, and CDC is discharged. This state will be maintained until all three comparators are high and the CDC pin is fully discharged.
APPLICATION INFORMATION (cont.)
5
UC1848
UC2848
UC3848
Oscillator
A capacitor from the CT pin to GND programs oscillator
frequency, as shown in Fig. 2. Frequency is determined
by:
F = 1 / (10k • CT).
The sawtooth wave shape is generated by a charging
current of 200
A and a discharge current of 1800 A. The
discharge time of the sawtooth is guaranteed dead time
for the output driver. If the maximum duty cycle control is
defeated by connecting DMAX to VREF, the maximum
duty cycle is limited by the oscillator to 90%. If an adjustment is required, an additional trim resistor RT from CT to
Ground can be used to adjust the oscillator frequency. RT
should not be less than 40k
. This will allow up to a 22%
decrease in frequency.
Inductor Current Waveform Synthesizer
Average current mode control is a very useful technique
to control the value of any current within a switching converter. Input current, output inductor current, switch current, diode current or almost any other current can be
controlled. In order to implement average current mode
control, the value of the current must be explicitly known
at all times. To control output inductor current (IL) in a
buck derived isolated converter, switch current provides
inductor current information, but only during the on time
of the switch. During the off time, switch current drops
abruptly to zero, but the inductor current actually diminishes with a slope dIL/dt = –V
O
/L. This down slope must
be synthesized in some manner on the primary side to
provide the entire inductor current waveform for the control circuit.
The patented current waveform synthesizer (Fig. 4) consists of a unidirectional voltage follower which forces the
voltage on capacitor CI to follow the on time switch current waveform. A programmable discharge current synthesizes the off time portion of the waveform. ION is the
input to the follower. The discharge current is programmed at IOFF.
The follower has a one volt offset, so that zero current
corresponds to one volt at CI. The best utilization of the
UC3848 is to translate maximum average inductor current to a 4V signal level. Given N and Ns (the turns ratio
of the power and current sense transformers), proper
scaling of IL to V(CI) requires a sense resistor Rs as calculated from:
Rs = 4V • Ns • N / IL(max).
Restated, the maximum average inductor current will be
limited to:
IL(max) = 4V • Ns • N/Rs.
IOFF and CI need to be chosen so that the ratio of
dV(CI)/dt to dIL/dt is the same during switch off time as
on time. Recommended nominal off current is 100
A.
This requires
CI = (100µA • N • Ns • L) / (Rs • V
O
(nom))
where L is the output inductor value and V
O
(nom) is the
converter regulated output voltage.
There are several methods to program IOFF. If accurate
average current control is required during short circuit operation, IOFF must track output voltage. The method
shown in Fig. 4 derives a voltage proportional to V
IN
• D
(Duty Cycle). (In a buck converter, output voltage is proportional to VIN • D.) A resistively loaded diode connection to the bootstrap winding yields a square wave whose
amplitude is proportional to VIN and is duty cycle modulated by the control circuit. Averaging this waveform with
a filter generates a primary side replica of secondary regulated V
O
. A single pole filter is shown, but in practice a
two or three pole filter provides better transient response.
Filtered voltage is converted by ROFF to a current to the
IOFF pin to control CI down slope.
Figure 3: Error amplifier gain and phase response
over frequency.
APPLICATION INFORMATION (cont.)
UDG-93008-1
6
UC1848
UC2848
If the system is not sensitive to short circuit requirements,
Figure 5 shows the simplest method of downslope generation: a single resistor (ROFF = 40k) from IOFF to VREF.
The discharge current is then 100µA. The disadvantage
to this approach is that the synthesizer continues to generate a down slope when the switch is off even during
short circuit conditions. Actual inductor down slope is
closer to zero during a short circuit. The penalty is that
the average current is understated by an amount approximately equal to the nominal inductor ripple current. Output short circuit is therefore higher than the designed
maximum output current.
A third method of generating IOFF is to add a second
winding to the output inductor core (Fig. 6). When the
power switch is off and inductor current flows in the free
wheeling diode, the voltage across the inductor is equal
to the output voltage plus the diode drop. This voltage is
then transformed by the second winding to the primary
side of the converter. The advantages to this approach
are its inherent accuracy and bandwidth. Winding the
second coil on the output inductor core while maintaining
the required isolation makes this a more costly solution.
In the example, ROFF = V
O
/ 100µA. The 4 • ROFF resistor is added to compensate the one volt input level of
the IOFF pin. Without this compensation, a minor current
foldback behavior will be observed.
APPLICATION INFORMATION (cont.)
Figure 5: Fixed IOFF. Figure 6: Second inductor winding generation of
IOFF.
UDG-93010
UDG-93011
Figure 4: Inductor current waveform synthesizer.
UDG-93009
7
UC1848
UC2848
UC3848
Figure 7: Duty cycle control.
UDG-93012-1
UDG-93013-1
Maximum Volt-Second Circuit
A maximum volt-second product can be programmed by
a resistor (Rvs) from VS to VIN and a capacitor (Cvs)
from VS to ground (Figure 7). VS is discharged while the
switch is off. When the output turns on, VS is allowed to
charge. Since the threshold of the VS comparator is
much less than VIN, the charging profile at Vs will be essentially linear. If VS crosses the 4.0V threshold before
the PWM turns the output off, the VS comparator will turn
the output off for the remainder of the cycle. The maximum volt-second product is
V
IN
• TON(max) = 4.0V • Rvs • Cvs.
Maximum Duty Cycle And Soft Start
A patented technique is used to accurately program maximum duty cycle. Programming is accomplished by a divider from VREF to DMAX (Fig. 7). The value
programmed is:
D(max) = Rd1 / (Rd1 + Rd2).
For proper operation, the integrating capacitor, C
DC
,
should be larger than C
DC
(min) >T(osc) / 80k, where
T(osc) is the oscillator period. CDCalso sets the soft start
time constant, so values of C
DC
larger than minimum
may be desired. The soft start time constant is approximately:
T(ss) = 20k • C
DC
.
Ground Planes
The output driver on the UC3848 is capable of 2A peak
currents. Careful layout is essential for correct operation
of the chip. A ground plane must be employed (Fig. 8). A
unique section of the ground plane must be designated
for high di/dt currents associated with the output stage.
This point is the power ground to which to PGND pin is
connected. Power ground can be separated from the rest
of the ground plane and connected at a single point, although this is not strictly necessary if the high di/dt paths
are well understood and accounted for. VCC should be
bypassed directly to power ground with a good high frequency capacitor. The sources of the power MOSFET
should connect to power ground as should the return
connection for input power to the system and the bulk input capacitor. The output should be clamped with a high
current Schottky diode to both VCC and PGND. Nothing
else should be connected to power ground.
VREF should be bypassed directly to the signal portion of
the ground plane with a good high frequency capacitor.
Low esr/esl ceramic 1
F capacitors are recommended
for both VCC and VREF. The capacitors from CT, CDC,
and CI should likewise be connected to the signal ground
plane.
APPLICATION INFORMATION (cont.)
8
UC1848
UC2848
UC3848
UNITRODE CORPORATION
7 CONTINENTALBLVD. • MERRIMACK, NH 03054
TEL. (603) 424-2410 FAX (603) 424-3460
Figure 8: Ground plane considerations.
Figure 9: Typical application - an average current-mode isolated forward converter.
UDG-93014
UDG-93015
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