Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
TP A0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
1
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
D
Integrated Depop Circuitry
D
High Power with PC Power Supply
– 2 W/Ch at 5 V into a 3-Ω Load
– 800 mW/Ch at 3 V
D
Fully Specified for Use With 3-Ω Loads
D
Ultra-Low Distortion
– 0.05% THD+N at 2 W and 3-Ω Load
D
Bridge-Tied Load (BTL) or Single-Ended
(SE) Modes
D
Stereo Input MUX
D
Surface-Mount Power Package
24-Pin TSSOP PowerP AD
D
Shutdown Control ...IDD = 5 µA
C
B
C
S
Right
MUX
RLINEIN
RHPIN
Left
MUX
LHPIN
LLINEIN
Bias, Mute,
Shutdown,
and SE/BTL
MUX Control
+
–
+
–
RBYPASS
MUTE IN
MUTE OUT
SHUTDOWN
LBYPASS
ROUT+
ROUT–
RV
DD
LV
DD
LOUT+
LOUT–
SE/BTL
HP/LINE
C
IR
R
IR
R
FR
C
FR
System
Control
C
IL
R
IL
NC
NC
4
5
6
8
9
11
19
20
21
R
FL
C
FL
100 kΩ
100 kΩ
V
DD
V
DD
C
OUTR
C
OUTL
10
3
16
7
14
18
15
22
1 kΩ
1 kΩ
1
2
3
4
5
6
7
8
9
10
11
12
24
23
22
21
20
19
18
17
16
15
14
13
GND/HS
TJ
LOUT+
LLINEIN
LHPIN
LBYPASS
LV
DD
SHUTDOWN
MUTE OUT
LOUT–
MUTE IN
GND/HS
GND/HS
NC
ROUT+
RLINEIN
RHPIN
RBYPASS
RV
DD
NC
HP/LINE
ROUT–
SE/BTL
GND/HS
PWP PACKAGE
(TOP VIEW)
Copyright 2000, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
PowerPAD is a trademark of Texas Instruments Incorporated.
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
2
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
description
The TPA0202 is a stereo audio power amplifier in a 24-pin TSSOP thermal package capable of delivering
greater than 2 W of continuous RMS power per channel into 3-Ω loads. The TP A0202 simplifies design and frees
up board space for other features. Full power distortion levels of less than 0.1% THD+N from a 5-V supply are
typical. Low-voltage applications are also well served by the TP A0202 providing 800-mW per channel into 3-Ω
loads with a 3.3-V supply voltage.
The TP A0202 has integrated depop circuitry that virtually eliminates transients that cause noise in the speakers
during power up and when using the mute and shutdown modes.
Amplifier gain is externally configured by means of two resistors per input channel and does not require external
compensation for settings of 2 to 20 in BTL mode (1 to 10 in SE mode). An internal input MUX allows two sets
of stereo inputs to the amplifier. In notebook applications, where internal speakers are driven as BTL and the
line (often headphone drive) outputs are required to be SE, the TP A0202 automatically switches into SE mode
when the SE/BTL
input is activated. Using the TP A0202 to drive line outputs up to 700 mW/channel into external
3-Ω loads is ideal for small non-powered external speakers in portable multimedia systems. The TP A0202 also
features a shutdown function for power sensitive applications, holding the supply current at 5 µA.
The PowerP AD package
†
(PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of approximately 35°C/W are readily realized in multilayer PCB
applications. This allows the TP A0202 to operate at full power into 3- Ω loads at ambient temperature of up to
85°C with 300 CFM of forced-air cooling. Into 8-Ω loads, the operating ambient temperature increases to 100°C.
AVAILABLE OPTIONS
PACKAGE
T
A
TSSOP
‡
(PWP)
–40°C to 85°C TPA0202PWP
‡
The PWP packages are available taped and reeled. T o order a taped
and reeled part, add the suffix R (e.g., TPA0202PWPR).
†
See Texas Instruments document,
PowerPAD Thermally Enhanced Package Application Report
(Literature Number SLMA002) for more
information on the PowerPAD package.
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
3
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
Terminal Functions
TERMINAL
NAME NO.
I/O
DESCRIPTION
GND/HS 1, 12,
13, 24
Ground connection for circuitry, directly connected to thermal pad
HP/LINE 16 I Input MUX control input, hold high to select LHP IN or RHP IN (5, 20), hold low to select LLINE IN or
RLINE IN (4, 21)
LBYPASS 6 Tap to voltage divider for left channel internal mid-supply bias
LHP IN 5 I Left channel headphone input, selected when HP/LINE terminal (16) is held high
LLINE IN 4 I Left channel line input, selected when HP/LINE terminal (16) is held low
LOUT+ 3 O Left channel + output in BTL mode, + output in SE mode
LOUT– 10 O Left channel – output in BTL mode, high-impedance state in SE mode
LV
DD
7 I Supply voltage input for left channel and for primary bias circuits
MUTE IN 11 I Mute all amplifiers, hold low for normal operation, hold high to mute
MUTE OUT 9 O Follows MUTE IN terminal (11), provides buffered output
NC 17, 23 No internal connection
RBYPASS 19 Tap to voltage divider for right channel internal mid–supply bias
RHPIN 20 I Right channel headphone input, selected when HP/LINE terminal (16) is held high
RLINEIN 21 I Right channel line input, selected when HP/LINE terminal (16) is held low
ROUT+ 22 O Right channel + output in BTL mode, + output in SE mode
ROUT– 15 O Right channel – output in BTL mode, high impedance state in SE mode
RV
DD
18 I Supply voltage input for right channel
SE/BTL 14 I Hold low for BTL mode, hold high for SE mode
SHUTDOWN 8 I Places entire IC in shutdown mode when held high, I
DD
= 5 µA
TJ 2 O Sources a current proportional to the junction temperature. This terminal should be left unconnected
during normal operation. For more information, see the
junction temperature measurement
section of
this document.
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
4
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
absolute maximum ratings over operating free-air temperature range (unless otherwise noted)
†
Supply voltage, VDD 6 V. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Input voltage, VI –0.3 V to VDD +0.3 V. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Continuous total power dissipation internally limited (see Dissipation Rating Table). . . . . . . . . . . . . . . . . . . . .
Operating free-air temperature range, T
A
–40°C to 85°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Operating junction temperature range, TJ –40°C to 150°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Storage temperature range, T
stg
–65°C to 150°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds 260°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
†
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
TA ≤ 25°C DERATING FACTOR TA = 70°C TA = 85°C
PWP
‡
2.7 W 21.8 mW/°C 1.7 W 1.4 W
‡
Please see the Texas Instruments document,
PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled
T exas Instruments Recommended
Board for PowerPAD
on page 33 of the before mentioned document.
recommended operating conditions
MIN NOM MAX UNIT
Supply Voltage, V
DD
3 5 5.5 V
VDD = 5 V,
250 mW/ch average power,
4-Ω stereo BTL drive,
with proper PCB design
–40 85
Operating free-air temperature, T
A
VDD = 5 V,
2 W/ch average power,
3-Ω stereo BTL drive,
with proper PCB design
and 300 CFM forced-air
cooling
–40 85
°C
p
VDD = 5 V 1.25 4.5
Common mode input voltage, V
ICM
VDD = 3.3 V 1.25 2.7
V
dc electrical characteristics, TA = 25°C
PARAMETER TEST CONDITIONS TYP†MAX UNIT
Stereo BTL 19 25 mA
Stereo SE 9 15 mA
V
DD
= 5
V
Mono BTL 9 15 mA
pp
Mono SE 3 10 mA
IDDSupply current
Stereo BTL 13 20 mA
Stereo SE 5 10 mA
V
DD
=
3.3 V
Mono BTL 5 10 mA
Mono SE 3 6 mA
V
OO
Output offset voltage (measured differentially) VDD = 5 V, Gain = 2, See Note 1 5 25 mV
I
DD(MUTE)
Supply current in mute mode VDD = 5 V 1.5 mA
I
DD(SD)
IDD in shutdown VDD = 5 V 5 15 µA
NOTE 1: At 3 V < VDD < 5 V the dc output voltage is approximately VDD/2.
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
5
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
ac operating characteristics, V
DD
= 5 V, T
A
= 25°C, R
L
= 3 Ω (unless otherwise noted)
PARAMETER TEST CONDITIONS TYP MAX UNIT
Output power (each channel) see
THD = 0.2%, BTL, See Figure 3 2
P
O
()
Note 2
THD = 1%,
BTL, See Figure 3 2.2
W
p
Po = 2W, f = 20 – 20 kHz, See Figure 5 200 m%
THD+N
Total harmonic distortion plus noise
VI = 1 V, RL = 10 kΩ, AV = 1 V/V 100 m%
B
OM
Maximum output power bandwidth AV = 10 V/V THD < 1 %, See Figure 5 >20 kHz
Phase margin RL = 4 Ω, Open Loop, See Figure 43 85°
pp
pp
f = 1 kHz, See Figure 37 80
Supply ripple rejection ratio
f = 20 – 20 kHz, See Figure 37 60
dB
Mute attenuation 85 dB
Channel-to-channel output separation f = 1 kHz, See Figure 39 85 dB
Line/HP input separation 100 dB
BTL attenuation in SE mode 100 dB
Z
I
Input impedance 2 MΩ
Signal-to-noise ratio Po = 500 mW, BTL 95 dB
V
n
Output noise voltage See Figure 35 21 µV(rms)
NOTE 2: Output power is measured at the output terminals of the IC at 1 kHz.
ac operating characteristics, V
DD
= 3.3 V, T
A
= 25°C, R
L
= 3 Ω
PARAMETER TEST CONDITIONS TYP MAX UNIT
Output power (each channel) see
THD = 0.2%, BTL, See Figure 10 800
P
O
()
Note 2
THD = 1%,
BTL, See Figure 10 900
mW
p
Po = 800 mW, f = 20 – 20 kHz, See Figure 11 350 m%
THD+N
Total harmonic distortion plus noise
VI = 1 V, RL = 10 kΩ, AV = 1 V/V 200 m%
B
OM
Maximum output power bandwidth AV = 10 V/V THD < 1 %, See Figure 1 1 >20 kHz
Phase margin RL = 4 Ω, Open Loop, See Figure 44 85°
pp
pp
f = 1 kHz, See Figure 37 70
Supply ripple rejection ratio
f = 20 – 20 kHz, See Figure 37 55
dB
Mute attenuation 85 dB
Channel-to-channel output separation f = 1 kHz, See Figure 40 85 dB
Line/HP input separation 100 dB
BTL attenuation in SE mode 100 dB
Z
I
Input impedance 2 MΩ
Signal-to-noise ratio Po = 500 mW, BTL 95 dB
V
n
Output noise voltage See Figure 37 21 µV(rms)
NOTE 2: Output power is measured at the output terminals of the IC at 1 kHz.
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
6
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
PARAMETER MEASUREMENT INFORMATION
MUX
R
I
C
I
R
F
4.7 µF
C
B
SE/BTL
HP/LINE
RL = 3 Ω or 8 Ω
Figure 1. BTL Test Circuit
MUX
R
I
C
I
R
F
C
B
C
O
SE/BTL
HP/LINE
RL = 3 Ω, 8 Ω, or 32 Ω
4.7 µF
V
DD
Figure 2. SE Test Circuit
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
7
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
p
vs Frequency
4, 5, 7, 8, 11, 12, 14, 15, 17, 18, 20,
21, 23, 24, 26, 27, 29, 30 32, 33
THD
+
N
Total harmonic distortion plus noise
vs Output power
3, 6, 9, 10, 13, 16, 19, 22, 25, 28, 31,
34
V
n
Output noise voltage vs Frequency 35,36
Supply ripple rejection ratio vs Frequency 37,38
Crosstalk vs Frequency 39 – 42
Open loop response vs Frequency 43,44
Closed loop response vs Frequency 45, 48
I
DD
Supply current vs Supply voltage 49
P
O
Output power
vs Supply voltage
vs Load resistance
50, 51
52, 53
P
D
Power dissipation vs Output power 54 – 57
Figure 3
0.1
0.01
0 0.25 0.5 0.75 1 1.25 1.5
1
10
1.75 2 2.25 2.5
PO – Output Power – W
VDD = 5 V
f = 1 kHz
BTL
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
RL = 8 Ω
RL = 3 Ω
Figure 4
0.01
10
20 100 1 k 10 k 20 k
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
1
0.1
VDD = 5 V
PO = 1.5 W
RL = 4 Ω
BTL
AV = –2 V/V
AV = –20 V/V
AV = –10 V/V (RL = 3 Ω, PO = 2 W)
AV = –10 V/V
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
8
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 5
PO = 1.5 W
PO = 0.25 W
VDD = 5 V
RL = 4 Ω
AV = –2 V/V
BTL
0.1
0.01
20 100 1 k
1
10
10 k 20 k
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
PO = 0.75 W
PO = 2 W, RL = 3 Ω
Figure 6
f = 20 kHz
f = 1 kHz
f = 20 Hz
0.1
0.01
0.01 0.1
1
10
110
PO – Output Power – W
VDD = 5 V
RL = 3 Ω
BTL
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
Figure 7
0.1
0.01
20 100 1 k
1
10
10 k 20 k
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
PO = 1 W
VDD = 5 V
RL = 8 Ω
AV = –2 V/V
BTL
PO = 0.25 W
PO = 0.5 W
Figure 8
0.1
0.01
20 100 1 k
1
10
10 k 20 k
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 5 V
PO = 1 W
RL = 8 Ω
BTL
AV = –2 V/V
AV =– 20 V/V
AV = –10 V/V
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
9
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 9
0.1
0.01
0.01 0.1
1
10
110
f = 20 kHz
f = 1 kHz
f = 20 Hz
PO – Output Power – W
VDD = 5 V
RL = 8 Ω
AV = –2 V/V
BTL
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
Figure 10
0.1
0.01
0 0.1 0.2 0.3 0.4 0.5 0.6
1
10
0.7 0.8 0.9 1
PO – Output Power – W
VDD = 3.3 V
f = 1 kHz
BTL
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
RL = 3 Ω
RL = 8 Ω
Figure 11
0.1
0.01
20 100 1 k
1
10
10 k 20 k
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 3.3 V
PO = 0.75 W
RL = 4 Ω
BTL
AV = –10 V/V
AV = –20 V/V
AV = –2 V/V
AV = –10 V/V (RL = 3 Ω, PO = 800 mW)
Figure 12
PO = 0.35 W
PO = 0.1 W
PO = 0.75 W
0.1
0.01
20 100 1 k
1
10
10 k 20 k
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 3.3 V
RL = 4 Ω
AV = –2 V/V
BTL
PO = 800 mW
(RL = 3 Ω)
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
10
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 13
0.1
0.01
0.1
1
10
12
f = 20 kHz
f = 1 kHz
f = 20 Hz
PO – Output Power – W
VDD = 3.3 V
RL = 3 Ω
AV = –2 V/V
BTL
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
Figure 14
0.1
0.01
20 100 1 k
1
10
10 k 20 k
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
AV = –20 V/V
AV = –10 V/V
AV = –2 V/V
VDD = 3.3 V
PO = 0.4 W
RL = 8 Ω
BTL
Figure 15
PO = 0.4 W
PO = 0.25 W
PO = 0.1 W
VDD = 3.3 V
RL = 8 Ω
AV = –2 V/V
BTL
0.1
0.01
20 100 1 k
1
10
10 k 20 k
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
Figure 16
0.1
0.01
0.01 0.1
1
10
110
f = 20 kHz
f = 1 kHz
f = 20 Hz
PO – Output Power – W
VDD = 3.3 V
RL = 8 Ω
AV = –2 V/V
BTL
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
11
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 17
0.1
0.01
20 100 1 k
1
10
10 k 20 k
AV = –10 V/V
AV = –5 V/V
AV = –1 V/V
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 5 V
PO = 0.5 W
RL = 4 Ω
SE
Figure 18
0.1
0.01
20 100 1 k
1
10
10 k 20 k
PO = 0.25 W
PO = 0.1 W
PO = 0.5 W
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 5 V
RL = 4 Ω
AV = –2 V/V
SE
Figure 19
f = 20 kHz
f =100 Hz
f = 1 kHz
VDD = 5 V
RL = 4 Ω
AV = –2 V/V
SE
0.1
0.01
0.001 0.01
1
10
0.1 1
PO – Output Power – W
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
Figure 20
0.1
0.01
20 100 1 k
1
10
10 k 20 k
AV = –10 V/V
AV = –5 V/V
AV = –1 V/V
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 5 V
PO = 0.25 W
RL = 8 Ω
SE
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
12
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 21
0.1
0.01
20 100 1 k
1
10
10 k 20 k
PO = 0.25 W
PO = 0.05 W
PO = 0.1 W
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 5 V
RL = 8 Ω
SE
Figure 22
0.1
0.01
0.001 0.1
1
10
1
PO – Output Power – W
VDD = 5 V
RL = 8 Ω
AV = –2 V/V
SE
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
0.01
f = 20 kHz
f = 1 kHz
f = 100 Hz
Figure 23
0.1
0.01
20 100 1 k
1
10
10 k 20 k
AV = –10 V/V
AV = –5 V/V
AV = –1 V/V
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 5 V
PO = 0.075 W
RL = 32 Ω
SE
Figure 24
0.1
0.01
20 100 1 k
1
10
10 k 20 k
PO = 75 mW
PO = 25 mW
PO = 50 mW
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 5 V
RL = 32 Ω
SE
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
13
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 25
0.1
0.01
0.001 0.01
1
10
0.1 1
f = 20 kHz
f = 1 kHz
f = 20 Hz
PO – Output Power – W
VDD = 5 V
RL = 32 Ω
SE
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
Figure 26
0.1
0.01
20 100 1 k
1
10
10 k 20 k
AV = –10 V/V
AV = –5 V/V
AV = –1 V/V
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 3.3 V
PO = 0.2 W
RL = 4 Ω
SE
Figure 27
0.1
0.01
20 100 1 k
1
10
10 k 20 k
PO = 0.05 W
PO = 0.1 W
PO = 0.2 W
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 3.3 V
RL = 4 Ω
SE
Figure 28
f = 100 Hz
f = 1 kHz
f = 20 kHz
VDD = 3.3 V
RL = 4 Ω
AV = –2 V/V
SE
0.1
0.01
0.001 0.01
1
10
10.1
PO – Output Power – W
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
14
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 29
AV = –10 V/V
AV = –5 V/V
AV = –1 V/V
VDD = 3.3 V
PO = 100 mW
RL = 8 Ω
SE
0.1
0.01
20 100 1 k
1
10
10 k 20 k
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
Figure 30
0.1
0.01
20 100 1 k
1
10
10 k 20 k
PO = 25 mW
PO = 50 mW
PO = 100 mW
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 3.3 V
RL = 8 Ω
SE
Figure 31
VDD = 3.3 V
RL = 8 Ω
SE
0.1
0.01
0.001 0.1
1
10
1
PO – Output Power – W
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
f = 20 kHz
f = 1 kHz
f = 100 Hz
0.01
Figure 32
0.1
0.01
20 100 1 k
1
10
10 k 20 k
AV = –10 V/V
AV = –5 V/V
AV = –1 V/V
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 3.3 V
PO = 30 mW
RL = 32 Ω
SE
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
15
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 33
0.1
0.01
0.001
20 100 1 k
1
10
10 k 20 k
PO = 10 mW
PO = 20 mW
PO = 30 mW
THD+N –Total Harmonic Distortion + Noise – %
f – Frequency – Hz
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 3.3 V
RL = 32 Ω
SE
Figure 34
0.1
0.01
0.001
0.001 0.01
1
10
0.1 1
f = 20 Hz
f = 1 kHz
f = 20 kHz
PO – Output Power – W
VDD = 3.3 V
RL = 32 Ω
SE
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
Figure 35
10
1
20 100 1 k
f – Frequency – Hz
OUTPUT NOISE VOLTAGE
vs
FREQUENCY
100
10 k 20 k
VDD = 5 V
BW = 22 Hz to 22 kHz
RL = 4Ω
VO BTL
V
O+
V
O–
– Output Noise Voltage –
V
n
Vµ
(rms)
Figure 36
10
1
20 100 1 k
f – Frequency – Hz
OUTPUT NOISE VOLTAGE
vs
FREQUENCY
100
10 k 20 k
VDD = 3.3 V
BW = 22 Hz to 22 kHz
RL = 4Ω
VO BTL
V
O+
V
O–
– Output Noise Voltage –
V
n
Vµ
(rms)
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
16
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 37
–50
–60
–80
–100
20 100 1 k
–30
–20
f – Frequency – Hz
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
0
10 k 20 k
–10
–40
–70
–90
VDD = 5 V
VDD = 3.3 V
RL = 4 Ω
CB = 4.7 µF
BTL
Supply Ripple Rejection Ratio – dB
Figure 38
–50
–60
–80
–100
20 100 1 k
–30
–20
f – Frequency – Hz
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
0
10 k 20 k
–10
–40
–70
–90
VDD = 5 V
VDD = 3.3 V
RL = 4 Ω
CB = 4.7 µF
SE
Supply Ripple Rejection Ratio – dB
Figure 39
Left to Right
Right to Left
–80
–90
–110
–120
–60
–50
–40
–70
–100
20 100 1 k
Crosstalk – dB
f – Frequency – Hz
CROSSTALK
vs
FREQUENCY
10 k 20 k
VDD = 5 V
PO = 1.5 W
RL = 4 Ω
BTL
Figure 40
–80
–90
–110
–120
–60
–50
–40
–70
–100
20 100 1 k
Crosstalk – dB
f – Frequency – Hz
CROSSTALK
vs
FREQUENCY
10 k 20 k
VDD = 3.3 V
PO = 0.75 W
RL = 4 Ω
BTL
Left to Right
Right to Left
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
17
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 41
–80
–90
–110
–120
–60
–50
–40
–70
–100
20 100 1 k
Crosstalk – dB
f – Frequency – Hz
CROSSTALK
vs
FREQUENCY
10 k 20 k
VDD = 5 V
PO = 75 mW
RL = 32 Ω
SE
Left to Right
Right to Left
Figure 42
–80
–90
–110
–120
–60
–50
–40
–70
–100
20 100 1 k
Crosstalk – dB
f – Frequency – Hz
CROSSTALK
vs
FREQUENCY
10 k 20 k
VDD = 3.3 V
PO = 35 mW
RL = 32 Ω
SE
Left to Right
Right to Left
40
20
–20
–40
0.01
Gain – dB
60
80
f – Frequency – kHz
OPEN LOOP RESPONSE
100
0
0.1 1 10 100 1000 10000
180°
90°
0°
–90°
–180°
VDD = 5 V
RL = 4 Ω
BTL
Gain
Phase
Figure 43
Phase
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
18
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
20
0
–20
–40
40
60
80
180°
90°
0°
–90°
–180°
0.01
Gain – dB
f – Frequency – kHz
OPEN LOOP RESPONSE
0.1 1 10 100 1000 10000
VDD = 3.3 V
RL = 4 Ω
BTL
Gain
Figure 44
Phase
Phase
5
3
2
0
20 100 1 k 10 k
Gain – dB
7
9
f – Frequency – Hz
CLOSED LOOP RESPONSE
10
100 k 200 k
8
6
4
1
–45°
0°
–90°
–135°
–180°
–225°
–270°
Phase
Phase
Gain
VDD = 5 V
AV = –2 V/V
PO = 1.5 W
BTL
Figure 45
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
19
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
5
3
2
0
20 100 1 k 10 k
Gain – dB
7
9
f – Frequency – Hz
CLOSED LOOP RESPONSE
10
100 k 200 k
8
6
4
1
–45°
0°
–90°
–135°
–180°
–225°
–270°
Phase
Phase
Gain
VDD = 3.3 V
AV = –2 V/V
PO = 0.75 W
BTL
Figure 46
Figure 47
–5
–7
–8
–10
20 100 1 k 10 k
Gain – dB
–3
–1
f – Frequency – Hz
CLOSED LOOP RESPONSE
0
100 k 200 k
–2
–4
–6
–9
–45°
0°
–90°
–135°
–180°
–225°
–270°
Phase
VDD = 5 V
AV = –1 V/V
PO = 0.5 W
SE
Phase
Gain
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
20
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
–5
–7
–8
–10
20 100 1 k 10 k
Gain – dB
–3
–1
f – Frequency – Hz
CLOSED LOOP RESPONSE
0
100 k 200 k
–2
–4
–6
–9
–45°
0°
–90°
–135°
–180°
–225°
–270°
Phase
VDD = 3.3V
AV = –1 V/V
PO = 0.25 W
SE
Phase
Gain
Figure 48
Figure 49
Stereo BTL
15
10
5
0
3
20
25
SUPPLY CURRENT
vs
SUPPLY VOLTAGE
30
465
VDD – Supply Voltage – V
Stereo SE
– Supply Current – mA
I
DD
Figure 50
1.5
1
0.5
0
2.5 3 3.5 4 4.5 5
2
2.5
3
5.5 6
RL = 4Ω
RL = 8 Ω
– Output Power – WP
O
OUTPUT POWER
vs
SUPPLY VOLTAGE
VDD – Supply Voltage – V
THD+N = 1%
BTL
Each Channel
RL = 3 Ω
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
21
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 51
0.4
0.2
0
2.5 3 3.5 4 4.5 5
0.6
0.8
1
5.5 6
RL = 4 Ω
RL = 8 Ω
RL = 32 Ω
THD+N = 1%
SE
Each Channel
– Output Power – WP
O
OUTPUT POWER
vs
SUPPLY VOLTAGE
VDD – Supply Voltage – V
Figure 52
RL – Load Resistance – Ω
1.5
1
0.5
0
04 8121620
2
2.5
3
24 28 32
THD+N = 1%
BTL
Each Channel
– Output Power – WP
O
OUTPUT POWER
vs
LOAD RESISTANCE
VDD = 5 V
VDD = 3.3 V
Figure 53
0.4
0.2
0
04 8121620
0.6
0.8
1
24 28 32
RL – Load Resistance – Ω
THD+N = 1%
SE
Each Channel
– Output Power – WP
O
OUTPUT POWER
vs
LOAD RESISTANCE
VDD = 5 V
VDD = 3.3 V
Figure 54
0.6
0.4
0.2
0
01
– Power Dissipation – W
1
1.2
POWER DISSIPATION
vs
OUTPUT POWER
1.4
1.5 2.5
0.8
PO – Output Power – W
P
D
RL = 4 Ω
RL = 8 Ω
VDD = 5 V
BTL
Each Channel
RL = 3 Ω
1.6
1.8
0.5 2
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
22
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 55
0.3
0.2
0.1
0
0 0.25 0.5
– Power Dissipation – W
0.5
0.6
POWER DISSIPATION
vs
OUTPUT POWER
0.7
0.75 1
0.4
PO – Output Power – W
P
D
RL = 4 Ω
RL = 8 Ω
VDD = 3.3 V
BTL
Each Channel
RL = 3 Ω
0.8
Figure 56
0.4
0.2
0
0 0.1 0.2 0.3
0.6
0.8
0.4 0.5 0.6
– Power Dissipation – W
POWER DISSIPATION
vs
OUTPUT POWER
PO – Output Power – W
P
D
RL = 4 Ω
RL = 8 Ω
VDD = 5 V
SE
Each Channel
RL = 32Ω
Figure 57
0.2
0
0 0.05 0.1 0.15
0.4
0.6
0.2 0.25
– Power Dissipation – W
POWER DISSIPATION
vs
OUTPUT POWER
PO – Output Power – W
P
D
RL = 4 Ω
RL = 8 Ω
VDD = 3.3V
SE
Each Channel
RL = 32Ω
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
23
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
THERMAL INFORMATION
The thermally enhanced PWP package is based on the 24-pin TSSOP, but includes a thermal pad (see Figure 58)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface-mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements (<2 mm) of many of
today’s advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerP AD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.
DIE
Side View (a)
End View (b)
Bottom View (c)
DIE
Thermal
Pad
Figure 58. Views of Thermally Enhanced PWP Package
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
24
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
bridged-tied load versus single-ended mode
Figure 59 shows a linear audio power amplifier (APA) in a BTL configuration. The TPA0202 BTL amplifier
consists of two linear amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration but initially consider power to the load. The differential drive to the speaker means
that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage
swing on the load as compared to a ground referenced load. Plugging 2 × V
O(PP)
into the power equation, where
voltage is squared, yields 4× the output power from the same supply rail and load impedance (see equation 1).
Power
+
V
(rms)
2
R
L
(1)
V
(rms)
+
V
O(PP)
22
Ǹ
R
L
2x V
O(PP)
V
O(PP)
–V
O(PP)
V
DD
V
DD
Figure 59. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-Ω speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvement —
which is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 60. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 µF to 1000 µF)
so they tend to be expensive, heavy , occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 2.
fc+
1
2pRLC
C
(2)
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
25
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
bridged-tied load versus single-ended mode (continued)
For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
R
L
C
C
V
O(PP)
V
O(PP)
V
DD
–3 dB
f
c
Figure 60. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4× the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the
thermal considerations
section.
BTL amplifier efficiency
Linear amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across the
output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc
voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the
output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from V
DD
.
The internal voltage drop multiplied by the RMS value of the supply current, IDDrms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS values of power in the load and in
the amplifier, the current and voltage waveform shapes must first be understood (see Figure 61).
V
(LRMS)
V
O
I
DD
I
DD(RMS)
Figure 61. Voltage and Current Waveforms for BTL Amplifiers
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
26
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
IDDrms
+
2V
P
p
R
L
P
SUP
+
VDDIDDrms
+
VDD2V
P
p
R
L
Efficiency
+
P
L
P
SUP
Efficiency of a BTL Configuration
+
p
V
P
2V
DD
+
p
ǒ
PLR
L
2
Ǔ
1ń2
2V
DD
(3)
Where:
(4)
PL+
VLrms
2
R
L
+
V
p
2
2R
L
VLrms
+
V
P
2
Ǹ
T able 1 employs equation 4 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-Ω loads and a 5-V supply , the maximum
draw on the power supply is almost 3.25 W.
Table 1. Efficiency Vs Output Power in 5-V 8-Ω BTL Systems
OUTPUT POWER
(W)
EFFICIENCY
(%)
PEAK-TO-PEAK
VOLTAGE
(V)
INTERNAL
DISSIPATION
(W)
0.25 31.4 2.00 0.55
0.50 44.4 2.83 0.62
1.00 62.8 4.00 0.59
1.25 70.2 4.47
†
0.53
†
High peak voltages cause the THD to increase.
A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 4, VDD is in the denominator. This
indicates that as VDD goes down, efficiency goes up.
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
27
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
For example, if the 5-V supply is replaced with a 3.3-V supply (TP A0202 has a maximum recommended V
DD
of 5.5 V) in the calculations of T able 1, then ef ficiency at 0.5 W would rise from 44% to 67% and internal power
dissipation would fall from 0.62 W to 0.25 W at 5 V. Then for a stereo 0.5-W system from a 3.3-V supply, the
maximum draw would only be 1.5 W as compared to 2.24 W from 5 V . In other words, use the efficiency analysis
to choose the correct supply voltage and speaker impedance for the application.
selection of components
Figure 62 and Figure 63 are a schematic diagrams of a typical notebook computer application circuits.
C
B
1 µF
C
S
0.1 µF
(see Note A)
Right
MUX
RLINEIN
RHPIN
Left
MUX
LHPIN
LLINEIN
Bias, Mute,
Shutdown,
and SE/BTL
MUX Control
+
–
+
–
RBYPASS
MUTE IN
MUTE OUT
SHUTDOWN
LBYPASS
ROUT+
ROUT–
RV
DD
LV
DD
LOUT+
LOUT–
SE/BTL
HP/LINE
C
IR
1 µF
R
IR
10 kΩ
R
FR
50 kΩ
C
FR
5 pF
System
Control
C
IL
1 µF
R
IL
10 kΩ
NC
NC
4
5
6
8
9
11
19
20
21
R
FL
50 kΩ
C
FL
5 pF
100 kΩ
100 kΩ
V
DD
V
DD
C
OUTR
330 µF
C
OUTL
330 µF
10
3
16
7
14
18
15
22
1 kΩ
1 kΩ
0.1 µF
NOTE A: A 0.1 µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger aluminum
electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 62. TPA0202 Minimum Configuration Application Circuit
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A – FEBRUARY 1998 – REVISED MARCH 2000
28
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
C
BR
0.1 µF
C
SR
0.1 µF
(see Note B)
Right
MUX
RLINEIN
RHPIN
Left
MUX
LHPIN
LLINEIN
Bias, Mute,
Shutdown,
and SE/BTL
MUX Control
+
–
+
–
RBYPASS
MUTE IN
MUTE OUT
SHUTDOWN
LBYPASS
ROUT+
ROUT–
RV
DD
LV
DD
LOUT+
LOUT–
SE/BTL
HP/LINE
C
IRLINE
1 µF
R
IRLINE
10 kΩ
R
FRLINE
50 kΩ
C
FRLINE
5 pF
System
Control
C
ILLINE
1 µF
R
ILLINE
10 kΩ
4
5
6
8
9
11
19
20
21
R
FLLINE
50 kΩ
C
FLLINE
5 pF
100 kΩ
100 kΩ
V
DD
C
OUTR
330 µF
C
OUTL
330 µF
10
3
16
7
14
18
15
22
1 kΩ
C
IRHP
1 µF
R
IRHP
10 kΩ
R
FRHP
10 kΩ
See Note A
C
BL
1 µF
C
ILHP
1 µF
R
ILHP
10 kΩ
R
FLHP
10 kΩ
C
SR
0.1 µF
(see Note B)
V
DD
1 kΩ
0.1 µF
NOTES: A. This connection is for ultra-low current in shutdown mode.
B. A 0.1 µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
aluminum electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 63. TPA0202 Full Configuration Application Circuit
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
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APPLICATION INFORMATION
gain setting resistors, RF and R
I
The gain for each audio input of the TP A0202 is set by resistors RF and RI according to equation 5 for BTL mode.
(5)
BTL Gain
+*
2
ǒ
R
F
R
I
Ǔ
BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the
voltage swing across the load. Given that the TPA0202 is a MOS amplifier, the input impedance is very high,
consequently input leakage currents are not generally a concern although noise in the circuit increases as the
value of RF increases. In addition, a certain range of RF values is required for proper start-up operation of the
amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the
amplifier be set between 5 kΩ and 20 kΩ. The effective impedance is calculated in equation 6.
(6)
Effective Impedance
+
R
FRI
RF)
R
I
As an example consider an input resistance of 10 kΩ and a feedback resistor of 50 kΩ. The BTL gain of the
amplifier would be –10 and the effective impedance at the inverting terminal would be 8.3 kΩ, which is well within
the recommended range.
For high performance applications metal film resistors are recommended because they tend to have lower noise
levels than carbon resistors. For values of RF above 50 kΩ the amplifier tends to become unstable due to a pole
formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capacitor of approximately 5 pF should be placed in parallel with R
F
when RF is greater than
50 kΩ. This, in effect, creates a low pass filter network with the cutoff frequency defined in equation 7.
(7)
f
c(lowpass)
+
1
2pR
FCF
–3 dB
f
c
For example, if RF is 100 kΩ and Cf is 5 pF then fc is 318 kHz, which is well outside of the audio range.
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
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APPLICATION INFORMATION
input capacitor, C
I
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency
determined in equation 8.
(8)
–3 dB
f
c
f
c(highpass)
+
1
2pR
ICI
The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where RI is 10 kΩ and the specification calls for a flat bass response down to 40 Hz.
Equation 8 is reconfigured as equation 9.
(9)
C
I
+
1
2pRIf
c
In this example, CI is 0.40 µF so one would likely choose a value in the range of 0.47 µF to 1 µF. A further
consideration for this capacitor is the leakage path from the input source through the input network (RI, CI) and
the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier
that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at V
DD
/2, which is likely higher
that the source dc level. Please note that it is important to confirm the capacitor polarity in the application.
power supply decoupling, C
S
The TPA0202 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 µF placed as close as possible to the device V
DD
lead works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near
the audio power amplifier is recommended.
TPA0202
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APPLICATION INFORMATION
midrail bypass capacitor, C
B
The midrail bypass capacitor, CB, is the most critical capacitor and serves several important functions. During
startup or recovery from shutdown mode, CB determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N. The capacitor is fed from a 100-kΩ source inside the amplifier. To keep the start-up pop as low as
possible, the relationship shown in equation 10 should be maintained.
(10)
1
ǒ
CB
100 kΩ
Ǔ
v
1
C
I
ǒ
RI)
R
F
Ǔ
As an example, consider a circuit where CB is 1 µF, CI is 0.22 µF, RF is 50 kΩ, and RI is 10 kΩ. Inserting these
values into the equation 10 we get 10 ≤ 75, which satisfies the rule. Bypass capacitor, CB, values of 0.1 µF to
1 µF ceramic or tantalum low-ESR capacitors are recommended for the best THD and noise performance.
In Figure 63, the full feature configuration, two bypass capacitors are used. This provides the maximum
separation between right and left drive circuits. When absolute minimum cost and/or component space is
required, one bypass capacitor can be used as shown in Figure 62. It is critical that terminals 6 and 19 be tied
together in this configuration.
load considerations
Extremely low impedance loads (below 4 Ω) coupled with certain external component selections, board layouts,
and cabling can cause oscillations in the system. Using a single air-cored inductor in series with the load
eliminates any spurious oscillations that might occur. An inductance of approximately 1 µH has been shown to
eliminate such oscillations. There are no special considerations when using 4 Ω and above loads with this
amplifier.
optimizing depop operation
Circuitry has been included in the TPA0202 to minimize the amount of popping heard at power-up and when
coming out of shutdown mode. Popping occurs whenever a voltage step is applied to the speaker. If high
impedances are used for the feedback and input resistors, it is possible for the input capacitor to drift downwards
from mid-rail during mute and shutdown. A high gain amplifier intensifies the problem as the small delta in
voltage is multiplied by the gain. So it is advantageous to use low-gain configurations, and to limit the size of
the gain-setting resistors. The time constant of the input coupling capacitor (CI) and the gain-setting resistors
(RI and RF) needs to be shorter than the time constant formed by the bypass capacitor (CB) and the output
impedance of the mid-rail generator, which is nominally 100 kΩ (see equation 10).
The effective output impedance of the mid-rail generator is actually greater than 100 kΩ due to a PNP transistor
clamping the input node (see Figure 64).
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
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APPLICATION INFORMATION
50 kΩ
BYPASS
V
DD
100 kΩ
100 kΩ
Figure 64. PNP Transistor Clamping of BYPASS Terminal
The PNP transistor limits the voltage drop across the 50 kΩ resistor by slewing the internal node slowly when
power is applied. At start-up, the xBYP ASS capacitor is at 0. The PNP is pulling the mid-point of the bias circuit
down, so the capacitor sees a lower effective voltage, and thus charges slower . This appears as a linear ramp
(while the PNP transistor is conducting), followed by the expected exponential ramp of an R-C circuit.
If the expression in equation 10 cannot be fulfilled or the small amount of pop is still unacceptable for the
application, then external circuitry must be added that can eliminate the pop heard during power up and while
transitioning out of mute or shutdown modes.
By holding the device in SE mode when the pop normally occurs, no pop can be heard through the
BTL-connected speakers (as the negative output is in a high impedance state when the amplifier is in SE mode).
From a hardware point of view, the easiest way to implement this is to drive the SE/BTL terminal from the
general-purpose input-output (GPIO) in the system. If the SE/BTL terminal is normally connected to a
headphone socket (as shown in Figure 65), then the GPIO signal must either be taken through an OR gate (see
Figure 65) or isolated with a diode (any signal diode) (see Figure 66).
R
m2
100 kΩ
R
m1
100 kΩ
V
DD
Left
Channel
From GPIO
Right
Channel
SE/BTL
0.1 µF
Figure 65. Implementation with an OR Gate
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
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APPLICATION INFORMATION
SE/BTL
R
m2
100 kΩ
R
m1
100 kΩ
V
DD
Left
Channel
From GPIO
Right
Channel
0.1 µF
Figure 66. Implementation with a Diode
The OR gate and diode isolate the GPIO terminal from the headphone switch. In these implementations, the
headphone switch has priority.
When the amplifier is in mute mode, the output stage continues to be biased. This causes the transition out of
mute mode to be very fast with only a short delay (from 100 ms to 500 ms). During power up or the transition
out of shutdown mode, a longer delay ( from 1 s to 2 s) is required. The exact delay time required is dependent
on the values of the external components used with the amplifier (see Figure 67).
System Control:
MUTE or SHUTDOWN
Delay
Output of Delay Circuit
(Input to SE/BTL
)
Figure 67. Transition Delay Timing
single-ended operation
In SE mode (see Figure 59 and Figure 60), the load is driven from the primary amplifier output for each channel
(OUT+, terminals 22 and 3).
In SE mode the gain is set by the RF and RI resistors and is shown in equation 1 1. Since the inverting amplifier
is not used to mirror the voltage swing on the load, the factor of 2, from equation 5, is not included.
(11)
SE Gain+*
ǒ
R
F
R
I
Ǔ
The output coupling capacitor required in single-supply SE mode also places additional constraints on the
selection of other components in the amplifier circuit. The rules described earlier still hold with the addition of
the following relationship (see equation 12):
(12)
1
ǒ
CB
25 kΩ
Ǔ
v
1
ǒ
CIR
I
Ǔ
Ơ
1
RLC
C
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
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APPLICATION INFORMATION
output coupling capacitor, C
C
In the typical single-supply SE configuration, an output coupling capacitor (C
C
) is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 14.
(14)
f
c(high)
+
1
2pRLC
C
–3 dB
f
c
The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher degrading the bass response. Large values of CC are required to pass low
frequencies into the load. Consider the example where a CC of 330 µF is chosen and loads vary from 3 Ω,
4 Ω, 8 Ω, 32 Ω, 10 kΩ, to 47 kΩ. Table 2 summarizes the frequency response characteristics of each
configuration.
Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode
R
L
C
C
LOWEST FREQUENCY
3 Ω 330 µF 161 Hz
4 Ω 330 µF 120 Hz
8 Ω 330 µF 60 Hz
32 Ω 330 µF
15 Hz
10,000 Ω 330 µF 0.05 Hz
47,000 Ω 330 µF 0.01 Hz
As Table 2 indicates, most of the bass response is attenuated into a 4-Ω load, an 8-Ω load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.
TPA0202
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APPLICATION INFORMATION
SE/BTL
operation
The ability of the TP A0202 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0202, two separate amplifiers drive OUT+ and OUT–. The SE/BTL input (terminal 14)
controls the operation of the follower amplifier that drives LOUT– and ROUT– (terminals 10 and 15). When
SE/BTL
is held low, the amplifier is on and the TP A0202 is in the BTL mode. When SE/BTL is held high, the OUT–
amplifiers are in a high output impedance state, which configures the TPA0202 as an SE driver from LOUT+
and ROUT+ (terminals 3 and 22). IDD is reduced by approximately one-half in SE mode. Control of the SE/BTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 68.
MUX
RLINE IN
RHP IN
+
–
Bypass
SE/BTL
HP/LINE
R
m2
100 kΩ
R
m1
100 kΩ
V
DD
C
OUTR
16
14
R
m3
1 kΩ
+
–
ROUT–
15
ROUT+ 22
Left
Channel
20
21
0.1 µF
Figure 68. TPA0202 Resistor Divider Network Circuit
Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 100-kΩ/1-kΩ divider pulls the SE/BTL
input low. When a plug is inserted, the 1-kΩ
resistor is disconnected and the SE/BTL input is pulled high. When the input goes high, the OUT– amplifier is
shutdown causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (CO) into the headphone jack.
As shown in the full feature application (Figure 63), the input MUX control can be tied to the SE/BTL input. The
benefits of doing this are described in the following input MUX operation section.
TPA0202
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APPLICATION INFORMATION
Input MUX operation
Working in concert with the SE/BTL feature, the HP/LINE MUX feature gives the audio designer the flexibility
of a multichip design in a single IC (see Figure 69). The primary function of the MUX is to allow different gain
settings for BTL versus SE mode. Speakers typically require approximately a factor of 10 more gain for similar
volume listening levels as compared to headphones. To achieve headphone and speaker listening parity, the
resistor values would need to be set as follows:
(15)
SE Gain
(HP)
+
*
ǒ
R
F(HP)
R
I(HP)
Ǔ
If, for example R
I(HP)
= 10 kΩ and R
F(HP)
= 10 kΩ then SE Gain
(HP)
= –1
(16)
BTL Gain
(LINE)
+*
2
ǒ
R
F(LINE)
R
I(LINE)
Ǔ
If, for example R
I(LINE)
= 10 kΩ and R
F(LINE)
= 50 kΩ then BTL Gain
(LINE)
= –10
ROUT+
ROUT–
C
IRLINE
R
IRLINE
15
22
C
IRHP
R
IRHP
R
FRHP
MUX
RLINE IN
RHP IN
SE/BTL
HP/LINE
V
DD
16
14
+
–
Left Channel
20
21
Right Channel
MID
R
FRLINE
0.1 µF
Figure 69. TPA0202 Example Input MUX Circuit
Another advantage of using the MUX feature is setting the gain of the headphone channel to –1. This provides
the optimum distortion performance into the headphones where clear sound is more important. Refer to the
SE/BTL operation section for a description of the headphone jack control circuit.
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
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APPLICATION INFORMATION
mute and shutdown modes
The TP A0202 employs both a mute and a shutdown mode of operation designed to reduce supply current, IDD,
to the absolute minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input
terminal should be held low during normal operation when the amplifier is in use. Pulling SHUTDOWN high
causes the outputs to mute and the amplifier to enter a low-current state, IDD = 5 µA. SHUTDOWN or MUTE
IN should never be left unconnected because amplifier operation would be unpredictable. Mute mode alone
reduces I
DD
to 1.5 mA.
Table 3. Shutdown and Mute Mode Functions
INPUTS
†
OUTPUT
AMPLIFIER STATE
SE/BTL
HP/LINE
MUTE IN SHUTDOWN MUTE OUT
INPUT OUTPUT
Low Low Low Low Low L/R Line BTL
X X — High — X Mute
X X High — High X Mute
Low High Low Low Low L/R HP BTL
High Low Low Low Low L/R Line SE
High High Low Low Low L/R HP SE
†
Inputs should never be left unconnected.
X = do not care
using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.
5-V versus 3.3-V operation
The TP A0202 operates over a supply range of 3 V to 5.5 V. This data sheet provides full specifications for 5-V
and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no
special considerations for 3.3-V versus 5-V operation as far as supply bypassing, gain setting, or stability goes.
For 3.3-V operation, supply current is reduced from 19 mA (typical) to 13 mA (typical). The most important
consideration is that of output power. Each amplifier in TPA0202 can produce a maximum voltage swing of
V
DD
– 1 V. This means, for 3.3-V operation, clipping starts to occur when V
O(PP)
= 2.3 V as opposed to
V
O(PP)
= 4 V at 5 V . The reduced voltage swing subsequently reduces maximum output power into an 8-Ω load
before distortion becomes significant.
Operation from 3.3-V supplies, as can be shown from the efficiency formula in equation 4, consumes
approximately two-thirds the supply power for a given output-power level than operation from 5-V supplies.
When the application demands less than 500 mW, 3.3-V operation should be strongly considered, especially
in battery-powered applications to improve the efficiency.
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
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APPLICATION INFORMATION
headroom and thermal considerations
Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions.
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA0202 data sheet, one can see that when the
TP A0202 is operating from a 5-V supply into a 3-Ω speaker that 2 W peaks are available. Converting watts to
dB:
PdB+
10Log
ǒ
P
W
P
ref
Ǔ
+
10Log
ǒ
2
1
Ǔ
+
3.0 dB
(17)
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
3.0 dB*15 dB
+*
12 dB(15 dB headroom
)
3.0 dB*12 dB
+*
9dB(12 dB headroom
)
3.0 dB*9dB
+*
6dB(9 dB headroom
)
3.0 dB*6dB
+*
3dB(6 dB headroom
)
3.0 dB*3dB+0dB(3 dB headroom
)
Converting dB back into watts:
PW+
10
PdBń10
P
ref
+
63 mW (15 dB headroom)
+
120 mW (12 dB headroom)
+
250 mW (9 dB headroom)
+
500 mW (6 dB headroom)
+
1000 mW (3 dB headroom)
(18)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with 0 dB of
headroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for
the system. Using the power dissipation curves for a 5-V, 3-Ω system, the internal dissipation in the TPA0202
and maximum ambient temperatures is shown in Table 4.
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
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APPLICATION INFORMATION
headroom and thermal considerations (continued)
Table 4. TPA0202 Power Rating, 5-V, 3-Ω, Stereo
PEAK OUTPUT POWER
(W)
AVERAGE OUTPUT POWER
POWER DISSIPATION
(W/Channel)
MAXIMUM AMBIENT
TEMPERATURE
2 2 W 1.7 –3°C
2 1000 mW (3 dB) 1.6 6°C
2 500 mW (6 dB) 1.4 24°C
2 250 mW (9 dB) 1.1 51°C
2 120 mW (12 dB) 0.8 78°C
2 63 mW (15 dB) 0.6 96°C
DISSIPATION RATING TABLE
PACKAGE
TA ≤ 25°C
DERATING FACTOR TA = 70°C TA = 85°C
PWP
†
2.7 W
21.8 mW/°C
1.7 W
1.4 W
PWP
‡
2.8 W
22.1 mW/°C 1.8 W 1.4 W
†
This parameter is measured with the recommended copper heat sink pattern on a 1-layer PCB, 4 in2 5-in × 5-in PCB,
1 oz. copper, 2-in × 2-in coverage.
‡
This parameter is measured with the recommended copper heat sink pattern on an 8-layer PCB, 6.9 in2 1.5-in × 2-in PCB,
1 oz. copper with layers 1, 2, 4, 5, 7, and 8 at 5% coverage (0.9 in2) and layers 3 and 6 at 100% coverage (6 in2).
The maximum ambient temperature depends on the heatsinking ability of the PCB system. Using the 0 CFM
and 300 CFM data from the dissipation rating table, the derating factor for the PWP package with 6.9 in2 of
copper area on a multilayer PCB is 22 mW/°C and 54 mW/°C respectively. Converting this to ΘJA:
Θ
JA
+
1
Derating
+
1
0.022
+
45°CńW
For 0 CFM :
(19)
To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated heat needs to be doubled for two channel operation. Given ΘJA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TP A0202 is
150 °C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
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APPLICATION INFORMATION
headroom and thermal considerations (continued)
TAMax+TJMax
*
Θ
JAPD
+
150*45(0.6 2)+
96°C(15 dB headroom, 0 CFM
)
(20)
NOTE:
Internal dissipation of 0.6 W is estimated for a 2-W system with 15 dB headroom per channel.
Table 4 shows that for some applications no airflow is required to keep junction temperatures in the specified
range. The TP A0202 is designed with thermal protection that turns the device off when the junction temperature
surpasses 150°C to prevent damage to the IC. Table 4 was calculated for maximum listening volume without
distortion. When the output level is reduced the numbers in the table change significantly. Also, using 8-Ω
speakers dramatically increases the thermal performance by increasing amplifier efficiency.
junction temperature measurement
Characterizing a PCB layout with respect to thermal impedance is very difficult, as it is usually impossible to
know the junction temperature of the IC in question. The TP A0202 terminal 2 (TJ) sources a current proportional
to the junction temperature. The circuit internal to TJ is shown in Figure 70.
5R
TJ
V
DD
R
R
Figure 70. TJ Terminal Internal Circuit
Connect an ammeter between TJ and ground to measure the current. As the resistors have a tolerance of ±20%,
this measurement must be calibrated on each device. The intent of this function is in characterization of the PCB
and end equipment and not a real-time measurement of temperature. Typically a 25°C reading is –120 µA for
a 3.3-V supply and –135 µA for a 5-V supply . The slope is approximately 0.25 µA/°C for both V
DD
= 3.3 V and
VDD = 5 V. To reduce quiescent current, do not ground TJ in normal operation. It can be connected to VDD or
left floating as it has a resistor connected across the base-emitter junction.
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
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POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
MECHANICAL INFORMATION
PWP (R-PDSO-G**) PowerPAD PLASTIC SMALL-OUTLINE
4073225/F 10/98
0,50
0,75
0,25
0,15 NOM
Thermal Pad
(See Note D)
Gage Plane
2824
7,70
7,90
20
6,40
6,60
9,60
9,80
6,60
6,20
11
0,19
4,50
4,30
10
0,15
20
A
1
0,30
1,20 MAX
1614
5,10
4,90
PINS **
4,90
5,10
DIM
A MIN
A MAX
0,05
Seating Plane
0,65
0,10
M
0,10
0°–8°
20 PINS SHOWN
NOTES: A. All linear dimensions are in millimeters.
B. This drawing is subject to change without notice.
C. Body dimensions do not include mold flash or protrusions.
D. The package thermal performance may be enhanced by bonding the thermal pad to an external thermal plane.
This pad is electrically and thermally connected to the backside of the die and possibly selected leads.
E. Falls within JEDEC MO-153
PowerPAD is a trademark of Texas Instruments Incorporated.
IMPORTANT NOTICE
T exas Instruments and its subsidiaries (TI) reserve the right to make changes to their products or to discontinue
any product or service without notice, and advise customers to obtain the latest version of relevant information
to verify, before placing orders, that information being relied on is current and complete. All products are sold
subject to the terms and conditions of sale supplied at the time of order acknowledgement, including those
pertaining to warranty, patent infringement, and limitation of liability.
TI warrants performance of its semiconductor products to the specifications applicable at the time of sale in
accordance with TI’s standard warranty. Testing and other quality control techniques are utilized to the extent
TI deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily
performed, except those mandated by government requirements.
CERT AIN APPLICATIONS USING SEMICONDUCTOR PRODUCTS MAY INVOLVE POTENTIAL RISKS OF
DEATH, PERSONAL INJURY, OR SEVERE PROPERTY OR ENVIRONMENTAL DAMAGE (“CRITICAL
APPLICATIONS”). TI SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED, AUTHORIZED, OR
WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT DEVICES OR SYSTEMS OR OTHER
CRITICAL APPLICATIONS. INCLUSION OF TI PRODUCTS IN SUCH APPLICA TIONS IS UNDERSTOOD T O
BE FULLY AT THE CUSTOMER’S RISK.
In order to minimize risks associated with the customer’s applications, adequate design and operating
safeguards must be provided by the customer to minimize inherent or procedural hazards.
TI assumes no liability for applications assistance or customer product design. TI does not warrant or represent
that any license, either express or implied, is granted under any patent right, copyright, mask work right, or other
intellectual property right of TI covering or relating to any combination, machine, or process in which such
semiconductor products or services might be or are used. TI’s publication of information regarding any third
party’s products or services does not constitute TI’s approval, warranty or endorsement thereof.
Copyright 2000, Texas Instruments Incorporated
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