Datasheet THS6012EVM, THS6012CDWP, THS6012CDWPR, THS6012IDWPR, THS6012IGQER Datasheet (Texas Instruments)
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THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
1
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
D
ADSL Differential Line Driver
D
400 mA Minimum Output Current Into 25-Ω
Load
D
High Speed
– 140 MHz Bandwidth (–3dB) With 25-Ω
Load
– 315 MHz Bandwidth (–3dB) With 100-Ω
Load
– 1300 V/µs Slew Rate, G = 5
D
Low Distortion
– –72 dB 3rd Order Harmonic Distortion at
f = 1 MHz, 25-Ω Load, and 20 V
PP
D
Independent Power Supplies for Low
Crosstalk
D
Wide Supply Range ±4.5 V to ±16 V
D
Thermal Shutdown and Short Circuit
Protection
D
Improved Replacement for AD815
D
Evaluation Module Available
description
The THS6012 contains two high-speed drivers
capable of providing 400 mA output current (min)
into a 25 Ω load. These drivers can be configured
differentially to drive a 50-Vp-p output signal over
low-impedance lines. The drivers are current
feedback amplifiers, designed for the high slew
rates necessary to support low total harmonic
distortion (THD) in xDSL applications. The THS6012 is ideally suited for asymmetrical digital subscriber line
(ADSL) applications at the central office, where it supports the high-peak voltage and current requirements of
this application.
Separate power supply connections for each driver are provided to minimize crosstalk. The THS6012 is
available in the small surface-mount, thermally enhanced 20-pin PowerPAD package.
HIGH-SPEED xDSL LINE DRIVER/RECEIVER FAMILY
DEVICE
DRIVER RECEIVER DESCRIPTION
THS6002
•
• Dual differential line drivers and receivers
THS6012 • 500-mA dual differential line driver
THS6022 • 250-mA dual differential line driver
THS6032 • Low-power ADSL central office line driver
THS6062 • Low-noise ADSL receiver
THS7002 • Low-noise programmable gain ADSL receiver
CAUTION: The THS6012 provides ESD protection circuitry. However, permanent damage can still occur if this device is subjected
to high-energy electrostatic discharges. Proper ESD precautions are recommended to avoid any performance degradation or loss
of functionality.
Copyright 2000, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments Incorporated.
VCC–
1OUT
V
CC+
1IN+
1IN–
NC
NC
NC
NC
NC
V
CC–
2OUT
V
CC+
2IN+
2IN–
NC
NC
NC
NC
NC
1
2
3
4
5
6
7
8
9
10
20
19
18
17
16
15
14
13
12
11
Cross Section View Showing PowerPAD
Thermally Enchanced SOIC (DWP)
PowerPAD Package
(TOP VIEW)
(SIDE VIEW)
MicroStar Junior (GQE) Package
(TOP VIEW)
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
2
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
AVAILABLE OPTIONS
PACKAGED DEVICE
T
A
PowerPAD PLASTIC
SMALL OUTLINE
†
(DWP)
MicroStar Junior
(GQE)
EVALUATION
MODULE
0°C to 70°C THS6012CDWP THS6012CGQE THS6012EVM
–40°C to 85°C THS6012IDWP THS6012IGQE —
†
The PWP packages are available taped and reeled. Add an R suffix to the device type (i.e.,
THS6012CPWPR)
functional block diagram
_
+
Driver 1
Driver 2
_
+
3
4
5
17
16
2
1
18
19
20
VCC+
V
CC+
V
CC–
V
CC–
1OUT
2OUT
1IN+
1IN–
2IN+
2IN–
Terminal Functions
TERMINAL
NAME DWP PACKAGE
TERMINAL NO.
GQE PACKAGE
TERMINAL NO.
1OUT 2 A3
1IN– 5 F1
1IN+ 4 D1
2OUT 19 A7
2IN– 16 F9
2IN+ 17 D9
V
CC+
3, 18 B1, B9
V
CC–
1, 20 A4, A6
NC 6, 7, 8 ,9, 10, 11, 12, 13,
14, 15
NA
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
3
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
pin assignments
98765
A
B
C
D
E
F
321
G
H
J
4
2IN+
1N+
1IN–
NC
NC
NC
NC
NC
NCNC NC NC NC
NC NC NC
NC NC NC
NC NC NC
NC NC
NC NC NC
V
CC+
NCNC
NCNC
NC
NC
NCNC NC
NCNC NC
NC NC
NC NC
NC
NC
NC
NC
NC NC
NC
2OUT
NC
NC
NC NCNC NC
NC
NC NC NC
NC
NC
1OUT
NC NC
2IN–
NC
NCNC
NC
V
CC+
NOTE: Shaded terminals are used for thermal connection to the ground plane.
MicroStarJunior (GQE) Package
(TOP VIEW)
V
CC–
V
CC–
NC
NCNCNC
NC
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
4
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
absolute maximum ratings over operating free-air temperature (unless otherwise noted)
†
Supply voltage, V
CC+
to V
CC–
33 V. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Input voltage, VI (driver and receiver) ±V
CC
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Output current, I
O
(driver) (see Note 1) 800 mA. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Differential input voltage, VID 6 V. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Continuous total power dissipation at (or below) TA = 25°C (see Note 1) 5.8 W. . . . . . . . . . . . . . . . . . . . . . . . . .
Operating free air temperature, T
A
–40°C to 85°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Storage temperature, T
stg
–65°C to 125°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds 300°C. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
†
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
NOTE 1: The THS6012 incorporates a PowerPad on the underside of the chip. This acts as a heatsink and must be connected to a thermal
dissipation plane for proper power dissipation. Failure to do so can result in exceeding the maximum junction temperature, which could
permanently damage the device. See the
Thermal Information
section of this document for more information about PowerPad
technology.
recommended operating conditions
MIN TYP MAX UNIT
pp
Split supply ±4.5 ±16
Suppl
y v
oltage, V
CC+
and V
CC–
Single supply 9 32
V
p
p
C suffix 0 70
°
O erating free-air tem erature, T
A
I suffix –40 85
°C
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
5
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
electrical characteristics, VCC = ±15 V, RL = 25 Ω, RF = 1 kΩ, TA = 25°C (unless otherwise noted)
dynamic performance
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
VI = 200 mV , G = 1,
RF = 680 Ω,RL = 25 Ω
VCC = ±15 V 140
VI = 200 mV , G = 1,
RF = 1 kΩ,RL = 25 Ω
VCC = ±5 V 100
VI = 200 mV , G = 2,
RF = 620 Ω,RL = 25 Ω
VCC = ±15 V 120
Small-signal bandwidth (–3 dB)
VI = 200 mV , G = 2,
RL = 25 Ω,RF = 820 Ω
VCC = ±5 V 100
MH
z
BW
VI = 200 mV , G = 1,
RF = 820 Ω,RL = 100 Ω
VCC = ±15 V 315
VI = 200 mV , G = 2,
RF = 560 Ω,RL = 100 Ω
VCC = ±15 V 265
VCC = ±5 V,
RF = 820 Ω
30
Bandwidth for 0.1 dB flatness
V
I
=
200 mV,G
=
1
VCC = ±15 V,
RF = 680 Ω
40
MH
z
p
VCC = ±15 V, V
O(PP)
= 20 V 20
Full power bandwidth (see Note 3)
VCC = ±5 V, V
O(PP)
= 4 V 35
MH
z
VCC = ±15 V, VO = 20 V
(PP)
, G = 5 1300
SR
Slew rate
VCC = ±5 V, VO = 5 V
(PP)
, G = 2 900
V/µs
t
s
Settling time to 0.1% 0 V to 10 V Step, G = 2 70 ns
noise/distortion performance
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
V
= ±15 V, R
= 680 Ω,
V
O(PP)
= 20 V –65
CC
,
F
,
G = 2, f = 1 MHz
V
O(PP)
= 2 V –79
THD
Total harmonic distortion
VCC = ±5 V, RF = 680 Ω,
G = 2, f = 1 MHz
V
O(PP)
= 2 V –76
dBc
V
n
Input voltage noise
VCC = ±5 V or ±15 V, f = 10 kHz,
G = 2, Single-ended
1.7
nV/√Hz
p
Positive (IN+)
V
= ±5 V or ±15 V, f = 10 kHz,
11.5
InInput noise current
Negative (IN–)
CC
,
G = 2
,
16
p
A/√H
z
G = 2, NTSC,
VCC = ±5 V 0.04%
ADDifferential gain error
,,
RL = 150 Ω, 40 IRE Modulation
VCC = ±15 V 0.05%
p
G = 2, NTSC,
VCC = ±5 V 0.07°
φDDifferential hase error
RL = 150 Ω, 40 IRE Modulation
VCC = ±15 V 0.08°
Crosstalk Driver to driver VI = 200 mV, f = 1 MHz –62 dB
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
6
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
electrical characteristics, VCC = ±15 V, RL = 25 Ω, RF = 1 kΩ, TA = 25°C (unless otherwise noted)
(continued)
dc performance
PARAMETER TEST CONDITIONS
†
MIN TYP MAX UNIT
p
p
VCC = ±5 V 1.5
Open loop transresistance
VCC = ±15 V 5
MΩ
p
TA = 25°C 2 5
VIOInput offset voltage
V
CC
= ±5 V or
±15 V
TA = full range 7
mV
Input offset voltage drift VCC = ±5 V or ±15 V, TA = full range 20 µV/°C
p
TA = 25°C 1.5 4
Differential input offset voltage
V
CC
=
±5 V or ±15 V
TA = full range 5
mV
TA = 25°C 3 9
Negative
TA = full range 12
µ
A
p
TA = 25°C 4 10
IIBInput bias current
Positive
V
CC
= ±5 V or
±15 V
TA = full range 12
µ
A
TA = 25°C 1.5 8
Differential
TA = full range 11
µ
A
Differential input offset voltage drift VCC = ±5 V or ±15 V, TA = full range 10 µV/°C
input characteristics
PARAMETER TEST CONDITIONS
†
MIN TYP MAX UNIT
p
VCC = ±5 V ±3.6 ±3.7
V
ICR
Common-mode input voltage range
VCC = ±15 V ±13.4 ±13.5
V
Common-mode rejection ratio
62 70
CMRR
Differential common-mode rejection ratio
V
CC
= ±5 V or
±15 V
,
T
A
=
full range
100
dB
R
I
Input resistance 300 kΩ
C
I
Differential input capacitance 1.4 pF
output characteristics
PARAMETER TEST CONDITIONS
†
MIN TYP MAX UNIT
VCC = ±5 V
3
to
–2.8
3.2
to
–3
p
Single ended
R
L
= 25
Ω
VCC = ±15 V
11.8
to
–11.5
12.5
to
–12.2
V
VOOutput voltage swing
VCC = ±5 V
6
to
–5.6
6.4
to
–6
Differential
R
L
= 50
Ω
VCC = ±15 V
23.6
to
–23
25
to
–24.4
V
p
VCC = ±5 V, RL = 5 Ω 500
IOOutput current (see Note 2)
VCC = ±15 V, RL = 25 Ω 400 500
mA
I
OS
Short-circuit output current (see Note 2) 800 mA
R
O
Output resistance Open loop 13 Ω
NOTE 2: A heat sink is required to keep the junction temperature below absolute maximum when an output is heavily loaded or shorted. See
absolute maximum ratings and
Thermal Information
section.
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
7
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
electrical characteristics, VCC = ±15 V, RL = 25 Ω, RF = 1 kΩ, TA = 25°C (unless otherwise noted)
power supply
PARAMETER TEST CONDITIONS
†
MIN TYP MAX UNIT
pp
p
Split supply ±4.5 ±16.5
VCCPower supply operating range
Single supply 9 33
V
VCC = ±5 V TA = full range 12
I
CC
Quiescent current (each driver)
TA = 25°C 11.5 13
mA
V
CC
=
±15 V
TA = full range 15
TA = 25°C –68 –74
pp
V
CC
=
±5 V
TA = full range –65
dB
PSRR
Power supply rejection ratio
TA = 25°C –64 –72
V
CC
=
±15 V
TA = full range –62
dB
†
Full range is 0°C to 70°C for the THS6012C and –40°C to 85°C for the THS6012I.
PARAMETER MEASUREMENT INFORMATION
+
–
1 kΩ
V
I
V
O
25 Ω
50 Ω
1 kΩ
Driver 1
+
–
1 kΩ
V
I
V
O
25 Ω
50 Ω
1 kΩ
Driver 2
Figure 1. Input-to-Output Crosstalk Test Circuit
V
I
V
O
+
–
R
G
R
F
R
L
25 Ω
50 Ω
–15 V
15 V
Figure 2. Test Circuit, Gain = 1 + (RF/RG)
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
8
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
p
p
vs Supply voltage 3
V
O(PP)
Peak-to-peak output voltage
vs Load resistance 4
V
IO
Input offset voltage vs Free-air temperature 5
I
IB
Input bias current vs Free-air temperature 6
CMRR Common-mode rejection ratio vs Free-air temperature 7
Input-to-output crosstalk vs Frequency 8
PSRR Power supply rejection ratio vs Free-air temperature 9
Closed-loop output impedance vs Frequency 10
pp
vs Supply voltage 11
ICCSupply current
vs Free-air temperature 12
SR Slew rate vs Output step 13, 14
V
n
Input voltage noise vs Frequency
I
n
Input current noise vs Frequency
15
Normalized frequency response vs Frequency 16, 17
Output amplitude vs Frequency 18–21
Normalized output response vs Frequency 22–25
Small and large frequency response 26, 27
vs Frequency 28, 29
Single-ended harmonic distortion
vs Output voltage 30, 31
DC input offset voltage 32, 33
Differential gain
Number of 150-Ω loads 34, 35
p
DC input offset voltage 32, 33
Differential phase
Number of 150-Ω loads 34, 35
Output step response 36–38
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
9
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 3
VCC – Supply Voltage – V
PEAK-TO-PEAK OUTPUT VOLTAGE
vs
SUPPLY VOLTAGE
5678 1112109
0
–5
–10
–15
10
5
TA = 25°C
RF = 1 kΩ
RL = 25 Ω
Gain = 1
14 1513
15
V
O(PP)
– Peak-to-Peak Output Voltage – V
Figure 4
10 100 1000
RL – Load Resistance – Ω
15
5
–5
–15
10
0
–10
VCC = ±15 V
VCC = ±5 V
PEAK-TO-PEAK OUTPUT VOLTAGE
vs
LOAD RESISTANCE
TA = 25°C
RF = 1 kΩ
Gain = 1
VCC = ±5 V
VCC = ±15 V
V
O(PP)
– Peak-to-Peak Output Voltage – V
Figure 5
TA – Free-Air Temperature – °C
–40 –20 0 20 80 1006040
VCC = ±5 V
INPUT OFFSET VOLTAGE
vs
FREE-AIR TEMPERATURE
VCC = ±15 V
V
IO
– Input Offset Voltage – mV
1
–1
–3
–5
0
–2
–4
2
G = 1
RF = 1 kΩ
Figure 6
TA – Free-Air Temperature – °C
–40 –20 0 20 80 1006040
INPUT BIAS CURRENT
vs
FREE-AIR TEMPERATURE
5
3
1
0
4
2
VCC = ±15 V
I
IB+
VCC = ±5 V
I
IB+
VCC = ±15 V
I
IB–
VCC = ±5 V
I
IB–
I
IB
– Input Bias Current – Aµ
G = 1
RF = 1 kΩ
See Figure 2
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
10
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 7
TA – Free-Air Temperature – °C
CMRR – Common-Mode Rejection Ratio – dB
–40 –20 0 20 806040
75
65
60
80
70
VCC = ±5 V
COMMON-MODE REJECTION RATIO
vs
FREE-AIR TEMPERATURE
VCC = ±15 V
1 kΩ
1 kΩ
V
I
+
–
V
O
1 kΩ
1 kΩ
Figure 8
f – Frequency – Hz
INPUT–TO–OUTPUT CROSSTALK
vs
FREQUENCY
–80
–90
10M100k
–70
–60
500M
Input–To–Output Crosstalk – dB
–50
–20
1M 100M
VCC = ± 15 V
RF = 1 Ω
RL = 25 Ω
Gain = 2
VI = 200 mV
See Figure 2
Driver 1 = Input
Driver 2 = Output
Driver 1 = Output
Driver 2 = Input
–40
–30
Figure 9
TA – Free-Air Temperature – °C
PSRR – Power Supply Rejection Ratio – dB
–40 –20 0 20 80 1006040
POWER SUPPLY REJECTION RATIO
vs
FREE-AIR TEMPERATURE
90
80
70
65
85
75
95
VCC = 15 V
VCC = 5 V
VCC =–5 V
VCC = –15 V
G = 1
RF = 1 kΩ
Figure 10
1
0.1
0.01
0.001
1M
f – Frequency – Hz
100k 10M 100M
10
100
Closed-Loop Output Impedance –
VCC = ±15 V
RF = 1 kΩ
Gain = 2
TA = 25°C
V
I(PP)
= 1 V
500M
Ω
V
O
+
–
50 Ω
1 kΩ
1 kΩ
V
I
THS6012
1 kΩ
(
V
I
V
O
=
1000
Z
o
)
– 1
CLOSED-LOOP OUTPUT IMPEDANCE
vs
FREQUENCY
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
11
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 11
±VCC – Supply Voltage – V
5678 1112109
6
5
10
8
9
7
I
CC
– Supply Current – mA
TA = 25°C
RF = 1 kΩ
Gain = +1
14 1513
12
11
SUPPLY CURRENT
vs
SUPPLY VOLTAGE
Figure 12
TA – Free-Air Temperature – °C
–40 –20 0 20 80 1006040
SUPPLY CURRENT
vs
FREE-AIR TEMPERATURE
10
6
2
0
8
4
12
VCC = ±15 V
VCC = ±5 V
I
CC
– Supply Current – mA
13
Figure 13
0
SLEW RATE
vs
OUTPUT STEP
Output Step (Peak–To–Peak) – V
1500
100
20
900
5
1100
700
10
1300
Slew Rate – V Sµ
500
300
VCC = ± 15V
Gain = 5
RF = 1 kΩ
RL = 25 Ω
15
+SR
–SR
Figure 14
0
SLEW RATE
vs
OUTPUT STEP
Output Step (Peak–To–Peak) – V
1000
100
5
700
1
800
600
23
900
Slew Rate – V Sµ
500
300
4
VCC = ± 5V
Gain = 2
RF = 1 kΩ
RL = 25 Ω
400
200
+SR
–SR
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
12
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
10
1
100
VCC = ±15 V
TA = 25°C
In+ Noise
f – Frequency – Hz
10 100 1k 10k 100k
nV/ Hz
Hz
– Voltage Noise –V
n
– Current Noise – pA/I
n
10
1
100
In– Noise
Vn Noise
INPUT VOLTAGE AND CURRENT NOISE
vs
FREQUENCY
Figure 15
Figure 16
f – Frequency – Hz
Normalized Frequency Response – dB
100 1M 10M 100M
–2
–4
–6
–8
–3
–5
–7
2
0
1
–1
NORMALIZED FREQUENCY RESPONSE
vs
FREQUENCY
VCC = ±15 V
VI = 200 mV
RL = 25 Ω
Gain = 1
TA = 25°C
RF = 510 Ω
RF = 300 Ω
RF = 750 Ω
RF = 1 kΩ
500M
Figure 17
NORMALIZED FREQUENCY RESPONSE
vs
FREQUENCY
f – Frequency – Hz
Normalized Frequency Response – dB
100K 1M 10M 100M
–1
–3
–5
–7
–2
–4
–6
1
2
0
VCC = ±15 V
Vin = 200 mV
RL = 25 Ω
Gain = 2
TA = 25°C
–8
–10
–9
500M
RF = 470 Ω
RF = 360 Ω
RF = 620 Ω
RF = 1 kΩ
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
13
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 18
f – Frequency – Hz
OUTPUT AMPLITUDE
vs
FREQUENCY
–5
–6
10M100k
–2
–1
500M
Output Amplitude – dB
0
3
1M 100M
1
2
–4
–3
VCC = ± 5 V
Gain = 1
RL = 25 Ω
VI = 200 mV
RF = 1 kΩ
RF = 1.5 kΩ
RF = 620 Ω
Figure 19
f – Frequency – Hz
OUTPUT AMPLITUDE
vs
FREQUENCY
1
0
10M100k
4
5
500M
Output Amplitude – dB
6
9
1M 100M
7
8
2
3
VCC = ± 5 V
Gain = 2
RL = 25 Ω
VI = 200 mV
RF = 820 Ω
RF = 1.2 kΩ
RF = 510 Ω
Figure 20
f – Frequency – Hz
OUTPUT AMPLITUDE
vs
FREQUENCY
–10
10M100k
20
30
500M
Output Level – dB
40
70
1M 100M
50
60
Gain = 1000
0
10
VCC = ± 5 V
RG =10 Ω
RL = 25 Ω
VO = 2 V
Gain = 100
Figure 21
f – Frequency – Hz
OUTPUT AMPLITUDE
vs
FREQUENCY
–10
10M100k
20
30
500M
Output Level – dB
40
70
1M 100M
50
60
Gain = 1000
0
10
VCC = ± 5 V
RG =10 Ω
RL = 25 Ω
VO = 2 V
Gain = 100
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
14
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 22
NORMALIZED OUTPUT RESPONSE
vs
FREQUENCY
1M
f – Frequency – Hz
100k 10M 100M
Normalized Output Response – dB
500M
–3
–5
–7
–9
–4
–6
–8
–1
0
–2
1
RL = 200 Ω
RL = 100 Ω
RL = 25 Ω
VCC = ±15 V
RF = 1 kΩ
Gain = 1
VI = 200 mV
RL = 50 Ω
Figure 23
NORMALIZED OUTPUT RESPONSE
vs
FREQUENCY
1M
f – Frequency – Hz
100k 10M 100M
Normalized Output Response – dB
VCC = ±15 V
RF = 1 kΩ
Gain = 2
VI = 200 mV
500M
–3
–5
–7
–9
–4
–6
–8
–1
0
–2
1
RL = 200 Ω
RL = 100 Ω
RL = 50 Ω
RL = 25 Ω
Figure 24
NORMALIZED OUTPUT RESPONSE
vs
FREQUENCY
1M
f – Frequency – Hz
100k 10M 100M
Normalized Output Response – dB
VCC = ±15 V
RL = 100 Ω
Gain = 1
VI = 200 mV
500M
–1
–3
–5
–7
–2
–4
–6
1
2
0
3
RF = 620 Ω
RF = 820 Ω
RF = 1 kΩ
Figure 25
NORMALIZED OUTPUT RESPONSE
vs
FREQUENCY
1M
f – Frequency – Hz
100k 10M 100M
Normalized Output Response – dB
VCC = ±15 V
RL = 100 Ω
Gain = 2
VI = 200 mV
500M
–1
–3
–5
–2
–4
–6
1
2
0
3
RF = 430 Ω
RF = 1 kΩ
RF = 620 Ω
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
15
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 26
f – Frequency – Hz
SMALL AND LARGE SIGNAL FREQUENCY RESPONSE
–27
–30
10M100k
–18
–15
500M
Output Level – dBV
–12
–3
1M 100M
–9
–6
VI = 500 mV
–24
–21
Gain = 1
VCC = ± 15 V
RF = 820 Ω
RL = 25 Ω
VI = 250 mV
VI = 125 mV
VI = 62.5 mV
Figure 27
f – Frequency – Hz
SMALL AND LARGE SIGNAL FREQUENCY RESPONSE
–21
–24
10M100k
–12
–9
500M
Output Level – dBV
–6
3
1M 100M
–3
0
VI = 500 mV
–18
–15
Gain = 2
VCC = ± 15 V
RF = 680 Ω
RL = 25 Ω
VI = 250 mV
VI = 125 mV
VI = 62.5 mV
Figure 28
f – Frequency – Hz
SINGLE–ENDED HARMONIC DISTORTION
vs
FREQUENCY
–90
–100
1M
–60
100k
–80
–70
10M
Single–Ended Harmonic Distortion (dBc)
–50
–40
2nd Harmonic
3rd Harmonic
VCC = ± 15 V
Gain = 2
RF = 680 Ω
RL = 25 Ω
V
O(PP)
= 2V
Figure 29
f – Frequency – Hz
SINGLE–ENDED HARMONIC DISTORTION
vs
FREQUENCY
–90
–100
1M
–60
100k
–80
–70
10M
Single–Ended Harmonic Distortion (dBc)
–50
–40
2nd Harmonic
3rd Harmonic
VCC = ± 5 V
Gain = 2
RF = 680 Ω
RL = 25 Ω
V
O(PP)
= 2V
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
16
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Figure 30
V
O(PP)
– Output Voltage – V
SINGLE–ENDED HARMONIC DISTORTION
vs
OUTPUT VOLTAGE
–90
–100
10
0
–80
–70
20
Single–Ended Harmonic Distortion (dBc)
–60
–50
515
2nd Harmonic
3rd Harmonic
VCC = ± 15 V
Gain = 2
RF = 680 Ω
RL = 25 Ω
f = 1 MHz
Figure 31
V
O(PP)
– Output Voltage – V
SINGLE–ENDED HARMONIC DISTORTION
vs
OUTPUT VOLTAGE
–90
–100
2
0
–80
–70
4
Single–Ended Harmonic Distortion – dBc
–60
–50
13
2nd Harmonic
3rd Harmonic
VCC = ± 5 V
Gain = 2
RF = 680 Ω
RL = 25 Ω
f = 1 MHz
DC Input Offset Voltage – V
DIFFERENTIAL GAIN AND PHASE
vs
DC INPUT OFFSET VOLTAGE
0.03
0.02
0.01
0
–0.5 –0.1–0.3 0.1 0.7
0.05
–0.7
0.3 0.5
0.04
VCC = ±15 V
RL = 150 Ω
RF = 1 kΩ
f = 3.58 MHz
Gain = 2
40 IRE Modulation
0.02
0
0.04
Differential Phase –
°
0.06
0.08
0.10
Phase
Differential Gain – %
Gain
Figure 32
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
17
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
DC Input Offset Voltage – V
DIFFERENTIAL GAIN AND PHASE
vs
DC INPUT OFFSET VOLTAGE
0.03
0.02
0.01
0
–0.5 –0.1–0.3 0.1 0.7
0.05
–0.7
0.3 0.5
0.04
VCC = ±5 V
RL = 150 Ω
RF = 1 kΩ
f = 3.58 MHz
Gain = 2
40 IRE Modulation
0.02
0
0.04
Differential Phase –
°
0.06
0.08
0.10
Gain
Differential Gain – %
Phase
Figure 33
Number of 150-Ω Loads
DIFFERENTIAL GAIN AND PHASE
vs
NUMBER OF 150-Ω LOADS
0.09
0.06
0.03
0
28
0.15
1
4
Differential Gain – %
0.12
VCC = ±15 V
RF = 1 kΩ
Gain = 2
f = 3.58 MHz
40 IRE Modulation
100 IRE Ramp
0.05
0
0.10
Differential Phase –
°
0.15
0.20
0.25
Gain
Phase
3567
Figure 34
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
18
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
Number of 150-Ω Loads
DIFFERENTIAL GAIN AND PHASE
vs
NUMBER OF 150-Ω LOADS
0.09
0.06
0.03
0
0.15
Differential Gain – %
0.12
0.05
0
0.10
Differential Phase –
°
0.15
0.20
0.25
VCC = ±5 V
RF = 1 kΩ
Gain = 2
f = 3.58 MHz
40 IRE Modulation
100 IRE Ramp
Gain
Phase
28143567
Figure 35
Figure 36
t – Time – ns
VCC = ±15 V
Gain = 2
RL = 25 Ω
RF = 1 kΩ
tr/tf= 300 ps
See Figure 3
400-mV STEP RESPONSE
100
–100
0
–200
V
O
– Output Voltage – mV
300
200
0 15010050 200 250 350300 400 450 500
400
–300
–400
Figure 37
t – Time – ns
10-V STEP RESPONSE
2
–2
0
–4
V
O
– Output Voltage – V
6
4
0 15010050 200 250 350300 400 450 500
8
–6
–8
VCC = ±15 V
Gain = 2
RL = 25 Ω
RF = 1 kΩ
tr/tf= 5 ns
See Figure 3
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
19
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
TYPICAL CHARACTERISTICS
t – Time – ns
VCC = ±15 V
Gain = 5
RL = 25 Ω
RF = 2 kΩ
tr/tf= 5 ns
See Figure 3
20-V STEP RESPONSE
4
–4
0
–8
V
O
– Output Voltage – V
12
8
0 15010050 200 250 350300 400 450 500
16
–12
–16
Figure 38
APPLICATION INFORMATION
The THS6012 contains two independent operational amplifiers. These amplifiers are current feedback topology
amplifiers made for high-speed operation. They have been specifically designed to deliver the full power
requirements of ADSL and therefore can deliver output currents of at least 400 mA at full output voltage.
The THS6012 is fabricated using Texas Instruments 30-V complementary bipolar process, HVBiCOM. This
process provides excellent isolation and high slew rates that result in the device’s excellent crosstalk and
extremely low distortion.
independent power supplies
Each amplifier of the THS6012 has its own power supply pins. This was specifically done to solve a problem
that often occurs when multiple devices in the same package share common power pins. This problem is
crosstalk between the individual devices caused by currents flowing in common connections. Whenever the
current required by one device flows through a common connection shared with another device, this current,
in conjunction with the impedance in the shared line, produces an unwanted voltage on the power supply . Proper
power supply decoupling and good device power supply rejection helps to reduce this unwanted signal. What
is left is crosstalk.
However, with independent power supply pins for each device, the effects of crosstalk through common
impedance in the power supplies is more easily managed. This is because it is much easier to achieve low
common impedance on the PCB with copper etch than it is to achieve low impedance within the package with
either bond wires or metal traces on silicon.
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
20
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
power supply restrictions
Although the THS6012 is specified for operation from power supplies of ±5 V to ±15 V (or singled-ended power
supply operation from 10 V to 30 V), and each amplifier has its own power supply pins, several precautions must
be taken to assure proper operation.
1. The power supplies for each amplifier must be the same value. For example, if the driver 1 uses ±15 volts,
then the driver 2 must also use ±15 volts. Using ±15 volts for one amplifier and ±5 volts for another amplifier
is not allowed.
2. T o save power by powering down one of the amplifiers in the package, the following rules must be followed.
• The amplifier designated driver 1 must always receive power. This is because the internal startup
circuitry uses the power from the driver 1 device.
• The –V
CC
pins from both drivers must always be at the same potential.
• Driver 2 is powered down by simply opening the +V
CC
connection.
The THS6012 incorporates a standard Class A-B output stage. This means that some of the quiescent current
is directed to the load as the load current increases. So under heavy load conditions, accurate power dissipation
calculations are best achieved through actual measurements. For small loads, however, internal power
dissipation for each amplifier in the THS6012 can be approximated by the following formula:
PD≅ǒ2VCCI
CC
Ǔ)ǒ
VCC_V
O
Ǔ
ǒ
V
O
R
L
Ǔ
Where:
P
D
= Power dissipation for one amplifier
VCC= Split supply voltage
I
CC
= Supply current for that particular amplifier
VO= Output voltage of amplifier
RL= Load resistance
To find the total THS6012 power dissipation, we simply sum up both amplifier power dissipation results.
Generally , the worst case power dissipation occurs when the output voltage is one-half the V
CC
voltage. One
last note, which is often overlooked: the feedback resistor (RF) is also a load to the output of the amplifier and
should be taken into account for low value feedback resistors.
device protection features
The THS6012 has two built-in protection features that protect the device against improper operation. The first
protection mechanism is output current limiting. Should the output become shorted to ground the output current
is automatically limited to the value given in the data sheet. While this protects the output against excessive
current, the device internal power dissipation increases due to the high current and large voltage drop across
the output transistors. Continuous output shorts are not recommended and could damage the device.
Additionally , connection of the amplifier output to one of the supply rails (±V
CC
) can cause failure of the device
and is not recommended.
The second built-in protection feature is thermal shutdown. Should the internal junction temperature rise above
approximately 180_C, the device automatically shuts down. Such a condition could exist with improper heat
sinking or if the output is shorted to ground. When the abnormal condition is fixed, the internal thermal shutdown
circuit automatically turns the device back on.
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
21
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
thermal information
The THS6012 is packaged in a thermally-enhanced DWP package, which is a member of the PowerP AD family
of packages. This package is constructed using a downset leadframe upon which the die is mounted
[see Figure 39(a) and Figure 39(b)]. This arrangement results in the lead frame being exposed as a thermal pad
on the underside of the package [see Figure 39(c)]. Because this thermal pad has direct thermal contact with
the die, excellent thermal performance can be achieved by providing a good thermal path away from the thermal
pad.
The PowerP AD package allows for both assembly and thermal management in one manufacturing operation.
During the surface-mount solder operation (when the leads are being soldered), the thermal pad can also be
soldered to a copper area underneath the package. Through the use of thermal paths within this copper area,
heat can be conducted away from the package into either a ground plane or other heat dissipating device. This
is discussed in more detail in the
PCB design considerations
section of this document.
The PowerPAD package represents a breakthrough in combining the small area and ease of assembly of
surface mount with the, heretofore, awkward mechanical methods of heatsinking.
DIE
Side View (a)
End View (b)
Bottom View (c)
DIE
Thermal
Pad
NOTE A: The thermal pad is electrically isolated from all terminals in the package.
Figure 39. Views of Thermally Enhanced DWP Package
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
22
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
recommended feedback and gain resistor values
As with all current feedback amplifiers, the bandwidth of the THS6012 is an inversely proportional function of
the value of the feedback resistor. This can be seen from Figures 17 – 20. The recommended resistors with a
±15 V power supply for the optimum frequency response with a 25-Ω load system are 680-Ω for a gain = 1 and
620-Ω for a gain = 2 or –1. Additionally, using a ±5 V power supply, it is recommended that a 1-kΩ feedback
resistor be used for a gain of 1 and a 820-Ω feedback resistor be used for a gain of 2 or –1. These should be
used as a starting point and once optimum values are found, 1% tolerance resistors should be used to maintain
frequency response characteristics. Because there is a finite amount of output resistance of the operational
amplifier, load resistance can play a major part in frequency response. This is especially true with these drivers,
which tend to drive low-impedance loads. This can be seen in Figure 1 1, Figure 23, and Figure 24. As the load
resistance increases, the output resistance of the amplifier becomes less dominant at high frequencies. To
compensate for this, the feedback resistor should change. For 100-Ω loads, it is recommended that the
feedback resistor be changed to 820 Ω for a gain of 1 and 560 Ω for a gain of 2 or –1. Although, for most
applications, a feedback resistor value of 1 kΩ is recommended, which is a good compromise between
bandwidth and phase margin that yields a very stable amplifier.
Consistent with current feedback amplifiers, increasing the gain is best accomplished by changing the gain
resistor, not the feedback resistor . This is because the bandwidth of the amplifier is dominated by the feedback
resistor value and internal dominant-pole capacitor. The ability to control the amplifier gain independently of the
bandwidth constitutes a major advantage of current feedback amplifiers over conventional voltage feedback
amplifiers. Therefore, once a frequency response is found suitable to a particular application, adjust the value
of the gain resistor to increase or decrease the overall amplifier gain.
Finally, it is important to realize the effects of the feedback resistance on distortion. Increasing the resistance
decreases the loop gain and increases the distortion. It is also important to know that decreasing load
impedance increases total harmonic distortion (THD). Typically, the third order harmonic distortion increases
more than the second order harmonic distortion.
offset voltage
The output offset voltage, (VOO) is the sum of the input offset voltage (VIO) and both input bias currents (IIB) times
the corresponding gains. The following schematic and formula can be used to calculate the output offset
voltage:
VOO+
V
IO
ǒ
1
) ǒ
R
F
R
G
Ǔ
Ǔ
"
I
IB
)
R
S
ǒ
1
) ǒ
R
F
R
G
Ǔ
Ǔ
"
I
IB–RF
+
–
V
I
+
R
G
R
S
R
F
I
IB–
V
O
I
IB+
Figure 40. Output Offset Voltage Model
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
23
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
noise calculations and noise figure
Noise can cause errors on very small signals. This is especially true for the amplifying small signals. The noise
model for current feedback amplifiers (CFB) is the same as voltage feedback amplifiers (VFB). The only
difference between the two is that the CFB amplifiers generally specify different current noise parameters for
each input while VFB amplifiers usually only specify one noise current parameter. The noise model is shown
in Figure 42. This model includes all of the noise sources as follows:
• e
n
= Amplifier internal voltage noise (nV/√Hz)
• IN+ = Noninverting current noise (pA/√Hz)
• IN– = Inverting current noise (pA/√Hz)
• e
Rx
= Thermal voltage noise associated with each resistor (eRx = 4 kTRx)
_
+
R
F
R
S
R
G
e
Rg
e
Rf
e
Rs
e
n
IN+
Noiseless
IN–
e
ni
e
no
Figure 41. Noise Model
The total equivalent input noise density (eni) is calculated by using the following equation:
eni+
ǒ
e
n
Ǔ
2
)ǒIN
)
R
S
Ǔ
2
)ǒIN–
ǒRFø
R
G
Ǔ
Ǔ
2
)
4kTRs)
4kTǒRFø
R
G
Ǔ
Ǹ
Where:
k = Boltzmann’s constant = 1.380658 × 10
–23
T = Temperature in degrees Kelvin (273 +°C)
RF || RG = Parallel resistance of RF and R
G
To get the equivalent output noise of the amplifier, just multiply the equivalent input noise density (eni) by the
overall amplifier gain (AV).
eno+
eniAV+
e
ni
ǒ
1
)
R
F
R
G
Ǔ
(Noninverting Case)
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
24
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
noise calculations and noise figure (continued)
As the previous equations show, to keep noise at a minimum, small value resistors should be used. As the
closed-loop gain is increased (by reducing RG), the input noise is reduced considerably because of the parallel
resistance term. This leads to the general conclusion that the most dominant noise sources are the source
resistor (RS) and the internal amplifier noise voltage (en). Because noise is summed in a root-mean-squares
method, noise sources smaller than 25% of the largest noise source can be effectively ignored. This can greatly
simplify the formula and make noise calculations much easier to calculate.
This brings up another noise measurement usually preferred in RF applications, the noise figure (NF). Noise
figure is a measure of noise degradation caused by the amplifier. The value of the source resistance must be
defined and is typically 50 Ω in RF applications.
NF+10log
ȧ
ȧ
ȱ
Ȳ
e
2
ni
ǒ
e
Rs
Ǔ
2
ȧ
ȧ
ȳ
ȴ
Because the dominant noise components are generally the source resistance and the internal amplifier noise
voltage, we can approximate noise figure as:
NF+10log
ȧ
ȧ
ȧ
ȧ
ȧ
ȱ
Ȳ
1
)
ȧ
ȡ
Ȣ
ǒ
e
n
Ǔ
2
)ǒIN
)
R
S
Ǔ
2
ȧ
ȣ
Ȥ
4kTR
S
ȧ
ȧ
ȧ
ȧ
ȧ
ȳ
ȴ
Figure 42 shows the noise figure graph for the THS6012.
NOISE FIGURE
vs
SOURCE RESISTANCE
18
14
0
20
16
12
Noise Figure – dB
Rs – Source Resistance – Ω
10 100 1k 10k
TA = 25°C
8
4
10
6
2
Figure 42. Noise Figure vs Source Resistance
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
25
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
driving a capacitive load
Driving capacitive loads with high performance amplifiers is not a problem as long as certain precautions are
taken. The first is to realize that the THS6012 has been internally compensated to maximize its bandwidth and
slew rate performance. When the amplifier is compensated in this manner, capacitive loading directly on the
output will decrease the device’s phase margin leading to high frequency ringing or oscillations. Therefore, for
capacitive loads of greater than 10 pF, it is recommended that a resistor be placed in series with the output of
the amplifier, as shown in Figure 44. A minimum value of 10 Ω should work well for most applications. For
example, in 75-Ω transmission systems, setting the series resistor value to 75 Ω both isolates any capacitance
loading and provides the proper line impedance matching at the source end.
+
_
THS6012
C
LOAD
1 kΩ
Input
Output
1 kΩ
10 Ω
Figure 43. Driving a Capacitive Load
PCB design considerations
Proper PCB design techniques in two areas are important to assure proper operation of the THS6012. These
areas are high-speed layout techniques and thermal-management techniques. Because the THS6012 is a
high-speed part, the following guidelines are recommended.
D
Ground plane – It is essential that a ground plane be used on the board to provide all components with a
low inductive ground connection. Although a ground connection directly to a terminal of the THS6012 is not
necessarily required, it is recommended that the thermal pad of the package be tied to ground. This serves
two functions. It provides a low inductive ground to the device substrate to minimize internal crosstalk and
it provides the path for heat removal.
D
Input stray capacitance – To minimize potential problems with amplifier oscillation, the capacitance at the
inverting input of the amplifiers must be kept to a minimum. T o do this, PCB trace runs to the inverting input
must be as short as possible, the ground plane must be removed under any etch runs connected to the
inverting input, and external components should be placed as close as possible to the inverting input. This
is especially true in the noninverting configuration. An example of this can be seen in Figure 44, which shows
what happens when 1.8 pF is added to the inverting input terminal in the noninverting configuration. The
bandwidth increases dramatically at the expense of peaking. This is because some of the error current is
flowing through the stray capacitor instead of the inverting node of the amplifier. Although, in the inverting
mode, stray capacitance at the inverting input has little effect. This is because the inverting node is at a
virtual ground
and the voltage does not fluctuate nearly as much as in the noninverting configuration.
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
26
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
PCB design considerations (continued)
f – Frequency – Hz
Normalized Frequency Response – dB
100 1M 10M 100M 500M
–1
–3
–5
–7
–2
–4
–6
3
1
2
0
NORMALIZED FREQUENCY RESPONSE
vs
FREQUENCY
VCC = ±15 V
VI = 200 mV
RL = 25 Ω
RF = 1 kΩ
Gain = 1
CI = 0 pF
(Stray C Only)
CI = 1.8 pF
+
–
1 kΩ
50 Ω
RL =
25 Ω
C
in
V
in
V
out
Figure 44. Driver Normalized Frequency Response vs Frequency
D
Proper power supply decoupling – Use a minimum of a 6.8-µF tantalum capacitor in parallel with a 0.1-µF
ceramic capacitor on each supply terminal. It may be possible to share the tantalum among several
amplifiers depending on the application, but a 0.1-µF ceramic capacitor should always be used on the
supply terminal of every amplifier. In addition, the 0.1-µF capacitor should be placed as close as possible
to the supply terminal. As this distance increases, the inductance in the connecting etch makes the capacitor
less effective. The designer should strive for distances of less than 0.1 inches between the device power
terminal and the ceramic capacitors.
Because of its power dissipation, proper thermal management of the THS6012 is required. Although there are
many ways to properly heatsink this device, the following steps illustrate one recommended approach for a
multilayer PCB with an internal ground plane.
1. Prepare the PCB with a top side etch pattern as shown in Figure 45. There should be etch for the leads as
well as etch for the thermal pad.
2. Place 18 holes in the area of the thermal pad. These holes should be 13 mils in diameter. They are kept
small so that solder wicking through the holes is not a problem during reflow.
3. It is recommended, but not required, to place six more holes under the package, but outside the thermal
pad area. These holes are 25 mils in diameter. They may be larger because they are not in the area to be
soldered so that wicking is not a problem.
4. Connect all 24 holes, the 18 within the thermal pad area and the 6 outside the pad area, to the internal
ground plane.
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
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POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
PCB design considerations (continued)
5. When connecting these holes to the ground plane, do not use the typical web or spoke via connection
methodology . Web connections have a high thermal resistance connection that is useful for slowing the heat
transfer during soldering operations. This makes the soldering of vias that have plane connections easier.
However, in this application, low thermal resistance is desired for the most ef ficient heat transfer. Therefore,
the holes under the THS6012 package should make their connection to the internal ground plane with a
complete connection around the entire circumference of the plated through hole.
6. The top-side solder mask should leave exposed the terminals of the package and the thermal pad area with
its five holes. The four larger holes outside the thermal pad area, but still under the package, should be
covered with solder mask.
7. Apply solder paste to the exposed thermal pad area and all of the operational amplifier terminals.
8. With these preparatory steps in place, the THS6012 is simply placed in position and run through the solder
reflow operation as any standard surface-mount component. This results in a part that is properly installed.
Addition 6 vias outside of thermal pad area
but under the package
(Via diameter = 25 mils)
Thermal pad area (0.19 x 0.21) with 18 vias
(Via diameter = 13 mils)
Figure 45. PowerPAD PCB Etch and Via Pattern
The actual thermal performance achieved with the THS6012 in its PowerPAD package depends on the
application. In the previous example, if the size of the internal ground plane is approximately 3 inches × 3 inches,
then the expected thermal coefficient, θ
JA
, is about 21.5_C/W. For a given θJA, the maximum power dissipation
is shown in Figure 46 and is calculated by the following formula:
PD+
ǒ
T
MAX–TA
q
JA
Ǔ
Where:
P
D
= Maximum power dissipation of THS6012 (watts)
T
MAX
= Absolute maximum junction temperature (150°C)
T
A
= Free-ambient air temperature (°C)
θ
JA
= θJC + θ
CA
θJC = Thermal coefficient from junction to case (0.37°C/W)
θ
CA
= Thermal coefficient from case to ambient
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
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POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
PCB design considerations (continued)
More complete details of the PowerP AD installation process and thermal management techniques can be found
in the Texas Instruments Technical Brief,
PowerPAD Thermally Enhanced Package.
This document can be
found at the TI web site (www.ti.com) by searching on the key word PowerPAD. The document can also be
ordered through your local TI sales office. Refer to literature number SLMA002 when ordering.
TA – Free-Air Temperature – °C
–40 –20 0 20 80 1006040
MAXIMUM POWER DISSIPATION
vs
FREE-AIR TEMPERATURE
5
3
1
0
4
2
6
7
Maximum Power Dissipation – W
8
9
TJ = 150°C
PCB Size = 3” x 3”
No Air Flow
θJA = 21.5°C/W
2 oz Trace and
Copper Pad
with Solder
θJA = 43.9°C/W
2 oz Trace and Copper Pad
without Solder
Figure 46. Maximum Power Dissipation vs Free-Air Temperature
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
29
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
ADSL
The THS6012 was primarily designed as a line driver and line receiver for ADSL (asymmetrical digital subscriber
line). The driver output stage has been sized to provide full ADSL power levels of 20 dBm onto the telephone
lines. Although actual driver output peak voltages and currents vary with each particular ADSL application, the
THS6012 is specified for a minimum full output current of 400 mA at its full output voltage of approximately 12
V . This performance meets the demanding needs of ADSL at the central office end of the telephone line. A typical
ADSL schematic is shown in Figure 47.
_
+
6.8 µF0.1 µF
–15 V
6.8 µF0.1 µF
15 V
1 kΩ
1 kΩ
+
+
V
I+
_
+
6.8 µF0.1 µF
–15 V
6.8 µF0.1 µF
15 V
1 kΩ
1 kΩ
+
+
V
I–
+
–
1 kΩ
0.1 µF
1 kΩ
2 kΩ
12.5 Ω
+
–
1 kΩ
0.1 µF
1 kΩ
2 kΩ
1:2
Telephone Line
12.5 Ω
–15 V
15 V
15 V
–15 V
0.01 µF
THS6062
THS6062
V
O+
V
O–
THS6012
Driver 1
THS6012
Driver 2
100 Ω
Figure 47. THS6012 ADSL Application
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
30
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
ADSL (continued)
The ADSL transmit band consists of 255 separate carrier frequencies each with its own modulation and
amplitude level. With such an implementation, it is imperative that signals put onto the telephone line have as
low a distortion as possible. This is because any distortion either interferes directly with other ADSL carrier
frequencies or it creates intermodulation products that interfere with ADSL carrier frequencies.
The THS6012 has been specifically designed for ultra low distortion by careful circuit implementation and by
taking advantage of the superb characteristics of the complementary bipolar process. Driver single-ended
distortion measurements are shown in Figures 29 – 32. It is commonly known that in the differential driver
configuration, the second order harmonics tend to cancel out. Thus, the dominant total harmonic distortion
(THD) will be primarily due to the third order harmonics. For these tests the load was 25 Ω. Additionally , distortion
should be reduced as the feedback resistance drops. This is because the bandwidth of the amplifier increases,
which allows the amplifier to react faster to any nonlinearities in the closed-loop system.
Another significant point is the fact that distortion decreases as the impedance load increases. This is because
the output resistance of the amplifier becomes less significant as compared to the output load resistance.
general configurations
A common error for the first-time CFB user is to create a unity gain buffer amplifier by shorting the output directly
to the inverting input. A CFB amplifier in this configuration oscillates and is not recommended. The THS6012,
like all CFB amplifiers, must have a feedback resistor for stable operation. Additionally, placing capacitors
directly from the output to the inverting input is not recommended. This is because, at high frequencies, a
capacitor has a very low impedance. This results in an unstable amplifier and should not be considered when
using a current-feedback amplifier. Because of this, integrators and simple low-pass filters, which are easily
implemented on a VFB amplifier, have to be designed slightly dif ferently . If filtering is required, simply place an
RC-filter at the noninverting terminal of the operational-amplifier (see Figure 49).
V
I
V
O
C1
+
–
R
G
R
F
R1
f
–3dB
+
1
2pR1C1
V
O
V
I
+ ǒ
1
)
R
F
R
G
Ǔ
ǒ
1
1)sR1C1
Ǔ
Figure 48. Single-Pole Low-Pass Filter
If a multiple pole filter is required, the use of a Sallen-Key filter can work very well with CFB amplifiers. This is
because the filtering elements are not in the negative feedback loop and stability is not compromised. Because
of their high slew-rates and high bandwidths, CFB amplifiers can create very accurate signals and help minimize
distortion. An example is shown in Figure 50.
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
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POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
general configurations (continued)
V
I
C2
R2R1
C1
R
F
R
G
R1 = R2 = R
C1 = C2 = C
Q = Peaking Factor
(Butterworth Q = 0.707)
(
=
1
Q
2 –
)
R
G
R
F
_
+
f
–3dB
+
1
2pRC
Figure 49. 2-Pole Low-Pass Sallen-Key Filter
There are two simple ways to create an integrator with a CFB amplifier. The first one shown in Figure 51 adds
a resistor in series with the capacitor. This is acceptable because at high frequencies, the resistor is dominant
and the feedback impedance never drops below the resistor value. The second one shown in Figure 52 uses
positive feedback to create the integration. Caution is advised because oscillations can occur because of the
positive feedback.
+
–
C1
R
F
R
G
V
O
V
I
THS6012
V
O
V
I
+ ǒ
R
F
R
G
Ǔ
ȧ
ȡ
Ȣ
S
)
1
RFC1
S
ȧ
ȣ
Ȥ
Figure 50. Inverting CFB Integrator
+
–
R
F
V
O
R
G
R2R1
C1
R
A
V
I
THS6012
For Stable Operation:
R2
R1 || R
A
≥
R
F
R
G
sR1C1
(
)
R
F
R
G
1 +
VO
≅ V
I
Figure 51. Non-Inverting CFB Integrator
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
32
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
APPLICATION INFORMATION
general configurations (continued)
Another good use for the THS6012 amplifiers is as very good video distribution amplifiers. One characteristic
of distribution amplifiers is the fact that the differential phase (DP) and the differential gain (DG) are
compromised as the number of lines increases and the closed-loop gain increases. Be sure to use termination
resistors throughout the distribution system to minimize reflections and capacitive loading.
+
–
620 Ω620 Ω
75 Ω
75 Ω
75 Ω
75 Ω
75 Ω
N Lines
V
O1
V
ON
THS6012
75 Ω Transmission Line
V
I
Figure 52. Video Distribution Amplifier Application
evaluation board
An evaluation board is available for the THS6012 (literature number SLOP132). This board has been configured
for proper thermal management of the THS6012. The circuitry has been designed for a typical ADSL application
as shown previously in this document. For more detailed information, refer to the
THS6012EVM User’s Manual
(literature number SLOU034). To order the evaluation board contact your local TI sales office or distributor.
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
33
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
MECHANICAL INFORMATION
DWP (R-PDSO-G20) PowerPAD PLASTIC SMALL-OUTLINE PACKAGE
Gage Plane
0.411 (10,44)
0.430 (10,92)
0.010 (0,25) NOM
0.010 (0,25)
Seating Plane
0.050 (1,27)
0.016 (0,40)
4073226/B 01/96
11
10
0.500 (12,70)
0.510 (12,95)
20
1
0.004 (0,10)
0.000 (0,00)
0.096 (2,43) MAX
0.020 (0,51)
0.014 (0,35)
0.293 (7,45)
0.299 (7,59)
0.050 (1,27)
M
0.010 (0,25)
0.004 (0,10)
+2°–8°
Thermal Pad 0.150 (3,81) 0.170 (4,31) NOM
(see Note C)
NOTES: A. All linear dimensions are in inches (millimeters).
B. This drawing is subject to change without notice.
C. The thermal performance may be enhanced by bonding the thermal pad to an external thermal plane.
PowerPAD is a trademark of Texas Instruments Incorporated.
THS6012
500-mA DUAL DIFFERENTIAL LINE DRIVER
SLOS226C– SEPTEMBER 1998 – REVISED FEBRUARY 2000
34
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265
MECHANICAL DATA
GQE (S-PLGA-N80) PLASTIC LAND GRID ARRAY
98765
J
H
G
F
E
D
321
C
B
A
4
4,00 TYP
Seating Plane
5,20
4,80
SQ
0,87
0,93
0,08 MAX
0,23
0,33
1,00 MAX
0,50
0,50
0,08
M
∅ 0,05
4200461/A 10/99
NOTES: A. All linear dimensions are in millimeters.
B. This drawing is subject to change without notice.
C. MicroStar Junior LGA configuration
MicroStar Junior LGA is a trademark of Texas Instruments Incorporated.
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