Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments
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SLES094A - OCTOBER 2003 - REVISED MARCH 2004
TM
FEATURES
D50 W per Channel (BTL) Into 6 Ω (Stereo)
D95-dB Dynamic Range With TAS5026
DLess Than 0.1% THD+N (1 W RMS Into 6 Ω)
DLess Than 0.2% THD+N (50 W RMS into 6 Ω)
DPower Efficiency Typically 90% Into 6-Ω Load
DSelf-Protecting Design (Undervoltage,
Overtemperature and Short Conditions) With
Error Reporting
DInternal Gate Drive Supply Voltage Regulator
DEMI Compliant When Used With
Recommended System Design
1
THD + NOISE vs OUTPUT POWER
RL = 6 Ω
TC = 75°C
APPLICATIONS
DDVD Receiver
DHome Theatre
DMini/Micro Component Systems
DInternet Music Appliance
DESCRIPTION
The TAS5112A is a high-performance, integrated stereo
digital amplifier power stage designed to drive 6-Ω
speakers at up to 50 W per channel. The device
incorporates TI’s PurePath Digitalt technology and is
used with a digital audio PWM processor (TAS50XX) and
a simple passive demodulation filter to deliver high-quality,
high-efficiency, true-digital audio amplification.
The efficiency of this digital amplifier is typically 90%,
reducing the size of both the power supplies and heatsinks
needed. Overcurrent protection, overtemperature
protection, and undervoltage protection are built into the
TAS5112A, safeguarding the device and speakers against
fault conditions that could damage the system.
1
THD + NOISE vs FREQUENCY
RL = 6 Ω
TC = 75°C
PO = 50 W
0.1
0.1
0.01
THD+N - Total Harmonic Distortion + Noise - %
0.01
100m
semiconductor products and disclaimers thereto appears at the end of this data sheet.
PurePath Digital and PowerPAD are trademarks of Texas Instruments.
Other trademarks are the property of their respective owners.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during
storage or handling to prevent electrostatic damage to the MOS gates.
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GENERAL INFORMATION
Terminal Assignment
The TAS5112A is offered in a thermally enhanced 56-pin
TSSOP DFD (thermal pad is on the top), shown as follows.
over operating free-air temperature range unless otherwise noted
TAS5112AUNITS
DVDD TO DGND–0.3 V to 4.2 V
GVDD TO GND33.5 V
PVDD_X TO GND (dc voltage)33.5 V
PVDD_X TO GND (spike voltage
OUT_X TO GND (dc voltage)33.5 V
OUT_X TO GND (spike voltage
BST_X TO GND (dc voltage)48 V
BST_X TO GND (spike voltage
GREG TO GND
PWM_XP, RESET, M1, M2, M3, SD,
OTW
Maximum operating junction
temperature, T
Storage temperature–40°C to 125°C
(1)
Stresses beyond those listed under “absolute maximum ratings”
may cause permanent damage to the device. These are stress
ratings only, and functional operation of the device at these or any
other conditions beyond those indicated under “recommended
operating conditions” is not implied. Exposure to absolutemaximum-rated conditions for extended periods may affect device
reliability.
(2)
The duration of voltage spike should be less than 100 ns; see
application note SLEA025.
(3)
GREG is treated as an input when the GREG pin is overdriven by
GVDD of 12 V .
(3)
J
(2)
)48 V
(2)
)48 V
(2)
)53 V
–0.3 V to DVDD + 0.3 V
–40°C to 150°C
14.2 V
(1)
Package Dissipation Ratings
R
PACKAGE
56-pin DFD TSSOP1.14See Note 4
(4)
The TAS5112A package is thermally enhanced for conductive
cooling using an exposed metal pad area. It is impractical to use the
device with the pad exposed to ambient air as the only heat sinking
of the device.
For this reason, R
thermal treatment, is provided in the Application Information section
of the data sheet. An example and discussion of typical system
R
values are provided in the Thermal Information section. This
θJA
example provides additional information regarding the power
dissipation ratings. This example should be used as a reference to
calculate the heat dissipation ratings for a specific application. TI
application engineering provides technical support to design
heatsinks if needed.
a system parameter that characterizes the
θJA,
θJC
(°C/W)
R
θJA
(°C/W)
Ordering Information
T
A
0°C to 70°CTAS5112ADFD56-pin small TSSOP
PACKAGEDESCRIPTION
For the most current specification and package
information, refer to our W eb site at www.ti.com.
2
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FUNCTION
(1)
DESCRIPTION
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
Terminal Functions
TERMINAL
NAMENO.
BST_A31PHigh-side bootstrap supply (BST), external capacitor to OUT_A required
BST_B42PHigh-side bootstrap supply (BST), external capacitor to OUT_B required
BST_C43PHS bootstrap supply (BST), external capacitor to OUT_C required
BST_D54PHS bootstrap supply (BST), external capacitor to OUT_D required
DGND23PDigital I/O reference ground
DREG16PDigital supply voltage regulator decoupling pin, capacitor connected to GND
DREG_RTN12PDigital supply voltage regulator decoupling return pin
DVDD25PI/O reference supply input (3.3 V)
GND1, 2, 22, 24,
27, 28, 29, 36,
37, 48, 49, 56
GREG3, 26PGate drive voltage regulator decoupling pin, capacitor to REG_GND
GVDD30, 55PVoltage supply to on-chip gate drive and digital supply voltage regulators
M1 (TST0)15IMode selection pin
M214IMode selection pin
M313IMode selection pin
OTW4OOvertemperature warning output, open drain with internal pullup resistor
OUT_A34, 35OOutput, half-bridge A
OUT_B38, 39OOutput, half-bridge B
OUT_C46, 47OOutput, half-bridge C
OUT_D50, 51OOutput, half-bridge D
PVDD_A32, 33PPower supply input for half-bridge A
PVDD_B40, 41PPower supply input for half-bridge B
PVDD_C44, 45PPower supply input for half-bridge C
PVDD_D52, 53PPower supply input for half-bridge D
PWM_AM20IInput signal (negative), half-bridge A
PWM_AP21IInput signal (positive), half-bridge A
PWM_BM18IInput signal (negative), half-bridge B
PWM_BP17IInput signal (positive), half-bridge B
PWM_CM10IInput signal (negative), half-bridge C
PWM_CP11IInput signal (positive), half-bridge C
PWM_DM8IInput signal (negative), half-bridge D
PWM_DP7IInput signal (positive), half-bridge D
RESET_AB19IReset signal, active low
RESET_CD9IReset signal, active low
SD_AB6OShutdown signal for half-bridges A and B, active-low
SD_CD5OShutdown signal for half-bridges C and D, active-low
(1)
I = input, O = Output, P = Power
PPower ground
3
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
FUNCTIONAL BLOCK DIAGRAM
PWM_AP
RESET
PWM
Receiver
Protection A
Protection B
GREG
Timing
Control
GREG
Gate
Drive
Gate
Drive
Gate
Drive
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BST_A
PVDD_A
OUT_A
GND
BST_B
PVDD_B
PWM_BP
OTW
SD
PWM
Receiver
To Protection
Blocks
OT
Protection
UVP
This diagram shows one channel.
Control
GREG
Timing
DREG
DREG
Gate
Drive
GREG
DREG_RTN
OUT_B
GND
DREG
GVDD
GREG
GREG
DREG_RTN
4
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Po
Output power
+ noise
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
RECOMMENDED OPERATING CONDITIONS
MINTYPMAXUNIT
DVDDDigital supply
GVDD
PVDD_xHalf-bridge supplyRelative to GND, RL= 6 Ω to 8 Ω029.530.5V
T
J
(1)
It is recommended for DVDD to be connected to DREG via a 100-Ω resistor.
Supply for internal gate drive and logic
regulators
Junction temperature0125_C
(1)
Relative to DGND33.33.6V
Relative to GND1629.530.5V
ELECTRICAL CHARACTERISTICS
PVDD_X = 29.5 V, GVDD = 29.5 V, DVDD connected to DREG via a 100-Ω resistor, RL = 6 Ω, 8X fs = 384 kHz, unless otherwise noted
TYPICALOVER TEMPERATURE
SYMBOLPARAMETERTEST CONDITIONS
AC PERFORMANCE, BTL Mode, 1 kHz
RL = 8 Ω, THD = 0.2%,
AES17 filter, 1 kHz
RL = 8 Ω, THD = 10%, AES17
filter, 1 kHz
RL = 6 Ω, THD = 0.2%,
AES17 filter, 1 kHz
RL = 6 Ω, THD = 10%, AES17
filter, 1 kHz
Po = 1 W/ channel, RL = 6 Ω,
AES17 filter
THD+N
V
n
SNRSignal-to-noise ratioA-weighted, AES17 filter96dBTyp
DRDynamic range
PVDD_x = 29.5 V, GVDD = 29.5 V, DVDD connected to DREG via a 100-Ω resistor, RL = 6 Ω, 8X fs = 384 kHz, unless otherwise noted
TYPICALOVER TEMPERATURE
SYMBOLPARAMETERTEST CONDITIONS
INPUT/OUTPUT PROTECTION
Set the DUT in normal
operation mode with all the
protections enabled. Sweep
uvp,G
OTW
OTE
OCOvercurrent protectionSee Note 1.6.7ATyp
STATIC DIGITAL SPECIFICATION
V
IL
OTW/SHUTDOWN (SD)
V
OL
(1)
To optimize device performance and prevent overcurrent (OC) protection tripping, the demodulation filter must be designed with special care.
See Demodulation Filter Design in the Application Information section of the data sheet and consider the recommended inductors and capacitors
for optimal performance. It is also important to consider PCB design and layout for optimum performance of the TAS5112A. It is recommended
to follow the TAS5112F2EVM (S/N 112) design and layout guidelines for best performance.
limit, GVDD
Overtemperature warning,
junction temperature
Overtemperature error,
junction temperature
PWM_AP, PWM_BP, M1,
M2, M3, SD, OTW
Low-level input voltage0.8VMax
Internally pull up R from
OTW/SD to DVDD
Low-level output voltageIO = 4 mA0.4VMax
SD output. Record the
GREG reading when SD is
triggered.
TA=25°CTA=25°C
6.9VMin
7.9VMax
125°CTyp
150°CTyp
DVDDVMax
-10µAMin
10µAMax
3022.5kΩMin
T
=
Case
75°C
2VMin
TA=40°C
TO 85°C
UNITS
MIN/TYP/
MAX
6
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Gate-Drive
PCB
SYSTEM CONFIGURATION USED FOR CHARACTERIZATION
Power Supply
External Power Supply
H-Bridge
Power Supply
TAS5112ADFD
ERR_RCVY
PWM_AP_1
PWM_AM_1
VALID_1
PWM PROCESSOR
TAS5026
PWM_AP_2
PWM_AM_2
VALID_2
1 µF
100 nF
100 Ω
100 nF
1 µF
1
GND
2
GND
3
GREG
4
OTW
5
SD_CD
6
SD_AB
7
PWM_DP
8
PWM_DM
9
RESET_CD
10
PWM_CM
11
PWM_CP
12
DREG_RTN
13
M3
14
M2
15
M1
16
DREG
17
PWM_BP
18
PWM_BM
19
RESET_AB
20
PWM_AM
21
PWM_AP
22
GND
23
DGND
24
GND
25
DVDD
26
GREG
27
GND
28
GND
BST_D
PVDD_D
PVDD_D
OUT_D
OUT_D
OUT_C
OUT_C
PVDD_C
PVDD_C
BST_C
BST_B
PVDD_B
PVDD_B
OUT_B
OUT_B
OUT_A
OUT_A
PVDD_A
PVDD_A
BST_A
GND
GVDD
GND
GND
GND
GND
GVDD
GND
56
55
1.5 Ω
54
53
52
51
50
49
48
47
46
45
44
43
1.5 Ω
42
41
40
39
38
37
36
35
34
33
32
31
1.5 Ω
30
29
33 nF
100 nF
1.5 Ω
1.5 Ω
†
33 nF
33 nF
100 nF
1.5 Ω
1.5 Ω
†
33 nF
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
100 nF
‡
L
PCB
10 µH
†
†
10 µH
100 nF
L
PCB
L
PCB
10 µH
10 µH
100 nF
L
PCB
470 nF
‡
‡
470 nF
‡
100 nF
100 nF
100 nF
100 nF
100 nF
4.7 kΩ
4.7 kΩ
4.7 kΩ
4.7 kΩ
1000 µF
1000 µF
†
Voltage suppressor diodes: 1SMA33CAT
‡
L
: Track in the PCB (1,0 mm wide and 50 mm long)
7
THD+N - Total Harmonic Distortion + Noise - %
THD+N - Total Harmonic Distortion + Noise - %
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
TYPICAL CHARACTERISTICS AND SYSTEM PERFORMANCE
OF TAS5112A EVM WITH TAS5026 PWM PROCESSOR
www.ti.com
TOTAL HARMONIC DISTORTION + NOISE
vs
FREQUENCY
1
RL = 6 Ω
TC = 75°C
PO = 50 W
0.1
PO = 10 W
PO = 1 W
0.01
0.001
201001k10k
f - Frequency - Hz
Figure 1
20k
NOISE AMPLITUDE
vs
FREQUENCY
0
RL = 6Ω
FFT = -60 dB
−20
TC = 75°C
TAS5026 Front End Device
−40
−60
−80
−100
Noise Amplitude - dBr
−120
−140
−160
0 2 4 6 8 10121416182022
f - Frequency - kHz
Figure 2
TOTAL HARMONIC DISTORTION + NOISE
10
RL = 6 Ω
TC = 75°C
1
0.1
0.01
100m
8
vs
OUTPUT POWER
110100
PO - Output Power - W
Figure 3
OUTPUT POWER
vs
H-BRIDGE VOLTAGE
60
TA = 75°C
50
40
RL = 6 Ω
30
- Output Power - W
20
O
P
10
0
048121620242832
VDD - Supply Voltage - V
Figure 4
RL = 8 Ω
www.ti.com
- System Output Stage Efficiency - %
P
- Output Power - W
k
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
SYSTEM OUTPUT STAGE EFFICIENCY
vs
OUTPUT POWER
100
90
80
70
60
50
40
30
20
η
10
0
0 5 10 15 20 25 30 35 40 45 50 55 60 65
PO - Output Power - W
Figure 5
f = 1 kHz
RL = 6 Ω
TC = 75°C
POWER LOSS
vs
OUTPUT POWER
11
f = 1 kHz
10
RL = 6 Ω
9
TC = 75°C
8
7
6
5
- Power Loss - W
4
tot
P
3
2
1
0
0 5 10 15 20 25 30 35 40 45 50 55 60 65
PO - Output Power - W
Figure 6
OUTPUT POWER
vs
CASE TEMPERATURE
60
PVDD = 29.5 V
58
RL = 6 Ω
56
54
52
50
48
O
46
44
42
40
020406080100120140
TC - Case Temperature - °C
Channel 2
Figure 7
Channel 1
AMPLITUDE
vs
FREQUENCY
3.0
2.5
2.0
1.5
1.0
0.5
0.0
−0.5
Amplitude - dBr
−1.0
−1.5
−2.0
−2.5
−3.0
101001k50
f - Frequency - Hz
RL = 8 Ω
Figure 8
RL = 6 Ω
10k
9
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
200
190
180
170
160
150
- On-State Resistance - mΩ
140
on
r
130
120
0 102030405060708090100
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ON-STATE RESISTANCE
vs
JUNCTION TEMPERATURE
TJ - Junction Temperature - °C
Figure 9
10
www.ti.com
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
THEORY OF OPERATION
POWER SUPPLIES
The power device only requires two supply voltages,
GVDD and PVDD_X.
GVDD is the gate drive supply for the device, regulated
internally down to approximately 12 V, and decoupled with
regards to board GND on the GREG pins through an
external capacitor. GREG powers both the low side and
high side via a bootstrap step-up conversion. The
bootstrap supply is charged after the first low-side turn-on
pulse. Internal digital core voltage DREG is also derived
from GVDD and regulated down by internal circuitry to
3.3 V.
The gate-driver regulator can be bypassed for reducing
idle loss in the device by shorting GREG to GVDD and
directly feeding in 12.0 V. This can be useful in an
application where thermal conduction of heat from the
device is difficult.
PVDD_X is the H-bridge power supply pin. T wo power pins
exists for each half-bridge to handle the current density . It
is important that the circuitry recommendations around the
PVDD_X pins are followed carefully both topology- and
layout-wise. For topology recommendations, see the
System Configuration Used for Characterization section.
Following these recommendations is important for
parameters like EMI, reliability, and performance.
POWERING UP
4.7-kΩ pulldown resistor on each PWM output node to
ground. This precharges the bootstrap supply capacitors
and discharges the output filter capacitor.
After GVDD has been applied, it takes approximately 800
µs to fully charge the BST capacitor. Within this time,
RESET must be kept low. After approximately 1 ms, the
back-end bootstrap capacitor is charged.
RESET can now be released if the modulator is powered
up and streaming valid PWM signals to the back-end
PWM_xP. Valid means a switching PWM signal which
complies with the frequency and duty cycle ranges stated
in the Recommended Operating Conditions.
A constant HIGH dc level on the PWM_xP is not permitted,
because it would force the high-side MOSFET ON until it
eventually ran out of BST capacitor energy and might
damage the device.
An unknown state of the PWM output signals from the
modulator is illegal and should be avoided, which in
practice means that the PWM processor must be powered
up and initialized before RESET is de-asserted HIGH to
the back end.
POWERING DOWN
For power down of the back end, an opposite approach is
necessary. The RESET must be asserted LOW before the
valid PWM signal is removed.
When PWM processors are used with TI PurePath Digital
amplifiers, the correct timing control of RESET and
PWM_xP is performed by the modulator.
> 1 ms
> 1 ms
RESET
GVDD
PVDD_X
PWM_xP
NOTE:
PVDD should not be powered up before GVDD.
During power up when RESET is asserted LOW, all
MOSFETs are turned off and the two internal half-bridges
are in the high-impedance state (Hi-Z). The bootstrap
capacitors supplying high-side gate drive are not charged
at this point. T o comply with the click and pop scheme and
use of non-TI modulators, it is recommended to use a
PRECAUTION
The TAS5112A must always start up in the
high-impedance (Hi-Z) state. In this state, the bootstrap
(BST) capacitor is precharged by a resistor on each PWM
output node to ground. See the system configuration. This
ensures that the back end is ready for receiving PWM
pulses, indicating either HIGH- or LOW-side turnon after
RESET is de-asserted to the back end.
With the following pulldown resistor and BST capacitor
size, the charge time is:
C = 33 nF, R = 4.7 kΩ
R × C × 5 = 775.5 µs
After GVDD has been applied, it takes approximately 800
µs to fully charge the BST capacitor. During this time,
RESET must be kept low. After approximately 1 ms the
back end BST is charged and ready. RESET can now be
released if the PWM modulator is ready and is streaming
valid PWM signals to the back end. Valid PWM signals are
switching PWM signals with a frequency between 350-400
kHz. A constant HIGH level on the PWM+ would force the
high-side MOSFET ON until it eventually ran out of BST
capacitor energy. Putting the device in this condition
should be avoided.
11
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
www.ti.com
In practice this means that the DVDD-to-PWM processor
(front-end) should be stable and initialization should be
completed before RESET is de-asserted to the back end.
CONTROL I/O
Shutdown Pin: SD
The SD pin functions as an output pin and is intended for
protection-mode signaling to, for example, a controller or
other front-end device. The pin is open-drain with an
internal pullup resistor to DVDD.
The logic output is, as shown in the following table, a
combination of the device state and RESET input:
SDRESETDESCRIPTION
00Reserved
01Device in protection mode, i.e., UVP and/or OC
and/or OT error
(2)
1
11Normal operation
Temperature Warning Pin: OTW
The OTW pin gives a temperature warning signal when
temperature exceeds the set limit. The pin is of the
open-drain type with an internal pullup resistor to DVDD.
Overall Reporting
The SD pin, together with the OTW pin, gives chip state
information as described in Table 1.
0Device set high-impedance (Hi-Z), SD forced high
(2)
SD is pulled high when RESET is asserted low independent
of chip state (i.e., protection mode). This is desirable to
maintain compatibility with some TI PWM front ends.
OTWDESCRIPTION
0Junction temperature higher than 125°C
1Junction temperature lower than 125°C
The device can be recovered by toggling RESET low and
then high, after all errors are cleared.
Overcurrent (OC) Protection
The device has individual forward current protection on
both high-side and low-side power stage FETs. The OC
protection works only with the demodulation filter present
at the output. See Demodulation Filter Design in the
Application Information section of the data sheet for design
constraints.
Overtemperature (OT) Protection
A dual temperature protection system asserts a warning
signal when the device junction temperature exceeds
125°C. The OT protection circuit is shared by all
half-bridges.
Undervoltage (UV) Protection
Undervoltage lockout occurs when GVDD is insufficient
for proper device operation. The UV protection system
protects the device under power-up and power-down
situations. The UV protection circuits are shared by all
half-bridges.
Reset Function
The reset has two functions:
D Reset is used for re-enabling operation after a
latching error event.
D Reset is used for disabling output stage
switching (mute function).
The error latch is cleared on the falling edge of reset and
normal operation is resumed when reset goes high.
Table 1. Error Signal Decoding
OTWSDDESCRIPTION
00Overtemperature error (OTE)
01Overtemperature warning (OTW)
10Overcurrent (OC) or undervoltage (UVP) error
11Normal operation, no errors/warnings
Chip Protection
The TAS5112A protection function is implemented in a
closed loop with, for example, a system controller and TI
PWM processor. The TAS5112A contains three individual
systems protecting the device against error conditions. All
of the error events covered result in the output stage being
set in a high-impedance state (Hi-Z) for maximum
protection of the device and connected equipment.
12
PROTECTION MODE
Autorecovery (AR) After Errors (PMODE0)
In autorecovery mode (PMODE0) the TAS5112A is
self-supported i n handling of error situations. All protection
systems are active, setting the output stage in the
high-impedance state to protect the output stage and
connected equipment. However, after a short time period
the device autorecovers, i.e., operation is automatically
resumed provided that the system is fully operational.
The autorecovery timing is set by counting PWM input
cycles, i.e., the timing is relative to the switching frequency.
The AR system is common to both half-bridges.
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A
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
Timing and Function
The function of the autorecovery circuit is as follows:
1. An error event occurs and sets the
protection latch (output stage goes Hi-Z).
2. The counter is started.
3. After n/2 cycles, the protection latch is
cleared but the output stage remains Hi-Z
(identical to pulling RESET
low).
4. After n cycles, operation is resumed
(identical to pulling RESET high) (n = 512).
Error
Protection
Latch
Shutdown
SD
Autorecovery
PWM
Counter
R-RESET
Figure 10. Autorecovery Function
Latching Shutdown on All Errors (PMODE1)
Table 3. Output Mode Selection
M3OUTPUT MODE
0Bridge-tied load output stage (BTL)
1Reserved
APPLICATION INFORMATION
DEMODULATION FILTER DESIGN AND
SPIKE CONSIDERATIONS
The output square wave is susceptible to overshoots
(voltage spikes). The spike characteristics depend on
many elements, including silicon design and application
design and layout. The device should be able to handle
narrow spike pulses, less than 65 ns, up to 65 volts peak.
For more detailed information, see TI application note
SLEA025.
The PurePath Digital amplifier outputs are driven by
heavy-duty DMOS transistors in an H-bridge
configuration. These transistors are either off or fully on,
which reduces the DMOS transistor on-state resistance,
R
, and the power dissipated in the device, thereby
DSon
increasing efficiency.
The result is a square-wave output signal with a duty cycle
that is proportional to the amplitude of the audio signal. It
is recommended that a second-order LC filter be used to
recover the audio signal. For this application, EMI is
considered important; therefore, the selected filter is the
full-output type shown in Figure 11.
TAS51xx
In latching shutdown mode, all error situations result in a
power down (output stage Hi-Z). Re-enabling can be done
by toggling the RESET pin.
All Protection Systems Disabled (PMODE2)
In PMODE2, all protection systems are disabled. This
mode is purely intended for testing and characterization
purposes and thus not recommended for normal device
operation.
MODE Pins Selection
The protection mode is selected by shorting M1/M2 to
DREG or DGND according to Table 2.
Table 2. Protection Mode Selection
M1 M2PROTECTION MODE
00Autorecovery after errors (PMODE 0)
01Latching shutdown on all errors (PMODE 1)
10All protection systems disabled (PMODE 2)
11Reserved
The output configuration mode is selected by shorting the
M3 pin to DREG or DGND according to Table 3.
Output A
Output B
L
L
C1A
C1B
C2
R
(Load)
Figure 11. Demodulation Filter
The main purpose of the output filter is to attenuate the
high-frequency switching component of the PurePath
Digital amplifier while preserving the signals in the audio
band.
Design of the demodulation filter affects the performance
of the power amplifier significantly. As a result, to ensure
proper operation of the overcurrent (OC) protection circuit
and meet the device THD+N specifications, the selection
of the inductors used in the output filter must be considered
according to the following. The rule is that the inductance
13
L - Inductance -
H
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
www.ti.com
should remain stable within the range of peak current seen
at maximum output power and deliver at least 5 µH of
inductance at 15 A.
If this rule is observed, the TAS5112A does not have
distortion issues due to the output inductors, and
overcurrent conditions do not occur due to inductor
saturation in the output filter.
Another par a meter to be considered is the idle current loss
in the inductor. This can be measured or specified as
inductor dissipation (D). The target specification for
dissipation is less than 0.05.
In general, 10-µH inductors suffice for most applications.
The frequency response of the amplifier is slightly altered
by the change in output load resistance; however, unless
tight control of frequency response is necessary (better
than 0.5 dB), it is not necessary to deviate from 10 µH.
The graphs in Figure 12 display the inductance vs current
characteristics of two inductors that are recommended for
use with the TAS5112A.
INDUCTANCE
vs
CURRENT
11
In order to minimize the EMI effect of unbalanced ripple
loss in the inductors, 0.1-µF 50-V SMD capacitors (X7R or
better) (C1A and C1B in Figure 11) should be added from
the output of each inductor to ground.
THERMAL INFORMATION
The thermally augmented package provided with the
TAS5112A is designed to be interfaced directly to
heatsinks using a thermal interface compound (for
example, Wakefield Engineering type 126 thermal
grease.) The heatsink then absorbs heat from the ICs and
couples it to the local air. If the heatsink is carefully
designed, this process can reach equilibrium and heat can
be continually removed from the ICs. Because of the
efficiency o f the TAS5112A, heatsinks can be smaller than
those required for linear amplifiers of equivalent
performance.
R
is a system thermal resistance from junction to
θ
JA
ambient ai r. As such, it is a system parameter with roughly
the following components:
D R
(the thermal resistance from junction to
θ
JC
case, or in this case the metal pad)
D Thermal grease thermal resistance
D Heatsink thermal resistance
µ
The selection of the capacitor that is placed across the
output of each inductor (C2in Figure 11) is simple. To
complete the output filter, use a 0.47-µF capacitor with a
voltage rating at least twice the voltage applied to the
output stage (PVDD).
This capacitor should be a good quality polyester dielectric
such as a Wima MKS2-047ufd/100/10 or equivalent.
10
DFB1310A
9
DASL983XX-1023
8
7
6
5
4
051015
I - Current - A
Figure 12. Inductance Saturation
R
has been provided in the General Information
θ
JC
section.
The thermal grease thermal resistance can be calculated
from the exposed pad area and the thermal grease
manufacturer’s area thermal resistance (expressed in
°C-in2/W). The area thermal resistance of the example
thermal grease with a 0.002-inch thick layer is about 0.1
°C-in2/W. The approximate exposed pad area is as
follows:
56-pin HTSSOP0.045 in
2
Dividing the example thermal grease area resistance by
the surface area gives the actual resistance through the
thermal grease for both ICs inside the package:
56-pin HTSSOP2.27 °C/W
The thermal resistance of thermal pads is generally
considerably higher than a thin thermal grease layer.
Thermal tape has an even higher thermal resistance.
Neither pads nor tape should be used with either of these
two packages. A thin layer of thermal grease with careful
clamping of the heatsink is recommended. It may be
difficult to achieve a layer 0.001-inch thick or less, so the
modeling below is done with a 0.002-inch thick layer,
which may be more representative of production thermal
grease thickness.
14
www.ti.com
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
Heatsink thermal resistance is generally predicted by the
heatsink vendor, modeled using a continuous flow
dynamics (CFD) model, or measured.
Thus, for a single monaural IC, the system R
= R
θ
JA
+
θ
JC
thermal grease resistance + heatsink resistance.
Table 4, Table 5, and Table 6 indicate modeled
parameters for one or two TAS5112A ICs on a single
heatsink. The final junction temperature is set at 110°C in
all cases. It is assumed that the thermal grease is 0.002
inch thick and that it is similar in performance to Wakefield
Type 126 thermal grease. It is important that the thermal
grease layer is ≤0.002 inches thick and that thermal pads
or tape are not used in the pad-to-heatsink interface due
to the high power density that results in these extreme
power cases.
Table 4. Case 1 (2 × 50 W Unclipped Into 6 Ω,
Both Channels in Same IC)
Ambient temperature25°C
Power to load (per channel)50 W (unclipped)
Power dissipation4.5 W
Delta T inside package
Delta T through thermal grease
Required heatsink thermal resistance4.2°C/W
Junction temperature110°C
System R
R
Junction temperature85°C + 25°C = 110°C
(1)
θJA
* power dissipation85°C
θJA
This case represents a stereo system with only one package. See
Case 2 and Case 2A if doing a full-power, 2-channel test in a
multichannel system.
(1)
56-Pin HTSSOP
10.2°C, note 2 ×
channel dissipation
37.1°C, note 2 ×
channel dissipation
19°C/W
Table 5. Case 2 (2 × 50 W Unclipped Into 6 Ω,
Channels in Separate Packages)
Ambient temperature25°C
Power to load (per channel)50 W (unclipped)
Power dissipation4.5 W
Delta T inside package5.1°C
Delta T through thermal grease18.6°C
Required heatsink thermal resistance6.9°C/W
Junction temperature110°C
System R
R
Junction temperature85°C + 25°C = 110°C
(1)
θJA
* power dissipation85°C
θJA
In this case, the power is separated into two packages. Note that
this allows a considerably smaller heatsink because twice as much
area is available for heat transfer through the thermal grease. For
this reason, separating the stereo channels into two ICs is
recommended in full-power stereo tests made on multichannel
systems.
19°C/W
(1)
56-Pin HTSSOP
Table 6. Case 2A (2 × 60 W Into 6 Ω, Channels in
Separate IC Packages)
Ambient temperature25°C
Power to load (per channel)60 W (10% THD)
Power dissipation per channel5.4 W
Delta T inside package
Delta T through thermal grease
Required heatsink thermal resistance5.3°C/W
Junction temperature110°C
System R
R
Junction temperature85°C + 25°C = 110°C
(1)
θJA
* power dissipation85°C
θJA
In this case, the power is also separated into two packages, but
overdriving causes clipping to 10% THD. In this case, the high
power requires extreme care in attachment of the heatsink to
ensure that the thermal grease layer is ≤ 0.002 inches thick. Note
that this power level should not be attempted with both channels in
a single IC because of the high power density through the thermal
grease layer.
(1)
56-Pin HTSSOP
6.1°C, note 2 ×
channel dissipation
22.3°C, note 2 ×
channel dissipation
15.9°C/W
15
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
Thermal
Pad
8,20 mm
7,20 mm
www.ti.com
Other things that can affect the audible click level:
D The spectrum of the click seems to follow the
speaker impedance vs. frequency curve—the
higher the impedance, the higher the click
energy.
D Crossover filters used between woofer and
tweeter in a speaker can have high impedance
in the audio band, which should be avoided if
possible.
Another way to look at it is that the speaker impulse
response is a major contributor to how the click energy is
shaped in the audio band and how audible the click will be.
The following mode transitions feature click and pop
reduction.
3,90 mm
2,98 mm
CLICK AND POP REDUCTION
TI modulators feature a pop and click reduction system
that controls the timing when switching starts and stops.
Going from nonswitching to switching operation causes a
spectral energy burst to occur within the audio bandwidth,
which is heard in the speaker as an audible click, for
instance, after having asserted RESET LH during a
system start-up.
To make this system work properly, the following design
rules must be followed when using the TAS5112A back
end:
D The relative timing between the PWM_AP/M_x
signals and their corresponding VALID_x signal
should not be skewed by inserting delays,
because this increases the audible amplitude
level of the click.
D The output stage must start switching from a
fully discharged output filter capacitor. Because
the output stage prior to operation is in the
high-impedance state, this is done by having a
passive pulldown resistor on each speaker
output to GND (see System Configuration Usedfor Characterization).
STATE
(1)
Normal
Mute→ Normal
(1)
Normal
Error recovery → Normal
(1)
Normal
Hard Reset→ Normal
(1)
Normal = switching
→ MuteYes
Error recovery
→
(ERRCVY)
→ Hard ResetNo
(1)
(1)
(1)
CLICK AND
POP REDUCED
Yes
Yes
Yes
Yes
REFERENCES
1. TAS5000 Digital Audio PWM Processor data
manual—TI (SLAS270)
2. True Digital Audio Amplifier TAS5001 Digital AudioPWM Processor data sheet—TI (SLES009)
3. True Digital Audio Amplifier TAS5010 Digital AudioPWM Processor data sheet—TI (SLAS328)
4. True Digital Audio Amplifier TAS5012 Digital AudioPWM Processor data sheet—TI (SLES006)
5. TAS5026 Six-Channel Digital Audio PWMProcessor data manual—TI (SLES041)
6. TAS5036A Six-Channel Digital Audio PWMProcessor data manual—TI (SLES061)
7. TAS3103 Digital Audio Processor With 3D Effects
data manual—TI (SLES038)
8. Digital Audio Measurements application report—TI
(SLAA114)
10. System Design Considerations for True DigitalAudio Power Amplifiers application report—TI
(SLAA117)
16
www.ti.com
SLES094A - OCTOBER 2003 - REVISED MARCH 2004
MECHANICAL DATA
17
PACKAGE OPTION ADDENDUM
www.ti.com
16-May-2005
PACKAGING INFORMATION
Orderable DeviceStatus
(1)
Package
Type
Package
Drawing
Pins Package
Qty
Eco Plan
TAS5112ADCAACTIVEHTSSOPDCA5635Green(RoHS &
no Sb/Br)
TAS5112ADCAG4ACTIVEHTSSOPDCA5635Green (RoHS &
no Sb/Br)
TAS5112ADCARACTIVEHTSSOPDCA562000 Green (RoHS &
no Sb/Br)
TAS5112ADCARG4ACTIVEHTSSOPDCA562000 Green (RoHS &
no Sb/Br)
TAS5112ADFDACTIVEHTSSOPDFD5635Green (RoHS &
no Sb/Br)
TAS5112ADFDRACTIVEHTSSOPDFD562000 Green (RoHS &
no Sb/Br)
TAS5112ADFDRG4ACTIVEHTSSOPDFD562000 Green (RoHS &
no Sb/Br)
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Lead/Ball Finish MSL Peak Temp
CU NIPDAULevel-3-260C-168 HR
CU NIPDAULevel-3-260C-168 HR
CU NIPDAULevel-3-260C-168 HR
CU NIPDAULevel-3-260C-168 HR
CU NIPDAULevel-3-260C-168 HR
CU NIPDAULevel-3-260C-168 HR
CU NIPDAULevel-3-260C-168 HR
(3)
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS) or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
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Addendum-Page 1
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