TEXAS INSTRUMENTS TAS5112 Technical data

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments

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FEATURES

D 50 W per Channel (BTL) Into 6 Ω (Stereo) D 95-dB Dynamic Range With TAS5026 D Less Than 0.1% THD+N (1 W RMS Into 6 Ω) D Less Than 0.2% THD+N (50 W RMS into 6 Ω) D Power Efficiency Typically 90% Into 6-Ω Load D Self-Protecting Design (Undervoltage,

Overtemperature and Short Conditions) With Error Reporting

D Internal Gate Drive Supply Voltage Regulator D EMI Compliant When Used With

Recommended System Design

SLES048C − JULY 2003 − REVISED MARCH 2004

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APPLICATIONS

D DVD Receiver D Home Theatre D Mini/Micro Component Systems D Internet Music Appliance

DESCRIPTION

The TAS5112 is a high-performance, integrated stereo digital amplifier power stage designed to drive 6-Ω speakers at up to 50 W per channel. The device incorporates TI’s PurePath Digitalt technology and is used with a digital audio PWM processor (TAS50XX) and a simple passive demodulation filter to deliver high-quality, high-efficiency, true-digital audio amplification.

The efficiency of this digital amplifier is typically 90%, reducing the size of both the power supplies and heatsinks needed. Overcurrent protection, overtemperature protection, and undervoltage protection are built into the TAS5112, safeguarding the device and speakers against fault conditions that could damage the system.

0.1

THD+N − Total Harmonic Distortion + Noise − %

0.01 100m

PurePath Digital and PowerPAD are trademarks of Texas Instruments. Other trademarks are the property of their respective owners.

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THD + NOISE vs OUTPUT POWER

RL = 6 Ω TC = 75°C

1 10 100

PO − Output Power − W

semiconductor products and disclaimers thereto appears at the end of this data sheet.

0.1

0.01

THD+N − Total Harmonic Distortion + Noise − %

0.001 20 100 1k 10k

THD + NOISE vs FREQUENCY

RL = 6 Ω TC = 75°C

PO = 50 W

PO = 10 W

PO = 1 W

f − Frequency − Hz

20k

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D D

C C

B B

A A

DFD PACKAGE

SLES048C − JULY 2003 − REVISED MARCH 2004

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates.

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GENERAL INFORMATION

Terminal Assignment

The TAS5112 is offered in a thermally enhanced 56-pin TSSOP DFD (thermal pad is on the top), shown as follows.

(TOP VIEW)

GND GVDD BST_D PVDD_ PVDD_ OUT_D OUT_D GND GND OUT_C OUT_C PVDD_ PVDD_ BST_C BST_B PVDD_ PVDD_ OUT_B OUT_B GND GND OUT_A OUT_A PVDD_ PVDD_ BST_A GVDD GND

GND GND

GREG

OTW

SD_CD

SD_AB

PWM_DP

PWM_DM

ESET_CD

PWM_CM

PWM_CP

REG_RTN

M3 M2 M1

DREG

PWM_BP

PWM_BM

RESET_AB

PWM_AM

PWM_AP

GND

DGND

GND

DVDD

GREG

GND GND

1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28

56 55 54 53 52 51 50 49 48 47 46 45 44 43 42 41 40 39 38 37 36 35 34 33 32 31 30 29

ABSOLUTE MAXIMUM RATINGS

over operating free-air temperature range unless otherwise noted

TAS5112

DVDD TO DGND –0.3 V to 4.2 V GVDD TO GND 33.5 V PVDD_X TO GND (dc voltage) 33.5 V PVDD_X TO GND (spike voltage OUT_X TO GND (dc voltage) 33.5 V OUT_X TO GND (spike voltage BST_X TO GND (dc voltage) 48 V BST_X TO GND (spike voltage GREG TO GND PWM_XP, RESET, M1, M2, M3, SD,

OTW Maximum operating junction

temperature, T Storage temperature –40°C to 125°C

(1)

Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied. Exposure to absolutemaximum-rated conditions for extended periods may affect device reliability.

(2)

The duration of voltage spike should be less than 100 ns.

(3)

GREG is treated as an input when the GREG pin is overdriven by GVDD of 12 V .

(3)

(2)

) 48 V

(2)

) 48 V

(2)

) 53 V

UNITS

14.2 V

–0.3 V to DVDD + 0.3 V

–40°C to 150°C

ORDERING INFORMATION

0°C to 70°C TAS5112DFD 56-pin small TSSOP

(1)

For the most current specification and package information, refer to our Web site at www.ti.com.

PACKAGE DESCRIPTION

PACKAGE DISSIPATION RATINGS

PACKAGE

56-pin DAD TSSOP 1.14 See Note 1

(1)

The TAS5112 package is thermally enhanced for conductive cooling using an exposed metal pad area. It is impractical to use the device with the pad exposed to ambient air as the only heat sinking of the device. For this reason, R thermal treatment, is provided in the Application Information section of the data sheet. An example and discussion of typical system R

values are provided in the Thermal Information section. This

θJA

example provides additional information regarding the power dissipation ratings. This example should be used as a reference to calculate the heat dissipation ratings for a specific application. TI application engineering provides technical support to design heatsinks if needed.

a system parameter that characterizes the

θJA,

θJC

(°C/W)

θJA

(°C/W)

(1)

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FUNCTION

(1)

DESCRIPTION

SLES048C − JULY 2003 − REVISED MARCH 2004

Terminal Functions

TERMINAL

NAME NO.

BST_A 31 P High-side bootstrap supply (BST), external capacitor to OUT_A required BST_B 42 P High-side bootstrap supply (BST), external capacitor to OUT_B required BST_C 43 P HS bootstrap supply (BST), external capacitor to OUT_C required BST_D 54 P HS bootstrap supply (BST), external capacitor to OUT_D required DGND 23 P Digital I/O reference ground DREG 16 P Digital supply voltage regulator decoupling pin, capacitor connected to GND DREG_RTN 12 P Digital supply voltage regulator decoupling return pin DVDD 25 P I/O reference supply input (3.3 V) GND 1, 2, 22, 24,

27, 28, 29, 36,

37, 48, 49, 56 GREG 3, 26 P Gate drive voltage regulator decoupling pin, capacitor to REG_GND GVDD 30, 55 P Voltage supply to on-chip gate drive and digital supply voltage regulators M1 (TST0) 15 I Mode selection pin M2 14 I Mode selection pin M3 13 I Mode selection pin OTW 4 O Overtemperature warning output, open drain with internal pullup resistor OUT_A 34, 35 O Output, half-bridge A OUT_B 38, 39 O Output, half-bridge B OUT_C 46, 47 O Output, half-bridge C OUT_D 50, 51 O Output, half-bridge D PVDD_A 32, 33 P Power supply input for half-bridge A PVDD_B 40, 41 P Power supply input for half-bridge B PVDD_C 44, 45 P Power supply input for half-bridge C PVDD_D 52, 53 P Power supply input for half-bridge D PWM_AM 20 I Input signal (negative), half-bridge A PWM_AP 21 I Input signal (positive), half-bridge A PWM_BM 18 I Input signal (negative), half-bridge B PWM_BP 17 I Input signal (positive), half-bridge B PWM_CM 10 I Input signal (negative), half-bridge C PWM_CP 11 I Input signal (positive), half-bridge C PWM_DM 8 I Input signal (negative), half-bridge D PWM_DP 7 I Input signal (positive), half-bridge D RESET_AB 19 I Reset signal, active low RESET_CD 9 I Reset signal, active low SD_AB 6 O Shutdown signal for half-bridges A and B, active-low SD_CD 5 O Shutdown signal for half-bridges C and D, active-low

(1)

I = input, O = Output, P = Power

P Power ground

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SLES048C − JULY 2003 − REVISED MARCH 2004

FUNCTIONAL BLOCK DIAGRAM

PWM_AP

RESET

PWM

Receiver

Protection A

Protection B

GREG

Timing

Control

GREG

Gate

Drive

Gate

Drive

Gate

Drive

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BST_A

PVDD_A

OUT_A

GND

BST_B

PVDD_B

PWM_BP

OTW

PWM

Receiver

To Protection

Blocks

Protection

UVP

This diagram shows one channel.

Control

GREG

Timing

DREG

Gate

Drive

GREG

DREG_RTN

OUT_B

GND

DREG

GVDD

GREG

DREG_RTN

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Output power

+ noise

SLES048C − JULY 2003 − REVISED MARCH 2004

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RECOMMENDED OPERATING CONDITIONS

MIN TYP MAX UNIT

DVDD Digital supply GVDD PVDD_x Half-bridge supply Relative to GND, RL= 6 Ω to 8 Ω 0 29.5 30.5 V

(1)

It is recommended for DVDD to be connected to DREG via a 100-Ω resistor.

Supply for internal gate drive and logic regulators

Junction temperature 0 125 _C

(1)

Relative to DGND 3 3.3 3.6 V Relative to GND 16 29.5 30.5 V

ELECTRICAL CHARACTERISTICS

PVDD_X = 29.5 V, GVDD = 29.5 V, DVDD connected to DREG via a 100-Ω resistor, RL = 6 Ω, 8X fs = 384 kHz, unless otherwise noted

TYPICAL OVER TEMPERATURE

SYMBOL PARAMETER TEST CONDITIONS

AC PERFORMANCE, BTL Mode, 1 kHz

RL = 8 Ω, THD = 0.2%, AES17 filter, 1 kHz

RL = 8 Ω, THD = 10%, AES17 filter, 1 kHz

RL = 6 Ω, THD = 0.2%, AES17 filter, 1 kHz

RL = 6 Ω, THD = 10%, AES17 filter, 1 kHz

Po = 1 W/ channel, RL = 6 Ω, AES17 filter

THD+N

SNR Signal-to-noise ratio A-weighted, AES17 filter 96 dB Typ DR Dynamic range

INTERNAL VOLTAGE REGULATOR

DREG Voltage regulator

GREG Voltage regulator

IVGDD

IDVDD

OUTPUT STAGE MOSFETs

on,LS

on,HS

Total harmonic distortion + noise

Output integrated voltage noise

GVDD supply current, operating

DVDD supply current, operating

Forward on-resistance, low side

Forward on-resistance, high side

Po = 10 W/channel, RL = 6 Ω, AES17 filter

Po = 50 W/channel, RL = 6 Ω, AES17 filter

A-weighted, mute, RL = 6 Ω,, 20 Hz to 20 kHz, AES17 filter

f = 1 kHz, A-weighted, AES17 filter

Io = 1 mA, PVDD = 18 V−30.5 V

Io = 1.2 mA, PVDD = 18 V−30.5 V

fS = 384 kHz, no load, 50% duty cycle

fS = 384 kHz, no load 1 5 mA Max

TJ = 25°C 155 mΩ Typ

TA=25°C TA=25°C

3.1 V Typ

13.4 V Typ

24 mA Max

Case

75°C

0.03% Typ

0.04% Typ

0.2% Typ

260 µV Max

TA=40°C

TO 85°C

40 W Typ

50 W Typ

62 W Typ

96 dB Typ

UNITS

MIN/TYP/

MAX



Undervoltage protection

Undervoltage protection GVDD up and down. Monitor

7.4

VIHHigh-level input voltage

Leakage

Input leakage current

SLES048C − JULY 2003 − REVISED MARCH 2004

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ELECTRICAL CHARACTERISTICS

PVDD_x = 29.5 V, GVDD = 29.5 V, DVDD connected to DREG via a 100-Ω resistor, RL = 6 Ω, 8X fs = 384 kHz, unless otherwise noted

TYPICAL OVER TEMPERATURE

SYMBOL PARAMETER TEST CONDITIONS

INPUT/OUTPUT PROTECTION

Set the DUT in normal operation mode with all the protections enabled. Sweep

uvp,G

OTW

OTE OC Overcurrent protection See Note 1. 5.8 A Typ

STATIC DIGITAL SPECIFICATION

OTW/SHUTDOWN (SD)

(1)

To optimize device performance and prevent overcurrent (OC) protection tripping, the demodulation filter must be designed with special care. See Demodulation Filter Design in the Application Information section of the data sheet and consider the recommended inductors and capacitors for optimal performance. It is also important to consider PCB design and layout for optimum performance of the TAS5112. It is recommended to follow the TAS5112F2EVM (S/N 1 12) design and layout guidelines for best performance.

limit, GVDD

Overtemperature warning, junction temperature

Overtemperature error, junction temperature

PWM_AP, PWM_BP, M1, M2, M3, SD, OTW

Low-level input voltage 0.8 V Max

Internally pull up R from OTW/SD to DVDD

Low-level output voltage IO = 4 mA 0.4 V Max

SD output. Record the GREG reading when SD is triggered.

TA=25°C TA=25°C

6.9 V Min

7.9 V Max

125 °C Typ

150 °C Typ

DVDD V Max

−10 µA Min 10 µA Max

30 22.5 kΩ Min

Case

75°C

2 V Min

TA=40°C

TO 85°C

UNITS

MIN/TYP/

MAX

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Gate-Drive

SYSTEM CONFIGURATION USED FOR CHARACTERIZATION

Power Supply

External Power Supply

H-Bridge

Power Supply

TAS5112DFD

ERR_RCVY PWM_AP_1

PWM_AM_1

VALID_1

PWM PROCESSOR

TAS5026

PWM_AP_2

PWM_AM_2

VALID_2

1 µF

100 nF

100 Ω

100 nF

1 µF

GND

GREG

OTW

SD_CD

SD_AB

PWM_DP

PWM_DM

RESET_CD

PWM_CM

PWM_CP

DREG_RTN

DREG

PWM_BP

PWM_BM

RESET_AB

PWM_AM

PWM_AP

GND

DGND

GND

DVDD

GREG

GND

BST_D PVDD_D PVDD_D

OUT_D OUT_D

OUT_C

OUT_C PVDD_C PVDD_C

BST_C

BST_B PVDD_B PVDD_B

OUT_B OUT_B

OUT_A

OUT_A PVDD_A PVDD_A

BST_A

GND

GVDD

GND GND

GVDD

GND

56 55

1.5 Ω

54 53 52 51 50 49 48 47 46 45 44 43

1.5 Ω

42 41 40 39 38 37 36 35 34 33 32 31

1.5 Ω

30 29

33 nF

100 nF

1.5 Ω

33 nF

100 nF

1.5 Ω

33 nF

SLES048C − JULY 2003 − REVISED MARCH 2004

100 nF

PCB

10 µH

{

470 nF

10 µH

{

100 nF

PCB

10 µH

{

470 nF

10 µH

{

100 nF

PCB

100 nF

4.7 kΩ

1000 µF



: TRACK IN THE PCB (1,0 mm wide and 50 mm long)

PCB

{

Voltage Suppressor Diode: 1SMA33CAT

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THD+N − Total Harmonic Distortion + Noise − %

SLES048C − JULY 2003 − REVISED MARCH 2004

TYPICAL CHARACTERISTICS AND SYSTEM PERFORMANCE

OF TAS5112 EVM WITH TAS5026 PWM PROCESSOR

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TOTAL HARMONIC DISTORTION + NOISE

FREQUENCY

RL = 6 Ω TC = 75°C

PO = 50 W

0.1

PO = 10 W

PO = 1 W

0.01

0.001 20 100 1k 10k

f − Frequency − Hz

Figure 1

20k

NOISE AMPLITUDE

FREQUENCY

RL = 6 Ω FFT = −60 dB

−20 TC = 75°C

TAS5026 Front End Device

−40

−60

−80

−100

Noise Amplitude − dBr

−120

−140

−160 0 2 4 6 8 10121416182022

f − Frequency − kHz

Figure 2

TOTAL HARMONIC DISTORTION + NOISE

RL = 6 Ω TC = 75°C

0.1

0.01 100m

PO − Output Power − W

OUTPUT POWER

1 10 100

Figure 3

OUTPUT POWER

H-BRIDGE VOLTAGE

TA = 75°C

RL = 6 Ω

− Output Power − W

0 4 8 12 16 20 24 28 32

VDD − Supply Voltage − V

Figure 4

RL = 8 Ω

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− System Output Stage Efficiency − %

− Output Power − W



SLES048C − JULY 2003 − REVISED MARCH 2004

SYSTEM OUTPUT STAGE EFFICIENCY

OUTPUT POWER

100

80 70

60 50

40 30

0 5 10 15 20 25 30 35 40 45 50 55 60 65

PO − Output Power − W

Figure 5

f = 1 kHz RL = 6 Ω TC = 75°C

POWER LOSS

OUTPUT POWER

f = 1 kHz

RL = 6 Ω

TC = 75°C 8 7 6 5

− Power Loss − W

tot

3 2 1 0

0 5 10 15 20 25 30 35 40 45 50 55 60 65

PO − Output Power − W

Figure 6

OUTPUT POWER

CASE TEMPERATURE

PVDD = 29.5 V

RL = 6 Ω

56 54

52 50

48 46

44 42 40

0 20 40 60 80 100 120 140

TC − Case Temperature − °C

Channel 2

Channel 1

Figure 7

AMPLITUDE

FREQUENCY

3.0

2.5

2.0

1.5

1.0

0.5

0.0

−0.5

Amplitude − dBr

−1.0

−1.5

−2.0

−2.5

−3.0 10 100 1k 50

f − Frequency − Hz

RL = 6 Ω

Figure 8

RL = 8 Ω

10k



SLES048C − JULY 2003 − REVISED MARCH 2004

200

190

180

170

160

150

− On-State Resistance − mΩ

140

130

120

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ON-STATE RESISTANCE

JUNCTION TEMPERATURE

0 102030405060708090100

TJ − Junction Temperature − °C

Figure 9

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SLES048C − JULY 2003 − REVISED MARCH 2004

THEORY OF OPERATION

POWER SUPPLIES

The power device only requires two supply voltages, GVDD and PVDD_X.

GVDD is the gate drive supply for the device, regulated internally down to approximately 12 V, and decoupled with regards to board GND on the GREG pins through an external capacitor. GREG powers both the low side and high side via a bootstrap step-up conversion. The bootstrap supply is charged after the first low-side turn-on pulse. Internal digital core voltage DREG is also derived from GVDD and regulated down by internal circuitry to

3.3 V. The gate-driver regulator can be bypassed for reducing

idle loss in the device by shorting GREG to GVDD and directly feeding in 12.0 V. This can be useful in an application where thermal conduction of heat from the device is difficult.

PVDD_X is the H-bridge power supply pin. T wo power pins exists for each half-bridge to handle the current density . It is important that the circuitry recommendations around the PVDD_X pins are followed carefully both topologyand layout-wise. For topology recommendations, see the Typical System Configuration section. Following these recommendations is important for parameters like EMI, reliability, and performance.

POWERING UP

This precharges the bootstrap supply capacitors and discharges the output filter capacitor (see the Typical TAS5112 Application Configuration section).

After GVDD has been applied, it takes approximately 800 µs to fully charge the BST capacitor. Within this time, RESET must be kept low. After approximately 1 ms, the back-end bootstrap capacitor is charged.

RESET can now be released if the modulator is powered up and streaming valid PWM signals to the back-end PWM_xP. Valid means a switching PWM signal which complies with the frequency and duty cycle ranges stated in the Recommended Operating Conditions.

A constant HIGH dc level on the PWM_xP is not permitted, because it would force the high-side MOSFET ON until it eventually ran out of BST capacitor energy and might damage the device.

An unknown state of the PWM output signals from the modulator is illegal and should be avoided, which in practice means that the PWM processor must be powered up and initialized before RESET is de-asserted HIGH to the back end.

POWERING DOWN

For power down of the back end, an opposite approach is necessary. The RESET must be asserted LOW before the valid PWM signal is removed.

When PWM processors are used with TI PurePath Digital amplifiers, the correct timing control of RESET and PWM_xP is performed by the modulator.

> 1 ms

RESET

GVDD

PVDD_x

PWM_xP

NOTE: PVDD should not be powered up before GVDD.

During power up when RESET is asserted LOW, all MOSFETs are turned off and the two internal half-bridges are in the high-impedance state (Hi-Z). The bootstrap capacitors supplying high-side gate drive are not charged at this point. T o comply with the click and pop scheme and use of non-TI modulators it is recommended to use a 4-kΩ pulldown resistor on each PWM output node to ground.

PRECAUTION

The TAS5112 must always start up in the high-impedance (Hi-Z) state. In this state, the bootstrap (BST) capacitor is precharged by a resistor on each PWM output node to ground. See the system configuration. This ensures that the back end is ready for receiving PWM pulses, indicating either HIGH- or LOW-side turnon after RESET is de-asserted to the back end.

With the following pulldown resistor and BST capacitor size, the charge time is:

C = 33 nF, R = 4.7 kΩ R × C × 5 = 775.5 µs

After GVDD has been applied, it takes approximately 800 µs to fully charge the BST capacitor. During this time, RESET must be kept low. After approximately 1 ms the back end BST is charged and ready. RESET can now be released if the PWM modulator is ready and is streaming valid PWM signals to the back end. Valid PWM signals are switching PWM signals with a frequency between 350−400 kHz. A constant HIGH level on the PWM+ would force the high-side MOSFET ON until it eventually ran out of BST capacitor energy. Putting the device in this condition should be avoided.

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SLES048C − JULY 2003 − REVISED MARCH 2004

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In practice this means that the DVDD-to-PWM processor (front-end) should be stable and initialization should be completed before RESET is de-asserted to the back end.

CONTROL I/O

Shutdown Pin: SD

The SD pin functions as an output pin and is intended for protection-mode signaling to, for example, a controller or other front-end device. The pin is open-drain with an internal pullup resistor to DVDD.

The logic output is, as shown in the following table, a combination of the device state and RESET input:

SD RESET DESCRIPTION

0 0 Not used 0 1 Device in protection mode, i.e., UVP and/or OC

and/or OT error

(1)

1 1 Normal operation

(1)

SD is pulled high when RESET is asserted low independent of chip state (i.e., protection mode). This is desirable to maintain compatibility with some TI PWM front ends.

Temperature Warning Pin: OTW

The OTW pin gives a temperature warning signal when temperature exceeds the set limit. The pin is of the open-drain type with an internal pullup resistor to DVDD.

Overall Reporting

The SD pin, together with the OTW pin, gives chip state information as described in Table 1.

0 Device set high-impedance (Hi-Z), SD forced high

OTW DESCRIPTION

0 Junction temperature higher than 125°C 1 Junction temperature lower than 125°C

The device can be recovered by toggling RESET low and then high, after all errors are cleared.

Overcurrent (OC) Protection

The device has individual forward current protection on both high-side and low-side power stage FETs. The OC protection works only with the demodulation filter present at the output. See Demodulation Filter Design in the Application Information section of the data sheet for design constraints.

Overtemperature (OT) Protection

A dual temperature protection system asserts a warning signal when the device junction temperature exceeds 125°C. The OT protection circuit is shared by all half-bridges.

Undervoltage (UV) Protection

Undervoltage lockout occurs when GVDD is insufficient for proper device operation. The UV protection system protects the device under power-up and power-down situations. The UV protection circuits are shared by all half-bridges.

Reset Function

The function of the reset input is twofold:

D Reset is used for re-enabling operation after a

latching error event.

D Reset is used for disabling output stage

switching (mute function).

The error latch is cleared on the falling edge of reset and normal operation is resumed when reset goes high.

Table 1. Error Signal Decoding

OTW SD DESCRIPTION

0 0 Overtemperature error (OTE) 0 1 Overtemperature warning (OTW) 1 0 Overcurrent (OC) or undervoltage (UVP) error 1 1 Normal operation, no errors/warnings

Chip Protection

The TAS5112 protection function is implemented in a closed loop with, for example, a system controller and TI PWM processor. The TAS5112 contains three individual systems protecting the device against error conditions. All of the error events covered result in the output stage being set in a high-impedance state (Hi-Z) for maximum protection of the device and connected equipment.

PROTECTION MODE

Autorecovery (AR) After Errors (PMODE0)

In autorecovery mode (PMODE0) the TAS5112 is self-supported i n handling of error situations. All protection systems are active, setting the output stage in the high-impedance state to protect the output stage and connected equipment. However, after a short time the device autorecovers, i.e., operation is automatically resumed provided that the system is fully operational.

The autorecovery timing is set by counting PWM input cycles, i.e., the timing is relative to the switching frequency.

The AR system is common to both half-bridges.

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Timing and Function

The function of the autorecovery circuit is as follows:

1. An error event occurs and sets the protection latch (output stage goes Hi-Z).

2. The counter is started.

3. After n/2 cycles, the protection latch is cleared but the output stage remains Hi-Z (identical to pulling RESET

low).

4. After n cycles, operation is resumed (identical to pulling RESET high) (n = 512).

Error

Protection

Latch

Shutdown

Autorecovery

PWM

Counter

Table 3. Output Mode Selection

M3 OUTPUT MODE

0 Bridge-tied load output stage (BTL) 1 Reserved

APPLICATION INFORMATION

DEMODULATION FILTER DESIGN

The PurePath Digital amplifier outputs are driven by heavy-duty DMOS transistors in an H-bridge configuration. These transistors are either off or fully on, which reduces the DMOS transistor on-state resistance, R(DMOSon), and the power dissipated in the device, thereby increasing efficiency.

The result is a square-wave output signal with a duty cycle that is proportional to the amplitude of the audio signal. It is recommended that a second-order LC filter be used to recover the audio signal. For this application, EMI is considered important; therefore, the selected filter is the full-output type shown in Figure 11.

TAS51xx

Output A

R-RESET

Figure 10. Autorecovery Function

Latching Shutdown on All Errors (PMODE1)

In latching shutdown mode, all error situations result in a power down (output stage Hi-Z). Re-enabling can be done by toggling the RESET pin.

All Protection Systems Disabled (PMODE2)

In PMODE2, all protection systems are disabled. This mode is purely intended for testing and characterization purposes and thus not recommended for normal device operation.

MODE Pins Selection

The protection mode is selected by shorting M1/M2 to DREG or DGND according to Table 2.

Table 2. Protection Mode Selection

M1 M2 PROTECTION MODE

0 0 Autorecovery after errors (PMODE 0) 0 1 Latching shutdown on all errors (PMODE 1) 1 0 All protection systems disabled (PMODE 2) 1 1 Reserved

The output configuration mode is selected by shorting the M3 pin to DREG or DGND according to Table 3.

(Load)

Output B

C1A

C1B

Figure 11. Demodulation Filter

The main purpose of the output filter is to attenuate the high-frequency switching component of the PurePath Digital amplifier while preserving the signals in the audio band.

Design of the demodulation filter affects the performance of the power amplifier significantly. As a result, to ensure proper operation of the overcurrent (OC) protection circuit and meet the device THD+N specifications, the selection of the inductors used in the output filter must be considered according to the following. The rule is that the inductance should remain stable within the range of peak current seen at maximum output power and deliver at least 5 µH of inductance at 15 A.

If this rule is observed, the TAS5112 does not have distortion issues due to the output inductors and overcurrent conditions do not occur due to inductor saturation in the output filter.

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Another par a meter to be considered is the idle current loss in the inductor. This can be measured or specified as inductor dissipation (D). The target specification for dissipation is less than 0.05.

In general, 10-µH inductors suffice for most applications. The frequency response of the amplifier is slightly altered by the change in output load resistance; however, unless tight control of frequency response is necessary (better than 0.5 dB), it is not necessary to deviate from 10 µH.

The graphs in Figure 12 display the inductance vs current characteristics of two inductors that are recommended for use with the TAS5112.

INDUCTANCE

CURRENT

DASL983XX−1023

DFB1310A

THERMAL INFORMATION

The thermally augmented package provided with the TAS5112 is designed to be interfaced directly to heatsinks using a thermal interface compound (for example, Wakefield Engineering type 126 thermal grease.) The heatsink then absorbs heat from the ICs and couples it to the local air. If the heatsink is carefully designed, this process can reach equilibrium and heat can be continually removed from the ICs. Because of the efficiency of the TAS5112, heatsinks can be smaller than those required for linear amplifiers of equivalent performance.

is a system thermal resistance from junction to

ambient ai r. As such, it is a system parameter with roughly the following components:

D R

(the thermal resistance from junction to

case, or in this case the metal pad)

D Thermal grease thermal resistance D Heatsink thermal resistance

has been provided in the General Information

section. The thermal grease thermal resistance can be calculated

from the exposed pad area and the thermal grease manufacturer’s area thermal resistance (expressed in °C-in2/W). The area thermal resistance of the example thermal grease with a 0.002-inch thick layer is about 0.1 °C-in2/W. The approximate exposed pad area is as follows:

56-pin HTSSOP 0.045 in

Dividing the example thermal grease area resistance by the surface area gives the actual resistance through the thermal grease for both ICs inside the package:

The selection of the capacitor that is placed across the output of each inductor (C2 in Figure 11) is simple. To complete the output filter, use a 0.47-µF capacitor with a voltage rating at least twice the voltage applied to the output stage (PVDD).

This capacitor should be a good quality polyester dielectric such as a Wima MKS2-047ufd/100/10 or equivalent.

In order to minimize the EMI effect of unbalanced ripple loss in the inductors, 0.1-µF 50-V SMD capacitors (X7R or better) (C1A and C1B in Figure 11) should be added from the output of each inductor to ground.

0 5 10 15

I − Current − A

Figure 12. Inductance Saturation

56-pin HTSSOP 2.27 °C/W

The thermal resistance of thermal pads is generally considerably higher than a thin thermal grease layer. Thermal tape has an even higher thermal resistance. Neither pads nor tape should be used with either of these two packages. A thin layer of thermal grease with careful clamping of the heatsink is recommended. It may be difficult to achieve a layer 0.001-inch thick or less, so the modeling below is done with a 0.002-inch thick layer, which may be more representative of production thermal grease thickness.

Heatsink thermal resistance is generally predicted by the heatsink vendor, modeled using a continuous flow dynamics (CFD) model, or measured.

Thus, for a single monaural IC, the system R

= R

thermal-grease resistance + heatsink resistance. Table 4, Table 5, and Table 6 indicate modeled

parameters for one or two TAS5112 ICs on a single heatsink. The final junction temperature is set at 110°C in

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all cases. It is assumed that the thermal grease is 0.002 inch thick and that it is similar in performance to Wakefield Type 126 thermal grease. It is important that the thermal grease layer is ≤0.002 inches thick and that thermal pads or tape are not used in the pad-to-heatsink interface due to the high power density that results in these extreme power cases.

Table 4. Case 1 (2 × 50 W Unclipped Into 6 Ω,

Both Channels in Same IC)

Ambient temperature 25°C Power to load (per channel) 50 W (unclipped) Power dissipation 4.5 W

Delta T inside package

Delta T through thermal grease Required heatsink thermal resistance 4.2°C/W

Junction temperature 110°C System R R Junction temperature 85°C + 25°C = 110°C

(1)

θJA

* power dissipation 85°C

θJA

This case represents a stereo system with only one package. See Case 2 and Case 2A if doing a full-power, 2-channel test in a multichannel system.

(1)

56-Pin HTSSOP

10.2°C, note 2 × channel dissipation

37.1°C, note 2 × channel dissipation

19°C/W

Table 6. Case 2A (2 × 60 W Unclipped Into 6 Ω,

Channels in Separate IC Packages)

56-Pin HTSSOP

Ambient temperature 25°C Power to load (per channel) 60 W (10% THD) Power dissipation per channel 5.4 W

Delta T inside package

Delta T through thermal grease Required heatsink thermal resistance 5.3°C/W

Junction temperature 110°C System R R Junction temperature 85°C + 25°C = 110°C

(1)

θJA

* power dissipation 85°C

θJA

In this case, the power is also separated into two packages, but overdriving causes clipping to 10% THD. In this case, the high power requires extreme care in attachment of the heatsink to ensure that the thermal grease layer is ≤ 0.002 inches thick. Note that this power level should not be attempted with both channels in a single IC because of the high power density through the thermal grease layer.

6.1°C, note 2 × channel dissipation

22.3°C, note 2 × channel dissipation

15.9°C/W

(1)

Table 5. Case 2 (2 × 50 W Unclipped Into 6 Ω,

Channels in Separate Packages)

Ambient temperature 25°C Power to load (per channel) 50 W (unclipped) Power dissipation 4.5 W Delta T inside package 5.1°C Delta T through thermal grease 18.6°C Required heatsink thermal resistance 6.9°C/W Junction temperature 110°C System R R Junction temperature 85°C + 25°C = 110°C

(1)

θJA

* power dissipation 85°C

θJA

In this case, the power is separated into two packages. Note that this allows a considerably smaller heatsink because twice as much area is available for heat transfer through the thermal grease. For this reason, separating the stereo channels into two ICs is recommended in full-power stereo tests made on multichannel systems.

19°C/W

(1)

56-Pin HTSSOP

Thermal

Pad

3,90 mm 2,98 mm

8,20 mm 7,20 mm

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CLICK AND POP REDUCTION

TI modulators feature a pop and click reduction system that controls the timing when switching starts and stops.

Going from nonswitching to switching operation causes a spectral energy burst to occur within the audio bandwidth, which is heard in the speaker as an audible click, for instance, after having asserted RESET LH during a system start-up.

To make this system work properly, the following design rules must be followed when using the TAS5112 back end:

D The relative timing between the PWM_AP/M_x

signals and their corresponding VALID_x signal should not be skewed by inserting delays, because this increases the audible amplitude level of the click.

D The output stage must start switching from a

fully discharged output filter capacitor. Because the output stage prior to operation is in the high-impedance state, this is done by having a passive pulldown resistor on each speaker output to GND (see Typical System Configuration).

Other things that can affect the audible click level:

D The spectrum of the click seems to follow the

speaker impedance vs. frequency curve—the higher the impedance, the higher the click energy.

D Crossover filters used between woofer and

tweeter in a speaker can have high impedance in the audio band, which should be avoided if possible.

Another way to look at it is that the speaker impulse response is a major contributor to how the click energy is shaped in the audio band and how audible the click will be.

The following mode transitions feature click and pop reduction.

REFERENCES

1. TAS5000 Digital Audio PWM Processor data manual—TI (SLAS270)

2. True Digital Audio Amplifier TAS5001 Digital Audio PWM Processor data sheet—TI (SLES009)

3. True Digital Audio Amplifier TAS5010 Digital Audio PWM Processor data sheet—TI (SLAS328)

4. True Digital Audio Amplifier TAS5012 Digital Audio PWM Processor data sheet—TI (SLES006)

5. TAS5026 Six-Channel Digital Audio PWM Processor data manual—TI (SLES041)

6. TAS5036A Six-Channel Digital Audio PWM Processor data manual—TI (SLES061)

7. TAS3103 Digital Audio Processor With 3D Effects data manual—TI (SLES038)

8. Digital Audio Measurements application report—TI (SLAA114)

9. PowerPAD Thermally Enhanced Package technical brief—TI (SLMA002)

10. System Design Considerations for True Digital Audio Power Amplifiers application report—TI (SLAA117)

(1)

Normal Mute → Normal

(1)

Normal Error recovery → Normal

(1)

Normal Hard Reset → Normal

(1)

Normal = switching

STATE

→ Mute Yes

(1)

Error recovery

→

(ERRCVY)

(1)

→ Hard Reset No

(1)

CLICK AND

POP REDUCED

Yes Yes Yes

Yes

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MECHANICAL DATA

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TEXAS INSTRUMENTS TAS5112 Technical data

Specifications and Main Features

Frequently Asked Questions

User Manual