Texas Instrument LM3886 User Manual

Download

Page 1

LM3886

www.ti.com

SNAS091C –MAY 1999–REVISED MARCH 2013

LM3886 Overture™ Audio Power Amplifier Series

High-Performance 68W Audio Power Amplifier w/Mute

Check for Samples: LM3886

FEATURES

• 68W Cont. Avg. Output Power into 4Ω at VCC= ±28V

• 38W Cont. Avg. Output Power into 8Ω at VCC= ±28V

• 50W Cont. Avg. Output Power into 8Ω at VCC= ±35V

• 135W Instantaneous Peak Output Power Capability dynamically protected Safe Operating Area (SOA).

• Signal-to-Noise Ratio ≥ 92dB

• An Input Mute Function

• Output Protection from a Short to Ground or to the Supplies via Internal Current Limiting Circuitry

• Output Over-Voltage Protection against Transients from Inductive Loads

• Supply Under-Voltage Protection, not Allowing Internal Biasing to Occur when |VEE| + |VCC| ≤ 12V, thus Eliminating Turn-On and Turn-Off Transients

• 11-Lead TO-220 Package

• Wide Supply Range 20V - 94V

DESCRIPTION

The LM3886 is a high-performance audio power amplifier capable of delivering 68W of continuous average power to a 4Ω load and 38W into 8Ω with

0.1% THD+N from 20Hz–20kHz. The performance of the LM3886, utilizing its Self

Peak Instantaneous Temperature (°Ke) (SPiKe) protection circuitry, puts it in a class above discrete and hybrid amplifiers by providing an inherently,

SPiKe protection means that these parts are completely safeguarded at the output against overvoltage, undervoltage, overloads, including shorts to the supplies, thermal runaway, and instantaneous temperature peaks.

The LM3886 maintains an excellent signal-to-noise ratio of greater than 92dB with a typical low noise floor of 2.0µV. It exhibits extremely low THD+N values of 0.03% at the rated output into the rated load over the audio spectrum, and provides excellent linearity with an IMD (SMPTE) typical rating of

0.004%.

APPLICATIONS

• Component stereo

• Compact stereo

• Self-powered speakers

• Surround-sound amplifiers

• High-end stereo TVs

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

2Overture is a trademark of dcl_owner. 3All other trademarks are the property of their respective owners.

PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters.

Page 2

LM3886

SNAS091C –MAY 1999–REVISED MARCH 2013

Typical Application

www.ti.com

*Optional components dependent upon specific design requirements. Refer to External Components Description for a component functional description.

Figure 1. Typical Audio Amplifier Application Circuit

Connection Diagram

Top View

Preliminary: call your local Texas Instruments Sales Rep. or distributor for availability

Figure 2. TO-220 and PFM Plastic Packages

See Package Number NDJ0011B for Staggered Lead Non-Isolated Package or NDA0011B for Staggered

Lead Isolated Package

(1)

(1) The LM3886T package TA11B is a non-isolated package, setting the tab of the device and the heat sink at V−potential when the

LM3886 is directly mounted to the heat sink using only thermal compound. If a mica washer is used in addition to thermal compound,

(case to sink) is increased, but the heat sink will be isolated from V−.

Product Folder Links: LM3886

Page 3

LM3886

www.ti.com

SNAS091C –MAY 1999–REVISED MARCH 2013

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates.

Absolute Maximum Ratings

(1)(2)(3)

Supply Voltage |V+|+|V−| (No Signal) 94V Supply Voltage |V+|+|V−| (Input Signal) 84V Common Mode Input Voltage (V+or V−) and

|V+| + |V−| ≤ 80V

Differential Input Voltage

(4)

60V Output Current Internally Limited Power Dissipation ESD Susceptibility Junction Temperature

(5)

(6)

(7)

125W

3000V

150°C Soldering Information T Package (10 seconds) 260°C Storage Temperature −40°C to +150°C Thermal Resistance θ

1°C/W

43°C/W

(1) All voltages are measured with respect to the GND pin (pin 7), unless otherwise specified. (2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for

which the device is functional. Electrical Characteristics state DC and AC electrical specifications under particular test conditions and specific performance limits. This assumes that the device is within the Operating Ratings. The typical value is a good indication of device performance.

(3) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and

specifications. (4) The Differential Input Voltage Absolute Maximum Rating is based on supply voltages of V+= +40V and V−= −40V. (5) For operating at case temperatures above 25°C, the device must be derated based on a 150°C maximum junction temperature and a

thermal resistance of θJC= 1.0 °C/W (junction to case). Refer to Figure 50 in Application Information under THERMAL

CONSIDERATIONS.

(6) Human body model, 100 pF discharged through a 1.5 kΩ resistor. (7) The operating junction temperature maximum is 150°C, however, the instantaneous Safe Operating Area temperature is 250°C.

Operating Ratings

(1)(2)(3)

Temperature Range

MIN

≤ TA≤ T

MAX

−20°C ≤ TA≤ +85°C

Supply Voltage |V+| + |V−| 20V to 84V

which the device is functional. Electrical Characteristics state DC and AC electrical specifications under particular test conditions and

specific performance limits. This assumes that the device is within the Operating Ratings. The typical value is a good indication of

device performance. (3) Operation is specified up to 84V, however, distortion may be introduced from SPIKe Protection Circuitry if proper thermal considerations

are not taken into account. Refer to THERMAL CONSIDERATIONS for more information. (See Figure 5)

Product Folder Links: LM3886

Page 4

LM3886

SNAS091C –MAY 1999–REVISED MARCH 2013

Electrical Characteristics

(1)(2)

The following specifications apply for V+= +28V, V−= −28V, I

= −0.5 mA with RL= 4Ω unless otherwise specified. Limits

MUTE

www.ti.com

apply for TA= 25°C.

Parameter Test Conditions

|V+| + |V−| Power Supply Voltage

Mute Attenuation Pin 8 Open or at 0V, Mute: On

(5)

− V−≥ 9V 20 V (min)

pin7

Current out of Pin 8 > 0.5 mA, 115 80 dB (min) Mute: Off

(6)

Output Power (Continuous Average) THD + N = 0.1% (max)

f = 1 kHz; f = 20 kHz |V+| = |V−| = 28V, RL= 4Ω 68 60 W (min) |V+| = |V−| = 28V, RL= 8Ω 38 30 W (min) |V+| = |V−| = 35V, RL= 8Ω 50 W

Peak P

Instantaneous Peak Output Power 135 W

THD + N Total Harmonic Distortion Plus Noise 60W, RL= 4Ω, %

30W, RL= 8Ω, 0.03 % 20 Hz ≤ f ≤ 20 kHz 0.03 AV= 26 dB

(6)

+(6)

(8)

PSRR

Slew Rate Total Quiescent Power Supply Current VCM= 0V, Vo= 0V, Io= 0A 50 85 mA (max) Input Offset Voltage VCM= 0V, Io= 0 mA 1 10 mV (max) Input Bias Current VCM= 0V, Io= 0 mA 0.2 1 μA (max) Input Offset Current VCM= 0V, Io= 0 mA 0.01 0.2 μA (max) Output Current Limit |V+| = |V−| = 20V, tON= 10 ms, VO= 0V 11.5 7 A (min) Output Dropout Voltage

(8)

Power Supply Rejection Ratio V+= 40V to 20V, V−= −40V, 120 85 dB (min)

(7)

(9)

VIN= 2.0Vp-p, t

= 2 ns 19 8 V/μs (min)

RISE

|V+–VO|, V+= 28V, Io= +100 mA 1.6 2.0 V (max) |VO–V−|, V−= −28V, Io= −100 mA 2.5 3.0 V (max)

VCM= 0V, Io= 0 mA V+= 40V, V−= −40V to −20V, 105 85 dB (min) VCM= 0V, Io= 0 mA

(8)

CMRR

Common Mode Rejection Ratio V+= 60V to 20V, V−= −20V to −60V,

VCM= 20V to −20V, Io= 0 mA

(8)

VOL

Open Loop Voltage Gain |V+| = |V−| = 28V, RL= 2 kΩ, ΔVO= 40V 115 90 dB (min)

GBWP Gain-Bandwidth Product |V+| = |V−| = 30V

fO= 100 kHz, VIN= 50 mVrms

(6)

Input Noise IHF—A Weighting Filter

RIN= 600Ω (Input Referred)

SNR Signal-to-Noise Ratio PO= 1W, A-Weighted,

Measured at 1 kHz, RS= 25Ω PO= 60W, A-Weighted,

Measured at 1 kHz, RS= 25Ω

IMD Intermodulation Distortion Test 60 Hz, 7 kHz, 4:1 (SMPTE) 0.004

60 Hz, 7 kHz, 1:1 (SMPTE) 0.009

LM3886

Typical

(3)

(4)

Limit

84 V (max)

110 85 dB (min)

8 2 MHz (min)

2.0 10 μV (max)

92.5 dB

110 dB

(Limits)

Units

which the device is functional. Electrical Characteristics state DC and AC electrical specifications under particular test conditions and

specific performance limits. This assumes that the device is within the Operating Ratings. The typical value is a good indication of

device performance. (3) Typicals are measured at 25°C and represent the parametric norm. (4) Limits are speficied to AOQL (Average Outgoing Quality Level). (5) V−must have at least −9V at its pin with reference to ground in order for the under-voltage protection circuitry to be disabled. (6) AC Electrical Test; refer to Test Circuit #2 -AC Electrical Test Circuit. (7) The feedback compensation network limits the bandwidth of the closed-loop response and so the slew rate will be reduced due to the

high frequency roll-off. Without feedback compensation, the slew rate is typically larger. (8) DC Electrical Test; refer to Test Circuit #1- DC Electrical Test Circuit. (9) The output dropout voltage is the supply voltage minus the clipping voltage. Refer to Figure 14 in Typical Performance Characteristics.

Product Folder Links: LM3886

Page 5

LM3886

www.ti.com

Test Circuit #1- DC Electrical Test Circuit

Test Circuit #2 -AC Electrical Test Circuit

SNAS091C –MAY 1999–REVISED MARCH 2013

Product Folder Links: LM3886

Page 6

LM3886

SNAS091C –MAY 1999–REVISED MARCH 2013

Single Supply Application Circuit

www.ti.com

*Optional components dependent upon specific design requirements. Refer to the External Components Description

for a component functional description.

Figure 3. Typical Single Supply Audio Amplifier Application Circuit

Product Folder Links: LM3886

Page 7

LM3886

www.ti.com

Equivalent Schematic

(excluding active protection circuitry)

SNAS091C –MAY 1999–REVISED MARCH 2013

Product Folder Links: LM3886

Page 8

LM3886

SNAS091C –MAY 1999–REVISED MARCH 2013

(excluding active protection circuitry)

External Components Description

(See Figure 1 and Figure 3)

Components Functional Description

1. R

2. R

3. C

IN A A

4. C Provides AC coupling at the input and output of the amplifier for single supply operation.

5. R

6. C

7. Ri Inverting input resistance to provide AC Gain in conjunction with Rf1.

8. Ci

9. R

10. R

11. C

12. R

13. C

14. R

15. C

16. L

17. R

18. C

f1 f2

f M

M SN SN

(1)

19. S1 Mute switch that mutes the music going into the amplifier when opened.

(1) Optional components dependent upon specific design requirements. Refer to Application Information for more information.

Acts as a volume control by setting the voltage level allowed to the amplifier's input terminals. Provides DC voltage biasing for the single supply operation and bias current for the positive input terminal. Provides bias filtering.

Prevents currents from entering the amplifier's non-inverting input which may be passed through to the load upon powerdown of the system due to the low input impedance of the circuitry when the under-voltage circuitry is off. This phenomenon occurs when the supply voltages are below 1.5V.

(1)

Reduces the gain (bandwidth of the amplifier) at high frequencies to avoid quasi-saturation oscillations of the output transistor. The capacitor also suppresses external electromagnetic switching noise created from fluorescent lamps.

(1)

Feedback capacitor. Ensures unity gain at DC. Also a low frequency pole (highpass roll-off) at: fc= 1/(2πRi Ci) Feedback resistance to provide AC Gain in conjunction with Ri.

(1)

At higher frequencies feedback resistance works with Cfto provide lower AC Gain in conjunction with Rf1and Ri. A high frequency pole (lowpass roll-off) exists at:

fc= [Rf1Rf2(s + 1/Rf2Cf)]/[(Rf1+ Rf2)(s + 1/Cf(Rf1+ Rf2))] Compensation capacitor that works with Rf1and Rf2to reduce the AC Gain at higher frequencies. Mute resistance set up to allow 0.5 mA to be drawn from pin 8 to turn the muting function off.

→RMis calculated using: RM≤ (|VEE| − 2.6V)/I8 where I8 ≥ 0.5 mA. Refer to the Figure 44 and Figure 45 in Typical

Performance Characteristics.

Mute capacitance set up to create a large time constant for turn-on and turn-off muting.

(1)

Works with CSNto stabilize the output stage by creating a pole that eliminates high frequency oscillations.

(1)

Works with RSNto stabilize the output stage by creating a pole that eliminates high frequency oscillations.

fc= 1/(2πRSNCSN) Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce the Q of the

series resonant circuit due to capacitive load. Also provides a low impedance at low frequencies to short out R and pass audio signals to the load.

Provides power supply filtering and bypassing.

www.ti.com

OPTIONAL EXTERNAL COMPONENT INTERACTION

Although the optional external components have specific desired functions that are designed to reduce the bandwidth and eliminate unwanted high frequency oscillations they may cause certain undesirable effects when they interact. Interaction may occur for components whose reactances are in close proximity to one another. One example would be the coupling capacitor, CC, and the compensation capacitor, Cf. These two components act as low impedances to certain frequencies which will couple signals from the input to the output. Please take careful note of basic amplifier component functionality when designing in these components.

The optional external components shown in Figure 3 and described above are applicable in both single and split voltage supply configurations.

Product Folder Links: LM3886

Page 9

LM3886

www.ti.com

SNAS091C –MAY 1999–REVISED MARCH 2013

Typical Performance Characteristics

Safe Area Protection Response

Figure 4. Figure 5.

Supply Current vs

Supply Voltage Pulse Thermal Resistance

SPiKe

Figure 6. Figure 7.

Pulse Thermal Resistance Output Voltage

Figure 8. Figure 9.

Product Folder Links: LM3886

Supply Current vs

Page 10

LM3886

SNAS091C –MAY 1999–REVISED MARCH 2013

Typical Performance Characteristics (continued)

Pulse Power Limit Pulse Power Limit

Figure 10. Figure 11.

Supply Current vs Input Bias Current vs Case Temperature Case Temperature

www.ti.com

Figure 12. Figure 13.

Clipping Voltage vs Clipping Voltage vs

Supply Voltage Supply Voltage

Figure 14. Figure 15.

Product Folder Links: LM3886

Page 11

LM3886

www.ti.com

SNAS091C –MAY 1999–REVISED MARCH 2013

Typical Performance Characteristics (continued)

THD + N THD + N

vs vs

Frequency Frequency

Figure 16. Figure 17.

THD + N THD + N

vs vs

Frequency Output Power

Figure 18. Figure 19.

THD + N THD + N

vs vs

Output Power Output Power

Figure 20. Figure 21.

Product Folder Links: LM3886

Page 12

LM3886

SNAS091C –MAY 1999–REVISED MARCH 2013

Typical Performance Characteristics (continued)

THD + N THD + N

vs vs

Output Power Output Power

Figure 22. Figure 23.

THD + N THD + N

vs vs

Output Power Output Power

www.ti.com

Figure 24. Figure 25.

THD + N THD + N

vs vs

Output Power Output Power

Figure 26. Figure 27.

Product Folder Links: LM3886

Page 13

LM3886

www.ti.com

SNAS091C –MAY 1999–REVISED MARCH 2013

Typical Performance Characteristics (continued)

THD + N Distribution THD + N Distribution

Figure 28. Figure 29.

THD + N Distribution THD + N Distribution

Figure 30. Figure 31.

THD + N Distribution Load Resistance

Figure 32. Figure 33.

Product Folder Links: LM3886

Output Power vs

Page 14

LM3886

SNAS091C –MAY 1999–REVISED MARCH 2013

www.ti.com

Typical Performance Characteristics (continued)

Max Heatsink Thermal Resistance (°C/W)

at the Specified Ambient Temperature (°C)

Maximum Power Dissipation

Note: The maximum heat sink thermal resistance values, øSA, in the table above were calculated using a øCS= 0.2°C/W due to thermal compound.

Power Dissipation Power Dissipation

vs Output Power vs Output Power

Supply Voltage

Figure 34.

Figure 35. Figure 36.

Output Power

vs Supply Voltage IMD 60 Hz, 4:1

Figure 37. Figure 38.

Product Folder Links: LM3886

Page 15

LM3886

www.ti.com

SNAS091C –MAY 1999–REVISED MARCH 2013

Typical Performance Characteristics (continued)

IMD 60 Hz, 7 kHz, 4:1 IMD 60 Hz, 7 kHz, 4:1

Figure 39. Figure 40.

IMD 60 Hz, 1:1 IMD 60 Hz, 7 kHz 1:1

Figure 41. Figure 42.

IMD 60 Hz, 7 kHz, 1:1 Mute Current

Figure 43. Figure 44.

Mute Attenuation vs

Product Folder Links: LM3886

Page 16

LM3886

SNAS091C –MAY 1999–REVISED MARCH 2013

Typical Performance Characteristics (continued)

Mute Attenuation vs

Mute Current Large Signal Response

Figure 45. Figure 46.

Power Supply Common-Mode

Rejection Ratio Rejection Ratio

www.ti.com

Figure 47. Figure 48.

Open Loop

Frequency Response

Figure 49.

Product Folder Links: LM3886

Page 17

LM3886

www.ti.com

SNAS091C –MAY 1999–REVISED MARCH 2013

APPLICATION INFORMATION

GENERAL FEATURES

Mute Function: The muting function of the LM3886 allows the user to mute the music going into the amplifier by

drawing less than 0.5 mA out of pin 8 of the device. This is accomplished as shown in the Typical Application Circuit where the resistor RMis chosen with reference to your negative supply voltage and is used in conjuction with a switch. The switch (when opened) cuts off the current flow from pin 8 to V−, thus placing the LM3886 into mute mode. Refer to Figure 44 and Figure 45 in Typical Performance

Characteristics for values of attenuation per current out of pin 8. The resistance RMis calculated by the

following equation:

RM(|VEE| − 2.6V)/I8

where

• I8 ≥ 0.5 mA. (1)

Under-Voltage Protection: Upon system power-up the under-voltage protection circuitry allows the power

supplies and their corresponding caps to come up close to their full values before turning on the LM3886 such that no DC output spikes occur. Upon turn-off, the output of the LM3886 is brought to ground before the power supplies such that no transients occur at power-down.

Over-Voltage Protection: The LM3886 contains overvoltage protection circuitry that limits the output current to

approximately 11Apeak while also providing voltage clamping, though not through internal clamping diodes. The clamping effect is quite the same, however, the output transistors are designed to work alternately by sinking large current spikes.

SPiKe Protection: The LM3886 is protected from instantaneous peak-temperature stressing by the power

transistor array. Figure 4 in Typical Performance Characteristics shows the area of device operation where the SPiKe Protection Circuitry is not enabled. The waveform to the right of the SOA graph exemplifies how the dynamic protection will cause waveform distortion when enabled.

Thermal Protection: The LM3886 has a sophisticated thermal protection scheme to prevent long-term thermal

stress to the device. When the temperature on the die reaches 165°C, the LM3886 shuts down. It starts operating again when the die temperature drops to about 155°C, but if the temperature again begins to rise, shutdown will occur again at 165°C. Therefore the device is allowed to heat up to a relatively high temperature if the fault condition is temporary, but a sustained fault will cause the device to cycle in a Schmitt Trigger fashion between the thermal shutdown temperature limits of 165°C and 155°C. This greatly reduces the stress imposed on the IC by thermal cycling, which in turn improves its reliability under sustained fault conditions. Since the die temperature is directly dependent upon the heat sink, the heat sink should be chosen as discussed in THERMAL CONSIDERATIONS, such that thermal shutdown will not be reached during normal operation. Using the best heat sink possible within the cost and space constraints of the system will improve the long-term reliability of any power semiconductor device.

THERMAL CONSIDERATIONS

Heat Sinking

The choice of a heat sink for a high-power audio amplifier is made entirely to keep the die temperature at a level such that the thermal protection circuitry does not operate under normal circumstances. The heat sink should be chosen to dissipate the maximum IC power for a given supply voltage and rated load.

With high-power pulses of longer duration than 100 ms, the case temperature will heat up drastically without the use of a heat sink. Therefore the case temperature, as measured at the center of the package bottom, is entirely dependent on heat sink design and the mounting of the IC to the heat sink. For the design of a heat sink for your audio amplifier application refer to Determining the Correct Heat Sink.

Since a semiconductor manufacturer has no control over which heat sink is used in a particular amplifier design, we can only inform the system designer of the parameters and the method needed in the determination of a heat sink. With this in mind, the system designer must choose his supply voltages, a rated load, a desired output power level, and know the ambient temperature surrounding the device. These parameters are in addition to knowing the maximum junction temperature and the thermal resistance of the IC, both of which are provided by Texas Instruments.

Product Folder Links: LM3886

Page 18

LM3886

SNAS091C –MAY 1999–REVISED MARCH 2013

www.ti.com

As a benefit to the system designer we have provided Figure 34 for various loads in Typical Performance

Characteristics, giving an accurate figure for the maximum thermal resistance required for a particular amplifier

design. This data was based on θJC= 1°C/W and θCS= 0.2°C/W. We also provide a section regarding heat sink determination for any audio amplifier design where θCSmay be a different value. It should be noted that the idea behind dissipating the maximum power within the IC is to provide the device with a low resistance to convection heat transfer such as a heat sink. Therefore, it is necessary for the system designer to be conservative in his heat sink calculations. As a rule, the lower the thermal resistance of the heat sink the higher the amount of power that may be dissipated. This is of course guided by the cost and size requirements of the system. Convection cooling heat sinks are available commercially, and their manufacturers should be consulted for ratings.

Proper mounting of the IC is required to minimize the thermal drop between the package and the heat sink. The heat sink must also have enough metal under the package to conduct heat from the center of the package bottom to the fins without excessive temperature drop.

A thermal grease such as Wakefield type 120 or Thermalloy Thermacote should be used when mounting the package to the heat sink. Without this compound, thermal resistance will be no better than 0.5°C/W, and probably much worse. With the compound, thermal resistance will be 0.2°C/W or less, assuming under 0.005 inch combined flatness runout for the package and heat sink. Proper torquing of the mounting bolts is important and can be determined from heat sink manufacturer's specification sheets.

Should it be necessary to isolate V−from the heat sink, an insulating washer is required. Hard washers like beryluum oxide, anodized aluminum and mica require the use of thermal compound on both faces. Two-mil mica washers are most common, giving about 0.4°C/W interface resistance with the compound.

Silicone-rubber washers are also available. A 0.5°C/W thermal resistance is claimed without thermal compound. Experience has shown that these rubber washers deteriorate and must be replaced should the IC be dismounted.

Determining Maximum Power Dissipation

Power dissipation within the integrated circuit package is a very important parameter requiring a thorough understanding if optimum power output is to be obtained. An incorrect maximum power dissipation (PD) calculation may result in inadequate heat sinking, causing thermal shutdown circuitry to operate and limit the output power.

The following equations can be used to accurately calculate the maximum and average integrated circuit power dissipation for your amplifier design, given the supply voltage, rated load, and output power. These equations can be directly applied to Figure 35 in Typical Performance Characteristics.

Equation 2 exemplifies the maximum power dissipation of the IC and Equation 3, Equation 4 exemplify the

average IC power dissipation expressed in different forms.

= VCC2/2π2R

DMAX

where

• VCCis the total supply voltage (2)

DAVE

= (V

Opk/RL

)[VCC/π − V

Opk

/2]

where

• VCCis the total supply voltage and V

DAVE

= VCCV

/πRL− V

Opk

/2R

= VCC/π (3)

Opk

where

• VCCis the total supply voltage (4)

Determining the Correct Heat Sink

Once the maximum IC power dissipation is known for a given supply voltage, rated load, and the desired rated output power the maximum thermal resistance (in °C/W) of a heat sink can be calculated. This calculation is made using Equation 6 and is based on the fact that thermal heat flow parameters are analogous to electrical current flow properties.

It is also known that typically the thermal resistance, θJC(junction to case), of the LM3886 is 1°C/W and that using Thermalloy Thermacote thermal compound provides a thermal resistance, θCS(case to heat sink), of about

0.2°C/W as explained in Heat Sinking.

Product Folder Links: LM3886

Page 19

LM3886

www.ti.com

SNAS091C –MAY 1999–REVISED MARCH 2013

Referring to the figure below, it is seen that the thermal resistance from the die (junction) to the outside air (ambient) is a combination of three thermal resistances, two of which are known, θJCand θCS. Since convection heat flow (power dissipation) is analogous to current flow, thermal resistance is analogous to electrical resistance, and temperature drops are analogous to voltage drops, the power dissipation out of the LM3886 is equal to the following:

DMAX

= (T

Jmax

− T

Amb

)/θ

where

• θJA= θJC+ θCS+ θ

(5)

Figure 50.

But since we know P

= [(T

Jmax

− T

, θJC, and θSCfor the application and we are looking for θSA, we have the following:

DMAX

Amb

) − P

DMAX(θJC

+ θCS)]/P

DMAX

(6)

Again it must be noted that the value of θSAis dependent upon the system designer's amplifier application and its corresponding parameters as described previously. If the ambient temperature that the audio amplifier is to be working under is higher than the normal 25°C, then the thermal resistance for the heat sink, given all other things are equal, will need to be smaller.

Equation 2 and Equation 6 are the only equations needed in the determination of the maximum heat sink thermal

resistance. This is of course given that the system designer knows the required supply voltages to drive his rated load at a particular power output level and the parameters provided by the semiconductor manufacturer. These parameters are the junction to case thermal resistance, θJC, T

= 150°C, and the recommended Thermalloy

Jmax

Thermacote thermal compound resistance, θCS.

SIGNAL-TO-NOISE RATIO

In the measurement of the signal-to-noise ratio, misinterpretations of the numbers actually measured are common. One amplifier may sound much quieter than another, but due to improper testing techniques, they appear equal in measurements. This is often the case when comparing integrated circuit designs to discrete amplifier designs. Discrete transistor amps often “run out of gain” at high frequencies and therefore have small bandwidths to noise as indicated below.

Integrated circuits have additional open loop gain allowing additional feedback loop gain in order to lower harmonic distortion and improve frequency response. It is this additional bandwidth that can lead to erroneous signal-to-noise measurements if not considered during the measurement process. In the typical example above, the difference in bandwidth appears small on a log scale but the factor of 10 in bandwidth, (200 kHz to 2 MHz) can result in a 10 dB theoretical difference in the signal-to-noise ratio (white noise is proportional to the square root of the bandwidth in a system).

In comparing audio amplifiers it is necessary to measure the magnitude of noise in the audible bandwidth by using a “weighting” filter (see Note below). A “weighting” filter alters the frequency response in order to compensate for the average human ear's sensitivity to the frequency spectra. The weighting filters at the same time provide the bandwidth limiting as discussed in the previous paragraph.

Product Folder Links: LM3886

Page 20

LM3886

SNAS091C –MAY 1999–REVISED MARCH 2013

www.ti.com

NOTE

CCIR/ARM: A Practical Noise Measurement Method; by Ray Dolby, David Robinson and Kenneth Gundry, AES Preprint No. 1353 (F-3).

In addition to noise filtering, differing meter types give different noise readings. Meter responses include:

1. RMS reading,

2. average responding,

3. peak reading, and

4. quasi peak reading.

Although theoretical noise analysis is derived using true RMS based calculations, most actual measurements are taken with ARM (Average Responding Meter) test equipment.

Typical signal-to-noise figures are listed for an A-weighted filter which is commonly used in the measurement of noise. The shape of all weighting filters is similar, with the peak of the curve usually occurring in the 3 kHz–7 kHz region as shown below.

Figure 51.

SUPPLY BYPASSING

The LM3886 has excellent power supply rejection and does not require a regulated supply. However, to eliminate possible oscillations all op amps and power op amps should have their supply leads bypassed with lowinductance capacitors having short leads and located close to the package terminals. Inadequate power supply bypassing will manifest itself by a low frequency oscillation known as “motorboating” or by high frequency instabilities. These instabilities can be eliminated through multiple bypassing utilizing a large tantalum or electrolytic capacitor (10 μF or larger) which is used to absorb low frequency variations and a small ceramic capacitor (0.1 μF) to prevent any high frequency feedback through the power supply lines.

If adequate bypassing is not provided the current in the supply leads which is a rectified component of the load current may be fed back into internal circuitry. This signal causes low distortion at high frequencies requiring that the supplies be bypassed at the package terminals with an electrolytic capacitor of 470 μF or more.

LEAD INDUCTANCE

Power op amps are sensitive to inductance in the output lead, particularly with heavy capacitive loading. Feedback to the input should be taken directly from the output terminal, minimizing common inductance with the load.

Lead inductance can also cause voltage surges on the supplies. With long leads to the power supply, energy is stored in the lead inductance when the output is shorted. This energy can be dumped back into the supply bypass capacitors when the short is removed. The magnitude of this transient is reduced by increasing the size of the bypass capacitor near the IC. With at least a 20 μF local bypass, these voltage surges are important only if the lead length exceeds a couple feet (>1 μH lead inductance). Twisting together the supply and ground leads minimizes the effect.

Product Folder Links: LM3886

Page 21

LM3886

www.ti.com

SNAS091C –MAY 1999–REVISED MARCH 2013

LAYOUT, GROUND LOOPS AND STABILITY

The LM3886 is designed to be stable when operated at a closed-loop gain of 10 or greater, but as with any other high-current amplifier, the LM3886 can be made to oscillate under certain conditions. These usually involve printed circuit board layout or output/input coupling.

When designing a layout, it is important to return the load ground, the output compensation ground, and the low level (feedback and input) grounds to the circuit board common ground point through separate paths. Otherwise, large currents flowing along a ground conductor will generate voltages on the conductor which can effectively act as signals at the input, resulting in high frequency oscillation or excessive distortion. It is advisable to keep the output compensation components and the 0.1 μF supply decoupling capacitors as close as possible to the LM3886 to reduce the effects of PCB trace resistance and inductance. For the same reason, the ground return paths should be as short as possible.

In general, with fast, high-current circuitry, all sorts of problems can arise from improper grounding which again can be avoided by returning all grounds separately to a common point. Without isolating the ground signals and returning the grounds to a common point, ground loops may occur.

“Ground Loop” is the term used to describe situations occurring in ground systems where a difference in potential exists between two ground points. Ideally a ground is a ground, but unfortunately, in order for this to be true, ground conductors with zero resistance are necessary. Since real world ground leads possess finite resistance, currents running through them will cause finite voltage drops to exist. If two ground return lines tie into the same path at different points there will be a voltage drop between them. The first figure below shows a common ground example where the positive input ground and the load ground are returned to the supply ground point via the same wire. The addition of the finite wire resistance, R2, results in a voltage difference between the two points as shown below.

The load current ILwill be much larger than input bias current II, thus V1will follow the output voltage directly, i.e. in phase. Therefore the voltage appearing at the non-inverting input is effectively positive feedback and the circuit may oscillate. If there were only one device to worry about then the values of R1and R2would probably be small enough to be ignored; however, several devices normally comprise a total system. Any ground return of a separate device, whose output is in phase, can feedback in a similar manner and cause instabilities. Out of phase ground loops also are troublesome, causing unexpected gain and phase errors.

The solution to most ground loop problems is to always use a single-point ground system, although this is sometimes impractical. The third figure below is an example of a single-point ground system.

Product Folder Links: LM3886

Page 22

LM3886

SNAS091C –MAY 1999–REVISED MARCH 2013

www.ti.com

The single-point ground concept should be applied rigorously to all components and all circuits when possible. Violations of single-point grounding are most common among printed circuit board designs, since the circuit is surrounded by large ground areas which invite the temptation to run a device to the closest ground spot. As a final rule, make all ground returns low resistance and low inductance by using large wire and wide traces.

Occasionally, current in the output leads (which function as antennas) can be coupled through the air to the amplifier input, resulting in high-frequency oscillation. This normally happens when the source impedance is high or the input leads are long. The problem can be eliminated by placing a small capacitor, CC, (on the order of 50 pF to 500 pF) across the LM3886 input terminals. Refer to External Components Description relating to component interaction with Cf.

REACTIVE LOADING

It is hard for most power amplifiers to drive highly capacitive loads very effectively and normally results in oscillations or ringing on the square wave response. If the output of the LM3886 is connected directly to a capacitor with no series resistance, the square wave response will exhibit ringing if the capacitance is greater than about 0.2 μF. If highly capacitive loads are expected due to long speaker cables, a method commonly employed to protect amplifiers from low impedances at high frequencies is to couple to the load through a 10Ω resistor in parallel with a 0.7 μH inductor. The inductor-resistor combination as shown in Typical Application isolates the feedback amplifier from the load by providing high output impedance at high frequencies thus allowing the 10Ω resistor to decouple the capacitive load and reduce the Q of the series resonant circuit. The LR combination also provides low output impedance at low frequencies thus shorting out the 10Ω resistor and allowing the amplifier to drive the series RC load (large capacitive load due to long speaker cables) directly.

GENERALIZED AUDIO POWER AMPLIFIER DESIGN

The system designer usually knows some of the following parameters when starting an audio amplifier design:

Desired Power Output Input Level Input Impedance Load Impedance Maximum Supply Voltage Bandwidth

The power output and load impedance determine the power supply requirements, however, depending upon the application some system designers may be limited to certain maximum supply voltages. If the designer does have a power supply limitation, he should choose a practical load impedance which would allow the amplifier to provide the desired output power, keeping in mind the current limiting capabilities of the device. In any case, the output signal swing and current are found from (where POis the average output power):

(7) (8)

To determine the maximum supply voltage the following parameters must be considered. Add the dropout voltage (4V for LM3886) to the peak output swing, V current of I

). The regulation of the supply determines the unloaded voltage, usually about 15% higher.

opeak

, to get the supply rail value (i.e. ± (V

opeak

+ Vod) at a

Supply voltage will also rise 10% during high line conditions. Therefore, the maximum supply voltage is obtained from the following equation:

Max. supplies ≊ ± (V

+ Vod)(1 + regulation)(1.1) (9)

opeak

The input sensitivity and the output power specs determine the minimum required gain as depicted below:

(10)

Normally the gain is set between 20 and 200; for a 40W, 8Ω audio amplifier this results in a sensitivity of 894 mV and 89 mV, respectively. Although higher gain amplifiers provide greater output power and dynamic headroom capabilities, there are certain shortcomings that go along with the so called “gain.” The input referred noise floor is increased and hence the SNR is worse. With the increase in gain, there is also a reduction of the power bandwidth which results in a decrease in feedback thus not allowing the amplifier to respond quickly enough to nonlinearities. This decreased ability to respond to nonlinearities increases the THD + N specification.

Product Folder Links: LM3886

Page 23

LM3886

www.ti.com

SNAS091C –MAY 1999–REVISED MARCH 2013

The desired input impedance is set by RIN. Very high values can cause board layout problems and DC offsets at the output. The value for the feedback resistance, Rf1, should be chosen to be a relatively large value (10 kΩ–100 kΩ), and the other feedback resistance, Ri, is calculated using standard op amp configuration gain equations. Most audio amplifiers are designed from the non-inverting amplifier configuration.

DESIGN A 40W/4Ω AUDIO AMPLIFIER

Given:

Power Output 40W Load Impedance 4Ω Input Level 1V(max) Input Impedance 100 kΩ Bandwidth 20 Hz–20 kHz ± 0.25 dB

Equation 7 and Equation 8 give:

40W/4Ω V Therefore the supply required is: ±21.0V at 4.5A With 15% regulation and high line the final supply voltage is ±26.6V using Equation 9. At this point it is a good

idea to check the Power Output vs Supply Voltage to ensure that the required output power is obtainable from the device while maintaining low THD + N. It is also good to check the Power Dissipation vs Supply Voltage to ensure that the device can handle the internal power dissipation. At the same time designing in a relatively practical sized heat sink with a low thermal resistance is also important. Refer to Typical Performance

Characteristics graphs and THERMAL CONSIDERATIONS for more information.

The minimum gain from Equation 10 is: AV≥ 12.6 We select a gain of 13 (Non-Inverting Amplifier); resulting in a sensitivity of 973 mV. Letting RINequal 100 kΩ gives the required input impedance, however, this would eliminate the “volume control”

unless an additional input impedance was placed in series with the 10 kΩ potentiometer that is depicted in

Figure 1. Adding the additional 100 kΩ resistor would ensure the minumum required input impedance.

For low DC offsets at the output we let Rf1= 100 kΩ. Solving for Ri (Non-Inverting Amplifier) gives the following: Ri = Rf1/(AV− 1) = 100k/(13 − 1) = 8.3 kΩ; use 8.2 kΩ The bandwidth requirement must be stated as a pole, i.e., the 3 dB frequency. Five times away from a pole gives

0.17 dB down, which is better than the required 0.25 dB. Therefore:

fL= 20 Hz/5 = 4 Hz (11) fH= 20 kHz × 5 = 100 kHz (12)

At this point, it is a good idea to ensure that the Gain-Bandwidth Product for the part will provide the designed gain out to the upper 3 dB point of 100 kHz. This is why the minimum GBWP of the LM3886 is important.

GBWP ≥ AV× f3 dB = 13 × 100 kHz = 1.3 MHz (13) GBWP = 2.0 MHz (min) for the LM3886 (14)

Solving for the low frequency roll-off capacitor, Ci, we have:

Ci ≥ 1/(2π Ri fL) = 4.85 μF; use 4.7 μF. (15)

= 17.9V I

opeak

= 4.5A

Definition of Terms

Input Offset Voltage: The absolute value of the voltage which must be applied between the input terminals

through two equal resistances to obtain zero output voltage and current.

Input Bias Current: The absolute value of the average of the two input currents with the output voltage and

current at zero.

Input Offset Current:The absolute value of the difference in the two input currents with the output voltage and

current at zero.

Input Common-Mode Voltage Range (or Input Voltage Range): The range of voltages on the input terminals

Product Folder Links: LM3886

Page 24

LM3886

SNAS091C –MAY 1999–REVISED MARCH 2013

www.ti.com

for which the amplifier is operational. Note that the specifications are not specified over the full commonmode voltage range unless specifically stated.

Common-Mode Rejection: The ratio of the input common-mode voltage range to the peak-to-peak change in

input offset voltage over this range.

Power Supply Rejection: The ratio of the change in input offset voltage to the change in power supply voltages

producing it.

Quiescent Supply Current: The current required from the power supply to operate the amplifier with no load

and the output voltage and current at zero.

Slew Rate: The internally limited rate of change in output voltage with a large amplitude step function applied to

the input.

Class B Amplifier: The most common type of audio power amplifier that consists of two output devices each of

which conducts for 180° of the input cycle. The LM3886 is a Quasi-AB type amplifier.

Crossover Distortion: Distortion caused in the output stage of a class B amplifier. It can result from inadequate

bias current providing a dead zone where the output does not respond to the input as the input cycle goes through its zero crossing point. Also for ICs an inadequate frequency response of the output PNP device can cause a turn-on delay giving crossover distortion on the negative going transition through zero crossing at the higher audio frequencies.

THD + N: Total Harmonic Distortion plus Noise refers to the measurement technique in which the fundamental

component is removed by a bandreject (notch) filter and all remaining energy is measured including harmonics and noise.

Signal-to-Noise Ratio: The ratio of a system's output signal level to the system's output noise level obtained in

the absence of a signal. The output reference signal is either specified or measured at a specified distortion level.

Continuous Average Output Power: The minimum sine wave continuous average power output in watts (or

dBW) that can be delivered into the rated load, over the rated bandwidth, at the rated maximum total harmonic distortion

Music Power: A measurement of the peak output power capability of an amplifier with either a signal duration

sufficiently short that the amplifier power supply does not sag during the measurement, or when high quality external power supplies are used. This measurement (an IHF standard) assumes that with normal music program material the amplifier power supplies will sag insignificantly.

Peak Power: Most commonly referred to as the power output capability of an amplifier that can be delivered to

the load; specified by the part's maximum voltage swing.

Headroom: The margin between an actual signal operating level (usually the power rating of the amplifier with

particular supply voltages, a rated load value, and a rated THD + N figure) and the level just before clipping distortion occurs, expressed in decibels.

Large Signal Voltage Gain: The ratio of the output voltage swing to the differential input voltage required to

drive the output from zero to either swing limit. The output swing limit is the supply voltage less a specified quasi-saturation voltage. A pulse of short enough duration to minimize thermal effects is used as a measurement signal.

Output-Current Limit: The output current with a fixed output voltage and a large input overdrive. The limiting

current drops with time once SPiKe protection circuitry is activated.

Output Saturation Threshold (Clipping Point): The output swing limit for a specified input drive beyond that

required for zero output. It is measured with respect to the supply to which the output is swinging.

Output Resistance: The ratio of the change in output voltage to the change in output current with the output

around zero.

Power Dissipation Rating: The power that can be dissipated for a specified time interval without activating the

protection circuitry. For time intervals in excess of 100 ms, dissipation capability is determined by heat sinking of the IC package rather than by the IC itself.

Thermal Resistance: The peak, junction-temperature rise, per unit of internal power dissipation (units in °C/W),

above the case temperature as measured at the center of the package bottom.

Product Folder Links: LM3886

Page 25

LM3886

www.ti.com

SNAS091C –MAY 1999–REVISED MARCH 2013

The DC thermal resistance applies when one output transistor is operating continuously. The AC thermal resistance applies with the output transistors conducting alternately at a high enough frequency that the peak capability of neither transistor is exceeded.

Power Bandwidth: The power bandwidth of an audio amplifier is the frequency range over which the amplifier

voltage gain does not fall below 0.707 of the flat band voltage gain specified for a given load and output power. Power bandwidth also can be measured by the frequencies at which a specified level of distortion is obtained while the amplifier delivers a power output 3 dB below the rated output. For example, an amplifier rated at 60W with ≤ 0.25% THD + N, would make its power bandwidth measured as the difference between the upper and lower frequencies at which 0.25% distortion was obtained while the amplifier was delivering 30W.

Gain-Bandwidth Product: The Gain-Bandwidth Product is a way of predicting the high-frequency usefulness of

an op amp. The Gain-Bandwidth Product is sometimes called the unity-gain frequency or unity-gain cross frequency because the open-loop gain characteristic passes through or crosses unity gain at this frequency. Simply, we have the following relationship: A Assuming that at unity-gain (A

= 1 or (0 dB)) fu = fi = GBWP, then we have the following: GBWP = A

CL1

× f1= A

CL2

× f

CL2

× f2 This says that once fu (GBWP) is known for an amplifier, then the open-loop gain can be found at any frequency. This is also an excellent equation to determine the 3 dB point of a closed-loop gain, assuming that you know the GBWP of the device. Refer to Figure 52.

Biamplification: The technique of splitting the audio frequency spectrum into two sections and using individual

power amplifiers to drive a separate woofer and tweeter. Crossover frequencies for the amplifiers usually vary between 500 Hz and 1600 Hz. “Biamping” has the advantages of allowing smaller power amps to produce a given sound pressure level and reducing distortion effects produced by overdrive in one part of the frequency spectrum affecting the other part.

C.C.I.R./A.R.M.: Literally: International Radio Consultative Committee/Average Responding Meter

This refers to a weighted noise measurement for a Dolby B type noise reduction system. A filter characteristic is used that gives a closer correlation of the measurement with the subjective annoyance of noise to the ear. Measurements made with this filter cannot necessarily be related to unweighted noise measurements by some fixed conversion factor since the answers obtained will depend on the spectrum of the noise source.

S.P.L.: Sound Pressure Level—usually measured with a microphone/meter combination calibrated to a pressure

level of 0.0002 μBars (approximately the threshold hearing level). S.P.L. = 20 Log 10P/0.0002 dB where P is the R.M.S. sound pressure in microbars. (1 Bar = 1 atmosphere = 14.5 lb/in2= 194 dB S.P.L.).

Figure 52.

Product Folder Links: LM3886

Page 26

LM3886

SNAS091C –MAY 1999–REVISED MARCH 2013

www.ti.com

REVISION HISTORY

Changes from Revision B (March 2013) to Revision C Page

• Changed layout of National Data Sheet to TI format .......................................................................................................... 25

Product Folder Links: LM3886

Page 27

PACKAGE OPTION ADDENDUM

www.ti.com

11-Jan-2021

PACKAGING INFORMATION

Orderable Device Status

LM3886T/NOPB ACTIVE TO-220 NDJ 11 20 RoHS & Green SN Level-1-NA-UNLIM 0 to 70 LM3886T

LM3886TF ACTIVE TO-220 NDA 11 20 Non-RoHS &

LM3886TF/NOPB ACTIVE TO-220 NDA 11 20 RoHS-Exempt

(1)

The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device.

Package Type Package

(1)

Drawing

Pins Package

Qty

Eco Plan

(2)

Non-Green

& Non-Green

Lead finish/

Ball material

(6)

Call TI Call TI 0 to 70 LM3886TF

SN Level-1-NA-UNLIM 0 to 70 LM3886TF

MSL Peak Temp

(3)

Op Temp (°C) Device Marking

(4/5)

(2)

RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance

do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free".

RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based

flame retardants must also meet the <=1000ppm threshold requirement.

(3)

MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.

(4)

There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.

(5)

Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation

of the previous line and the two combined represent the entire Device Marking for that device.

(6)

Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two

lines if the finish value exceeds the maximum column width.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.

Samples

Addendum-Page 1

Page 28

PACKAGE OPTION ADDENDUM

www.ti.com

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

11-Jan-2021

Addendum-Page 2

Page 29

PACKAGE MATERIALS INFORMATION

www.ti.com 5-Jan-2022

TUBE

*All dimensions are nominal

Device Package Name Package Type Pins SPQ L (mm) W (mm) T (µm) B (mm)

LM3886T/NOPB NDJ TO-220 11 20 502 34 12192 0

LM3886TF NDA TO-220 11 20 502 37 12700 10.29 LM3886TF NDA TO-220 11 20 502 37 12700 10.29

LM3886TF/NOPB NDA TO-220 11 20 502 37 12700 10.29

Pack Materials-Page 1

Page 30

NDJ0011B

MECHANICAL DATA

www.ti.com

TA11B (Rev B)

Page 31

NDA0011B

MECHANICAL DATA

www.ti.com

TF11B (Rev D)

Page 32

IMPORTANT NOTICE AND DISCLAIMER

TI PROVIDES TECHNICAL AND RELIABILITY DATA (INCLUDING DATA SHEETS), DESIGN RESOURCES (INCLUDING REFERENCE DESIGNS), APPLICATION OR OTHER DESIGN ADVICE, WEB TOOLS, SAFETY INFORMATION, AND OTHER RESOURCES “AS IS” AND WITH ALL FAULTS, AND DISCLAIMS ALL WARRANTIES, EXPRESS AND IMPLIED, INCLUDING WITHOUT LIMITATION ANY IMPLIED WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE OR NON-INFRINGEMENT OF THIRD PARTY INTELLECTUAL PROPERTY RIGHTS.

These resources are intended for skilled developers designing with TI products. You are solely responsible for (1) selecting the appropriate TI products for your application, (2) designing, validating and testing your application, and (3) ensuring your application meets applicable standards, and any other safety, security, regulatory or other requirements.

These resources are subject to change without notice. TI grants you permission to use these resources only for development of an application that uses the TI products described in the resource. Other reproduction and display of these resources is prohibited. No license is granted to any other TI intellectual property right or to any third party intellectual property right. TI disclaims responsibility for, and you will fully indemnify TI and its representatives against, any claims, damages, costs, losses, and liabilities arising out of your use of these resources.

TI’s products are provided subject to TI’s Terms of Sale or other applicable terms available either on ti.com or provided in conjunction with such TI products. TI’s provision of these resources does not expand or otherwise alter TI’s applicable warranties or warranty disclaimers for TI products.

TI objects to and rejects any additional or different terms you may have proposed. IMPORTANT NOTICE

Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265

Texas Instrument LM3886 User Manual

Specifications and Main Features

Frequently Asked Questions

User Manual

FEATURES

DESCRIPTION

APPLICATIONS

Typical Application

Connection Diagram

Absolute Maximum Ratings

Operating Ratings

Electrical Characteristics

Test Circuit #1- DC Electrical Test Circuit

Test Circuit #2 -AC Electrical Test Circuit

Single Supply Application Circuit

Equivalent Schematic

External Components Description

OPTIONAL EXTERNAL COMPONENT INTERACTION

Typical Performance Characteristics

APPLICATION INFORMATION

GENERAL FEATURES

THERMAL CONSIDERATIONS

Heat Sinking

Determining Maximum Power Dissipation

Determining the Correct Heat Sink

SIGNAL-TO-NOISE RATIO

SUPPLY BYPASSING

LEAD INDUCTANCE

LAYOUT, GROUND LOOPS AND STABILITY

REACTIVE LOADING

GENERALIZED AUDIO POWER AMPLIFIER DESIGN

DESIGN A 40W/4Ω AUDIO AMPLIFIER

Definition of Terms

REVISION HISTORY