ST AN937 Application note

AN937

Application Note

Designing with L4971, 1.5 A high efficiency DC-DC converter

Introduction

The L4971 is a 1.5 A monolithic dc-dc converter, stepdown, operating at fix frequency continuous mode. It is designed in BCD60 II technology, and it is available in two plastic packages, DIP8 and SO16L.

One direct fixed output voltage at 3.3 V ±1% is available, adjustable for higher output voltage values, till 40 V, by an external voltage divider.

The operating input supply voltage ranges from 8 V to 55 V, while the absolute value, with no load, is 60V. New internal design solutions and superior technology performance allow to generate a device with improved efficiency in all the operating conditions and with reduced EMI due to an innovative internal driving circuit, and reduced external component counts.

While internal limiting current and thermal shutdown are today considered standard protection functions, mandatory for a safe load supply, oscillator with voltage feed forward improves line regulation and overall control loop.

Soft-start avoids output over voltage at turn-on, while, shorting this pin to ground, the device is completely disabled, going into zero consumption state.

Figure 1. Demonstration board

May 2011

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Contents	AN937

Contents

1	Device description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .		. 3
2	Power supply, UVLO and voltage reference . . . . . . . . . . . . . . . . . . . . . .		4
	2.1	Oscillator and voltage feed forward . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	4
	2.2	Current protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	6
	2.3	Soft-start and inhibit functions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	10
	2.4	Feedback disconnection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	14
	2.5	Zero load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	14
	2.6	Output overvoltage protection (OVP) . . . . . . . . . . . . . . . . . . . . . . . . . . . .	14
	2.7	Power stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	15
	2.8	Turn - on . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	15
	2.9	Turn - off . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	16
3	Typical application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .		17
	3.1	Electrical specification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	17
	3.2	Input capacitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	17
	3.3	Output voltage selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	19
	3.4	Inductor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	19
	3.5	Output capacitors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	21
	3.6	Compensation network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	22
	3.7	Error amplifier and compensation blocks . . . . . . . . . . . . . . . . . . . . . . . . .	23
	3.8	LC filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	23
	3.9	PWM gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	23
	3.10	Voltage divider and leading network . . . . . . . . . . . . . . . . . . . . . . . . . . . . .	23
4	Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .		28

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AN937	Device description

1 Device description

For a better understanding of the device and it’s working principle, a short description of the main building blocks is given here below, with packaging options and complete block diagram.

Figure 2 show s the two packaging options, with the pin function assignments.

Figure 2. Pin Connections

				N.C.	1	16	N.C.
GND	1	8	FB	GND	2	15	N.C.
SS_INH	2	7	COMP	SS_INH	3	14	FB
OSC	3	6	BOOT	OSC	4	13	COMP
OUT	4	5	VCC	OUT	5	12	BOOT
		D97IN595		OUT	6	11	VCC
				N.C.	7	10	N.C.
				N.C.	8	9	N.C.
						D97IN596

DIP8

SO16L

Figure 3. Block diagram.

					VCC
					5
	THERMAL	VOLTAGES
	SHUTDOWN	MONITOR
						CBOOT
2						CHARGE
			INTERNAL	INTERNAL
SS_INH	INHIBIT	SOFTSTART
			REFERENCE	SUPPLY	5.1V
		3.3V
7						6
COMP
8	E/A					BOOT
		PWM
FB			R
3.3V			Q			CBOOT
			S			CHARGE
					DRIVE	AT LIGHT
	OSCILLATOR					LOADS
	3		1		4
	OSC		GND		OUT	D97IN594

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Power supply, UVLO and voltage reference	AN937

2 Power supply, UVLO and voltage reference

The device is provided with an internal stabilized power supply (of about 12 V typ.) that powers the analog and digital control blocks and the bootstrap section.

From this pre regulator, a 3.3V reference voltage ±2%, is internally available.

2.1Oscillator and voltage feed forward

Just one pin is necessary to implement the oscillator function, with inherent voltage feedforward.

Figure 4. Oscillator internal circuit

	VCC
ROSC	TO PWM	5R
	COMPARATOR
	Osc	-	CLOCK
		+
	Q1	R	Q2
COSC
		1V
			D97IN655A

A resistor Rosc and a capacitor Cosc connected as shown in Figure 4, allow the setting of the desired switching frequency in agreement with the below formula:

1

FSW = -------------------------------------------------------------------------------------

( ) 6

Roasc Cosc In --5 + 100 Cosc

Where Fsw is in kHz, Rosc in kΩ and Cosc in nF.

The oscillator capacitor, Cosc, is discharged by an internal MOS transistor of 100 Ω of Rdson (Q1) and during this period the internal threshold is set at 1 V by a second MOS, Q2. When

the oscillator voltage capacitor reaches the 1V threshold the output comparator turn-off the MOS Q1 and turn-on the MOS Q2, restarting the Cosc charging.

The oscillator block, shown in Figure 4, generates a sawtooth wave signal that sets the switching frequency of the system.

This signal, compared with the output of the error amplifier, generates the PWM signal that will modulate the conduction time of the power output stage.

The way the oscillator has been integrated, does not require additional external components to benefit of the voltage feed forward function.

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AN937	Power supply, UVLO and voltage reference

The oscillator peak-to-valley voltage is proportional to the supply voltage, and the voltage feed forward is operative from 8 V to 55 V of input supply.

V VCC – 1 osc = --------------------

6

Also the V/ t of the sawtooth is directly proportional to the supply voltage. As Vcc increases, the Ton time of the power transistor decreases in such a way to provide to the choke, and finally also the load, the product Volt. sec constant.

Figure 6 show how the duty cycle varies as a result of the change on the V/ t of the sawtooth with the Vcc. The output of the error amplifier doesn’t change to maintain the output voltage constant and in regulation. With this function on board, the output response time is greatly reduced in presence of an abrupt change on the supply voltage, and the output ripple voltage at the mains frequency is greatly reduced too.

Figure 5. Device switching frequency vs Rosc and Cosc

fsw					D97IN630
(KHz)
500				Tamb=25˚C
		0.
200		82nF
		1.
100			2nF
			2.
			2nF
50			3.
			3nF
				4.
20				7nF
			5.
			6nF
10
5	20	40	60	80	R2(KΩ)
0

Figure 6. Voltage feed forward function

V1	Vi=30V
	Vi=15V
	Vc
	t
V2-3
	Vi=30V
	Vi=15V
	t
	D97IN684

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Figure 7. Maximum duty cycle vs. Rosc and Cosc as parameter

Dmax							D97IN685
		5.3nF
0.90			4.7nF
			2.2nF
			1.2nF
			0.8nF
0.80
0.70
0.60							32 ROSC(KΩ)
0	4	8	12 16	20	24	28

In fact, the slope of the ramp is modulated by the input ripple voltage, generally present in the order of some tents of Volt, for both off-line and dc-dc converters using mains transformers.

The charge and discharge time is approximately to:

Tch	= Rosc Cosc	6
		In --
		5

Tdis = 100 Cosc

The maximum duty cycle is a function of Tch, Tdis and an internal delay and is represented by the equation:

6 –9

Rosc Cosc In --5 – 80 10

Dmax = ---------------------------------------------------------------------------------

6

Rosc Cosc In --5 + 100 Cosc

and is represented in Figure 7:

2.2Current protection

The L4971 has two current limit levels, pulse by pulse and hiccup modes.

Increasing the output current till the pulse by pulse limiting current threshold (Ith1 typ. value of 2.5 A) the controller reduces the on-time till the value of TB = 300 ns that is a blanking time in which the current limit protection does not trigger. This minimum time is necessary to avoid undesirable intervention of the protection due to the spike current generated during the recovery time of the freewheeling diode.

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AN937	Power supply, UVLO and voltage reference

In this condition, because of this fixed blanking time, the output current is given by:

Imax = [R

[VCC TB FSW – Vf (1 – TB FSW)]

O

+ (R

D

+ R

L

)(1 – T

B

F

SW

) + (R

dson

+ R

L

)T

B

F

SW

]-

In Figure 8, the pulse by pulse protection is sufficient to limit the current.

In Figure 9 the pulse by pulse protection is no more effective to limit the current due to the minimum Ton fixed by the blanking time TB, and the hiccup protection intervenes because the output peak current reaches the relative threshold. At the pulse by pulse intervention (point A) the output voltage drops because of the Ton reduction, and the current is almost constant. Going versus the short circuit condition, the current is only limited by the series resistances RD and RL (see relation above) and could reach the hiccup threshold (point B), set 20% higher than the pulse by pulse. Once the hiccup limiting current is operating, in output short circuit condition, the delivered average output current decreases dramatically at very low values (point C).

Figure 8. Output characteristics

VO

A

		B
	C	IO
	D98IN908A
		2.5A 3A

Figure 9. Output characteristics

VO

A

D97IN809	2.5A	IO
		3A

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Figure 10. Current limiting internal schematic circuit
OSC S Q
R	VCC
VTh1	VTh2

+

+-

-

OUT

PWM

12V

+-

OSC	HICCUP	SOFT START
		LATCH
	THERMAL	S Q
	UNDERVOLTAGE	R
-	+
VFB	VREF
		+	CSS
			0.4
	D97IN658	-

Figure 11. Output current and soft start voltage

5.4A

4.5A

Figure 10 shows the internal current limiting circuitry. Vth1 is the pulse by pulse while Vth2 is the hiccup threshold.

The sense resistor is in series with a small MOS realized as a partition of the main DMOS.

The Vth2 comparator (20% higher than Vth1) sets the soft start latch, initializing the discharge of the soft start capacitor with a constant current (about 22 mA). Reaching about 0.4 V, the valley comparator resets the soft start latch, restarting a new recharge cycle.

Figure 11 shows the typical waveforms of the current in the output inductor and the soft start voltage (pin 2).

During the recharging of the soft start capacitor, the Ton increases gradually and, if the short circuit is still present, when Ton>TB and the output peak current reaches the threshold, the hiccup protection intervenes again. So, the value of the soft start capacitor must not be too high (in this case the Ton increases slowly thus taking much time to reach the TB value) to avoid that during the soft start slope, the current exceeds the limit before the protection activation.

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AN937	Power supply, UVLO and voltage reference

The following diagrams of Figure 13 and Figure 14 show the maximum allowed soft-start capacitor as a function of the input voltage, inductor value and switching frequency. A minimum value of the soft start capacitance is necessary to guarantee, in short circuit condition, the functionality of the limiting current circuitry.

In fact, with a capacitor too small, the frequency of the current peaks (see Figure 11) is high and the mean current value in short circuit increases.

Example: for a maximum input voltage of 55 V at 100 kHz, with an inductor of 260 mH, it is possible to use a soft start capacitor lower than 470 µF. With such a value, the soft start time (see Figure 19) of about 10 ms for an output voltage of 5 V.

Figure 12. Maximum soft start capacitance with fsw = 100 kHz

L								D97IN745
					680nF
(μH)
								470nF
		fsw=100KHz
400
								330nF
300
200							220nF
100								100nF
0
15	20	25	30	35	40	45	50	VCCmax(V)

Figure 13. Maximum soft start capacitance with fsw = 200 kHz

L	D97IN746
(μH)	56nF
fsw=200KHz
300	47nF
200	33nF
	22nF

100

0

15 20 25 30 35 40 45 50 VCCmax(V)

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