ST AN2844 Application note

AN2844

Applica t ion note

15 W wide range SMPS for metering

based on ESBT™ STC03DE220HV and L6565 PWM controller

1 Introduction

This document describes a 15-W flyback switched mode power supply (SMPS) application
that uses an emitter-switched bipolar transistor (ESBT™) switch (STC03DE220HV) and
L6565 quasi-resonant pulse-width modulation (PWM) controller.

The application is a universal, cost-effective flyback converter used in metering applications,
with an excellent wide-voltage input range from 125 to 1250 VDC, achieved using the ESBT
as the main switch and a quasi-resonant PWM driver.

This document is associated with the release of the demonstration board
STEVAL-ISA057V1 (Figure 1).

Figure 1. STEV AL- ISA057V1

March 2009 Rev 1 1/37

www.st.com

Contents AN2844

Contents

1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

2 Adapter features . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

2.1 Main characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

2.2 Circuit description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

3 Waveforms and results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10

4 Electrical performances . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17

5 Functional check . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

5.1 Startup behavior at full loads . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

5.2 Power-down . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

5.3 Short-circuit tests . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

6 Thermal measurements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

7 Bill of materials . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26

8 PCB layout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

9 Transformer specification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

9.1 Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

9.2 Mechanical aspects . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

10 Ordering information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33

11 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34

12 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35

13 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36

2/37

AN2844 List of tabl es

List of tables

Table 1. Line and load regulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17

Table 2. Efficiency at 125 VDC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19

Table 3. Effic iency at 500 VDC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19

Table 4. Efficiency at 1000 VDC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19

Table 5. Power consumpt ion in no-load conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

Table 6. Temperature of key componen ts at 125 VDC - full load

(3 A output current). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

Table 7. Temperature of key componen ts at 1250 VDC - low load

(3 A output current). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

Table 8. Bill of materials . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26

Table 9. Winding characteristics. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

Table 10. Document revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36

3/37

List of figures AN2844

List of figures

Figure 1. STEVAL- ISA057V1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

Figure 2. Electrical diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

Figure 3. ESBT - internal schematic and symbol. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

Figure 4. V
Figure 5. V
Figure 6. V

Figure 7. V

Figure 8. 620 VDC and maximum output power in steady-state conditions . . . . . . . . . . . . . . . . . . . 12

Figure 9. 620 VDC and no output power (no load) in steady-state conditions. . . . . . . . . . . . . . . . . . 13

Figure 10. 620 VDC and maximum output power in steady-state conditions
Figure 11. 620 VDC and maximum output power in steady-state conditions

Figure 12. 1250 VDC and maximum output power in steady-state conditions . . . . . . . . . . . . . . . . . . 14

Figure 13. 1250 VDC and no output power (no load) in steady-state conditions. . . . . . . . . . . . . . . . . 15

Figure 14. 1250 VDC and maximum output power in steady-state conditions
Figure 15. 1250 VDC and maximum output power in steady-state conditions
Figure 16. Efficiency versus output power at V

Figure 17. Efficiency versus output power at V
Figure 18. Efficiency versus output power at V

Figure 19. Connector description. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

Figure 20. Startup at 125 VDC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

Figure 21. Power-down at 125 VDC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

Figure 22. Short-circuit test at V
Figure 23. Short-circuit test at V
Figure 24. Short-circuit test at V

Figure 25. Silk screen - top side . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

Figure 26. Silk screen - bottom side . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

Figure 27. Copper tracks - top side . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29

Figure 28. Copper tracks - bottom side . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29

Figure 29. Electrical diagram of the transformer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

Figure 30. Winding position . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

Figure 31. Hole arrangement. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32

= 125 VDC and maximum output power in steady-state conditions. . . . . . . . . . . . . . . 10

IN

= 125 VDC and no output power (no load) in steady-state conditions . . . . . . . . . . . . . 11

IN

= 125 VDC and maximum output power in steady-state conditions

IN

- switch-ON highlighted. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

= 125 VDC and maximum output power in steady-state conditions

IN

- switch-OFF highlighted. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

- switch-ON highlighted. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

- switch-OFF highlighted. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14

- switch-ON highlighted. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

- switch-OFF highlighted. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

= 125 VDC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17

IN

= 500 VDC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

IN

= 1000 VDC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

IN

= 125 VDC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

IN

= 620 VDC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

IN

= 1250 VDC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

IN

4/37

AN2844 Adapter features

2 Adapter features

2.1 Main characteristics

The following is a list of the specifications and main parameters of the demonstration board.

●

Minimum input voltage: 125 VDC

●

Maximum input voltage: 1250 VDC

●

Output voltage: 5 V - 3.0 A

●

Maximum output power: 15 W

●

Short-circuit protection based on auto-res tart feature

●

Minimum switching frequency limited to 30 kHz

●

Overall converter efficiency > 60%

●

Non-galvanic isolated solution used in most metering applications

●

PCB type and size:
– material used for PCB: FR-4
– double-sided layout
– thickness of copper: 35 µm
– total diameter of demonstration board: 58 x 120 mm.

2.2 Circuit description

This device is a flyback converter, a very popular and well-known topology in switch-mode
power supply applications where the required output power is in the range of
5 to 200 W. The popularity of this type of converter comes from the simplicity of its design,
the small number of components and its resulting low cost compared to other topologies in
the same output power range.

The converter is based on the L6565 PWM driver that operates in quasi-resonant mode,
meaning zero voltage or valley switching during the turn-OFF phase. Current mode control
is the pri mary con tr o l method. An ESBT switch must be used for t hi s appl ication. The ESBT
is a cascade configuration of a high-voltage bipolar junction transistor (BJT) and a low­voltage power MOSFET. STMicroelectronics™ optimizes the performance of both devices
and offers this kind of switch in one single package so as to simplify the application’s design,
reduce electromagnetic interference and price, and increase reliability and performance
(see Figure 3). The ESBT switch, compa r e d to a high-volta ge switch, offers a low ON-state
voltage drop like a BJT. The switch is very robust, easy to drive and has a relatively fast
switching speed similar to that of a MOSFET.

For more detailed information on the design of the discontinuous conduction-mode flyba ck
converter, refer to the application note AN1889 "ESBT STC03DE170HV in
3-phase auxiliary power supply".

5/37

Adapter features AN2844

Figure 2. Electrical diagram

GND

+

1

2

5 V / 3 A

J2

OUT

connector

C5

100 nF

AM003536

T1 transf_ETD_29

C1

15 W SMPS 5 V / 3 A

Lp = 5.8 mH

Np / Ns = 70.2

1

Primary

2.2 nF / 2000 V
R3

R2

D1 BAV103

D3

STPS10L60

12

2

3

Primary

D2

STTH112D5STTH112

120 kΩ / 0.6 W

D4 STPS1150

120 kΩ / 0.6 W

Q1 2STF1360

C2

100 nF

C4

C3

R5

D6

Secondary

467

n.c.

BAV103

2.2 kΩ
R7

10

1000 µF

1000 µF

56 kΩ

R12 aux.

C7

D7

15 V

68 nF

C6

R16

C9

47 µF / 50 V

12 kΩ

2.2 µF

D8

12 kΩ

R21

C12

12 nF

Q3

BC847

Q2

STC03DE220HV

1

4

2

3

22 Ω

1.5 ΩR20
R26

C10

10 nF

R22

BAV103

910 Ω / 0.6 W

R27

GND
INV

FF

V

R42

10 kΩ

270 kΩ

Q7

BC847

47 nF

C19

Q6

BC847

C18

33 pF

C17

470 pF

6
1

3

C15

68 µF

/ 25 V

R38

10 kΩ

470 Ω

R39

R40

C14

100 pF

33 kΩ

R37

D9

BAV103

82 kΩ

R34

R36

R35

5
8

7

R33

R32

U1

ZCD

CC

V
OUT

910 Ω

10 kΩ

3.9 Ω

3.9 Ω

1 kΩ

L6565

2

COMP

CS

4

56 kΩR456 kΩ

R6

R10

R1 10 Ω

F1

Fuse 1 A

123

IN

J1

connector

R15

56 kΩ

R11

680 kΩ

R9

1.8

MΩ

R8

R13

200 kΩ

33 µFC8450 V

6/37

D10

Q4

STP03D200

56 kΩ

56 kΩ

R19

R25

1.8 MΩR14

1.8 MΩR18

200 kΩ

200 kΩ

R17

200 kΩ

R23

33 µF

450 V

C11

Q5

56 kΩ

1.8 MΩR24

R28

1.8 MΩR29

200 kΩ

C13

BC847

R30

33 µF

450 V

R31

1.8 MΩ

200 kΩ

18 V

R41

C16

22 kΩ

10 nF

AN2844 Adapter features

The operational voltage of the converter ranges from 125 to 1250 VDC, which enables the
demonstration board to be used in various technologies, particular ly in metering
applications. The output voltage is 5 V and the maximum output power is 15 W.

The board is protected with a 1-A fuse in the primary area. A negative temperature
coefficient (NTC) resistor is inserted in series with the input line to protect the demonstration
board from inrush current. For voltage purposes on the input DC line, the main 450 V
filtering capacitors are connected in series.

Figure 3. ESBT - internal schematic and symbol

C

C

B

B
G

G

S

S

AM003535

A non-dissipative active startup circuit has been implemented to optimize the converter’s
efficiency. The alternative option of using a pure resistive startup circuit was rejected due to
unacceptable power losses. The active startup circuit has been design ed with Q4, Q5 and
related passive parts. The R4, R6, R11, R15, R19 and R25 resistors provide the supply
current to the PWM driver during the start-up phases and have been calculated from the
minimal input supply voltage of the converter and required supply current of the PWM
controller L6565, plus the related supply current required to charge the C15 filtering
capacitor within a reasonable time to maintain an acceptable startup time. The balance
resistors R8, R13, R17 and R23 are used to ensure the same voltage drop across each
input capacitor. R28 and R30 supply the current into the base of the Q4 high-voltage
Darlington transistor. When the rising voltage on the C15 capacitor reaches the start-up
threshold of the L6565 PWM controller, the voltage on the transformer's auxiliary winding
turns on the Q5 transistor, which in turn shorts the base of the Q4 transistor. This means
that the active startup is blocked. The main power dissipation under normal working
conditions of the startup circuit is due to the balance resistors. Refer to AN2454 "Universal
input voltage power supply for ESBT-based breaker and metering applications" for more
information on active startup issues.

The self-supply circuit that provides the supply energy to the controller has been built
around Q1, which acts as a linear voltage regulator. This voltage regulator offers a stable
output supply voltage, which guarantees the performance of the converter’s overall input
voltage range at very low or no loads. The voltage regulator is mandatory in such
applications with a wide input voltage range. The value of the auxiliary voltage is set with the
Zener diode D7 and is approximately 14.5 V. The primary side of the converter incorporates
the L6565 PWM controller that includes all the features required for building a complete
system working in QR mode with a minimal number of external components.

Information relating to the zero voltage switching comes from the transformer’s auxiliary
winding. The auxiliary winding is also used for the controller’s self-supply. To k eep a
relatively constant output power across the entire input voltage range, the line voltage is fed
through resistors R9, R14, R18, R24, R29 and R31 into the line voltage feed-forward pin
(V

) of the L6565.

FF

7/37

Adapter features AN2844

This information is used to change the set point of the pulse-by-pulse current limitation. In
the standard application circuit with L6565 as U1, an OFF-time limitation circuit with a Q3
transistor has been added. This transistor limits the maximum switching frequency of the
converter to approximately 70 kHz. The input information coming from the OUT pin is fed
into Q3 with a delay. This feature provides some anticipated time dur ing the OFF time by
blocking the ZCD pin of the converter, and enables the reflected voltage on the auxiliary
winding to be sensed after that time. This means that the controller has skipped one or
several v alleys of the flyback voltage during the turn-OFF phases and has limited the
maximum switching frequency . The circuit’s time constant is set with capacitor C12 and
resistor R36. The limitations of the application’s frequency keep power losses on the
primary ESBT switch within reasonable values and contribute to the converter’s overall
stability. Refer to the L6565 datasheet for detailed information on the L6565 and function of
the circuit. All features, including the calculation of all setting components, are described in
STMicroelectronics’ application note AN1326 "L 6565 quasi-resonant controller". This
application note also describes the ZVS concept.

The output voltage is controlled by a non-galvanic isolated primary feedback loop with
resistor dividers R38 and R42, and frequency response compensation components R39,
R40 and C14. All formulas related to the calculation of the frequency response
compensation during first trials and testing in this type of application are described in
STMicroelectronics’ application note AN2495 "8 0 W very wide input voltage range 3-phase
SMPS design based on L6565 and ESBT STC04IE170HV".

In SMPS applications, where the load can vary, the current of the primary switch also
fluctuates. To minimize power losses on the ESBT switch, the base current should be
proportional to the collector current or at least constant with the initial current overpeak of
the switching pulse - this is called the modifying envelope. It is impor tant to avoid radical
over-saturation of the device at low loads and aim to optimize performance at full loads. To
achieve these driving requirements with a cost-effective solution, a simple driving circuit
providing a constant current into the base of the ESBT has been designed. This type of
solution is simple, cost-effective and minimizes power losses. The bias current for the base
of the ESBT is provided directly by the auxiliary power supply through the R22 resistor. For
a related base bias current, the value of this resistor is calculated according to the collector
current. According to the STC03DE220HV datasheet, for a maximum peak collector current
of 0.6 A, the gain and related base current should be 16 mA. During the storage time, when
the collector current for a c ertain period flows trough the B-C junction before this junction
recovers from conduction, the current flows into the C10 capacitor which stores some
energy and provides it again for the next switching cycle to create an initial base current
spike. Current that is not stored can also flow through D8 to the auxiliary supply area. Note
that during the storage time the collector current flows through the base and is stored in the
base capacitor C10, so that the quasi totality of energy is recovered. The R20 resistor limits
the inrush current floating from the C10 capacitor to the base of the Q2 transistor. In this
topology, the base current always has the same value and does not follow the variation of
the collector current, which appears while unloading the output. The constant base bias
current can caus e over-saturation of th e BJT struct u r e in the ESBT in low- or no-load
conditions. However, in a case like this, the driver guarantees the appropriate switching of
the ESBT th r ough the ZVS pin when the device is fully switched OFF and the carriers in the
BJT are fully recombined. This solution – which is very simple and cost-effective – implies
that the ESBT be driven with a constant current.

8/37

AN2844 Adapter features

ESBTs with this type of configuration offer very good performance in terms of power losses,
and have a low cost compared to other available switches such as very high voltage power
MOSFETs. For further information on driving networks, refer to STMicroelectronics’
application note AN2454 "Universal input voltage power supply for ESBT-based breaker and
metering applications".

The main T1 transformer used is a layered-type transformer, which uses a standard ETD29
core with a bobbin. The ETD29 bobbin has been chosen because of its strong voltage
isolation capacities at such high input voltages. In terms of just power requirements, an even
smaller core area than the ETD29 could be used. A sandwich topology has been used for
the design of the winding, offering better coupling of windings compared to standard
topologies with only one primary winding.

This transformer has been designed according to STMicroelectronics’ released application
notes, with a flyback voltage of 250 V. The turn ratio between the primary and secondary
side has been calculated and is approximately 70. Refer to AN1326 "Quasi-resonant

controller" and AN2495 "80 W very wide input voltage range 3-phase SMPS design based
on L6565 and ESBT STC04IE170HV" for all necessary calculations.

As is common in flyback applications, the total voltage across the switch can reach very high
voltages. The calculation is done with the formula:

V

= V

OFF

where Vfl is the flyback voltage = (V
the primary side while Ns is the number of turns on the secondary side. V

inmax

+ Vfl + V

spike

+ VF diode) x Np/Ns. Np is the number of turns on

OUT

spike

is the
maximum overvoltage allowed by the clamping network and has been fixed to 200 V.
Allowing for som e m argin, a related switch STC03DE220HV with a breakdown voltage of
2200 V fills the requirements for these types of applicatio n.

A clamp network is used for leakage inductance demagnetization. In this particular case,
a C1 capacitor with related passive resistors R2, R3 and bl ocking diodes D2 and D5 used in
series because of voltage stresses, has been selected for this purpose.

The secondary side comprises a S chot tky barri er diode D3 as rectifier, and filtering
capacitors C3 and C4 featuring low serial resistance. The short-circuit protection features for
the converter have been designed with transistors Q7 and Q6 and related passive parts.
The Q7 transistor senses the output voltage through the resistor dividers R38 and R42. In
normal conditions, the Q7 transistor keeps the Q6 transistor turned off. During a short-circuit
condition where the output voltage is very low or e qual to zero, the Q7 transistor is closed.
Energy stored in the tank capacitor C9 can start to provide the supply current for Q6, which
starts to block the function of the converter through the L6565’s ZCD pin. This condition
continues until all the energy from the C9 capacitor has been discharged. The time cycle is
set with the R37 resistor and capacity of C9. Once all the energy in C9 has been
discharged, the converter star ts to work again. If the short connection on the output is still
present, the short-circuit protection repeats until the short circuit is removed.

9/37

Waveforms and results AN2844

3 Waveforms and results

Figure 4 to Figure 15 show the main waveforms in steady-state conditions, and depict the

function of the converter with full loads or no loads and with var ious inpu t voltages. The
figures also show the turn-ON and turn-OFF behavior in various conditions. Of particular
interest is the behavior of the base current, where an initial high-peak pulse is needed to
minimize the effect of the dynamic saturation voltage.

Figure 4. V

= 125 VDC and maximum output power in steady-state conditions

IN

10/37

AN2844 Waveforms and results

Figure 5. VIN = 125 VDC and no output power (no load) in steady-state conditions

Figure 6. V

= 125 VDC and maximum output power in steady-state conditions

IN

- swi tch-ON hig hli g hte d

11/37

Waveforms and results AN2844

Figure 7. VIN = 125 VDC and maximum output power in steady-state conditions

- switch-OFF hig hl i ght e d

Figure 8. 620 VDC and maximum output power in steady-state conditions

12/37