HB LLC resonant converter for LCD TV & flat panels
Introduction
This note describes the performances of a 200 W reference board, with wide-range mains
operation and power-factor-correction (PFC). Its electrical specification is tailored to a
typical high-end application for LCD TV or monitor applications.
The main features of this design are the very low no-load input consumption (<0.5 W) and
the very high global efficiency, better than 87% at full load and nominal mains voltage (115 230 V
The circuit consists of three main blocks; the first is a front-end PFC pre-regulator based on
the L6563 PFC controller. The second stage is a multi-resonant half-bridge converter whose
control is implemented through the STMicroelectronics L6599 resonant controller. A further
auxiliary flyback converter based on the VIPer12A-E off-line primary switcher completes the
architecture. This third block is mainly intended for microprocessor supply and display power
management operations.
ac
).
Figure 1.L6599 and L6563 200W evaluation board (EVAL6599-200W)
Main characteristics and circuit descriptionAN2393
1 Main characteristics and circuit description
The main characteristics of the SMPS are listed below:
●Universal input mains range: 90to 264 V
●Output voltages:
–24 V@6 A continuous operation
–12 V@ 5 A continuous operation
–3.3 V@ 0.7 A continuous operation
–5 V@ 1 A continuous operation
●Mains harmonics: Compliance with EN61000-3-2 specifications
●St-by mains consumption: Typical 0.5 W @230 V
●
Overall efficiency: better than 88% at full load
●EMI: Compliance with EN55022-class B specifications
●Safety: Compliance with EN60950 specifications
●PCB single layer: 132x265 mm, mixed PTH/SMT technologies
The circuit consists of three stages. A front-end PFC pre-regulator implemented by the
controller L6563 (Figure 2), a half-bridge resonant DC/DC converter based on the resonant
controller L6599 (Figure 3) and a 7 W flyback converter intended for stand-by management
(Figure 4) utilizing the VIPer12A-E off-line primary switcher.
and frequencies between 45 and 65 Hz
ac
ac
The PFC stage delivers a stable 400 V
supply and provides for the reduction of the mains
DC
harmonics, in order to meet the requirements of the European norm EN61000-3-2 and the
JEIDA-MITI norm for Japan.
The PFC controller is the L6563 (U1), working in FOT (fixed off-time) mode and integrating
all functions needed to operate the PFC and interface the downstream resonant converter.
Note:The FOT control is implemented through components C15, C17, D5, Q3, R14, R17 and R29
(see AN1792 for a complete description of a FOT PFC pre-regulator).
The power stage of the PFC is a conventional boost converter, connected to the output of
the rectifier bridge through a differential mode filtering cell (C5, C6 and L3) for EMI
reduction. It includes a coil (L4), diode (D3) and two capacitors (C7 and C8).
The boost switch is represented by the Power MOSFET (Q2) which is directly driven by the
L6563 output drive thanks to the high current capability of the IC.
The divider (R30, R31 and R32) provides the L6563 (MULT Pin 3) with the information of the
instantaneous voltage that is used to modulate the boost current and to derive some further
information like the average value of the AC line used by the V
(voltage feed-forward)
FF
function. This function is used to keep the output voltage almost independent of the mains
one.
The first divider (R3, R6, R8, R10 and R11) is dedicated to detecting the output voltage
while the second divider (R5, R7, R9, R16 and R25) is used to protect the circuit in case of
voltage loop fail.
The second stage is an LLC resonant converter, with half bridge topology, working in ZVS
(zero voltage switching) mode.
The controller is the L6599 integrated circuit that incorporates the necessary functions to
drive properly the two half-bridge MOSFETs by a 50 percent fixed duty cycle with dead-time,
4/35
AN2393Main characteristics and circuit description
changing the frequency according to the feedback signal in order to regulate the output
voltages against load and input voltage variations.
The main features of the L6599 are a non-linear soft-start, a current protection mode used
to program the hiccup mode timing, a dedicated pin for sequencing or brown-out (LINE) and
a stand-by pin (STBY) for burst mode operation at light loads (not used in this design).
The transformer uses the magnetic integration approach, incorporating the resonant series
and shunt inductances. Thus, no additional external coils are needed for the resonance. The
transformer configuration chosen for the secondary winding is center-tap, and the output
rectifiers are Schottky type diodes, in order to limit the power dissipation. The feedback loop
is implemented by means of a classical configuration using a TL431 (U4) to adjust the
current in the optocoupler diode (U3). A weighted resistive divider (R53, R57, R58, R60 and
R61) is used to detect both output voltages in order to get a better overall voltage regulation.
The optocoupler transistor modulates the current from Pin 4, so the frequency will change
accordingly, thus achieving the output voltage regulation. Resistors R46 and R54 set the
maximum operating frequency.
In case of a short circuit, the current entering the primary winding is detected by the lossless
circuit (C34, C39, D11, D12, R43, and R45) and the resulting signal is fed into Pin 6.
In case of overload, the voltage on Pin 6 will overpass an internal threshold that triggers a
protection sequence via Pin 2, keeping the current flowing in the circuit at a safe level.
The third stage is a small flyback converter based on the VIPer12A-E, a current mode
controller with integrated Power MOSFET, capable of delivering (approximately) 7 W output
power on the output voltages (5 V and 3.3 V). The regulated output voltage is the 3.3 V
output and, also in this case, the feedback loop bases on the TL431 (U7) and optocoupler
(U6) to control the output voltage.
This converter is able to operate in the whole mains voltage range, even when the PFC
stage is not working. From the auxiliary winding on the primary side of the flyback
transformer (T2), a voltage Vs is available, intended to supply the other controllers (L6563
and L6599) in addition to the VIPer12A-E itself.
The PFC stage and the resonant converter can be switched on and off through the circuit
based mainly on components Q7, Q8, D22 and U8, which, depending on the level of the
signal ST-BY, supplies or removes the auxiliary voltage (V
) necessary to start-up the
AUX
controllers of the PFC and resonant stages. In this way, when the AC input voltage is applied
to the power supply, the small flyback converter switches on first; then, when the ST-BY
signal is low, the PFC pre-regulator becomes operative, and last the resonant converter can
deliver the output power to the load.
Note that if Pin 9 of Connector J3 is left floating (no signal ST-BY present), the PFC and
resonant converter will be not operating, and only +5V and +3.3V supplies are available on
the output. In order to enable the +24 V and +12 V outputs, Pin 9 of Connector J3 must be
pulled down to ground.
5/35
Main characteristics and circuit descriptionAN2393
Figure 2.PFC pre-regulator electrical diagram
Vdc
+400V
C9
2nF2-Y1
220uF/450V
C8
R2
NTC 2R5-S237
C7
470nF/630V
D3
STTH8R06
1-2
D1
1N5406
L4
PQ35-900uH
4-5
Q2
STP12NM50FP
D6
LL4148
R18
6R8
R24
0R68
R23
0R68
R22
0R68
R21
2R2
R19
1k
C18
330pF
Vrect
C6
680nF/630V
L3
DM-LSR-72uH-3A
C5
330nF/630V
Vaux
+
-
D2
D15XB60
~
~
C11
C4
Jumper
C3
L1
CM-TF2628V-5mH-3A
C2
R1
F1
6.3A/250V
1
J1
2nF2-Y2
680nF-X2
C10
2nF2-Y2
Jumper
330nF-X2
100nF-X2
1M5
2
CON2-IN
R4
47
C13
10uF/50V
C12
100nF
R6
680kR8680k
R3
680k
R5
Vdc
2M2
R11
R10
R7
2M2
R9
D5
C15
100pF
R14
1k5
R17
C17
15k
U1
L6563
100k
R13
56k
C14
100nF
C16
1uF
2M2
LL4148
15k
220pF
CSCS
PWM-Latch
R29
1k5
Q3
BC857C
R20
1k0
C21
2nF2
R28
GD
ZCD
VCC
RUN
GND
PWM-STOP
INV
COMP
MULTCSVFF
TBO
PFC-OK PWM-LATCH
R16
5k1
LINE
240k
R26
150k
C20
470nF
C19
10nF
R25
30k
C22
10nF
R32
10k
R31
620k
R30
620k
Vrect
6/35
AN2393Main characteristics and circuit description
Figure 3.Resonant converter electrical diagram
1234567
J2
+24V
8
CON8
C25
470uF/35V
C30
+12V
R61
R53
33k
R58
0R
R51
2k2
R57
C38
C35
2200uF/35V
470uF/25V
2200uF/25V
15k
3k9
R60
3k9
L5
2uH2
D8A
Vdc
Q5
R33
D7
D8B
STPS20H100CF
0R
R35
LL4148
STPS20H100CF
13
14
15
16
T1
T-RE S-ER4 9
4
2
C28
22nF/630V
Q6
STP14NK50Z
47
STP14NK50Z
R39
0R
R40
47
D9
LL4148
L6
C29
2200uF/35V
2uH2
D10A
STPS20L40CF
12
Vaux
R38
47
C37
2200uF/25V
D10B
STPS20L40CF
91011
R43
150
C34
220pF/630V
C32
100nF
C31
10uF/50V
D11
LL4148
D12
LL4148
R45
75R
C39
220nF
C41
10uF/50V
D13
C-12V
R59
R56
1k0
R52
5k6
U3A
SFH617A-2
27k
C44
47nF
U4
TL431
C27 100nF
NC
LVG
OUT
VCC
HVG
VBOOT
U2
L6599
CSS
DELAYCFRFMIN
STBY
ISEN
LINEGND
CC by rework
16k
DISPFC-STOP
C33
4nF7
R42
10
LINE
R36
0R
R34
2k7
C23
2uF2
C24
470nF
C26
270pF
R37
1M0
R41
C40
10nF
R47
10k
R46
5k6
PWM-Latch
U3B
SFH617A-2
R54
5k6
7/35
Main characteristics and circuit descriptionAN2393
Q10
BC847C
C54
100nF
U7
TL431
C53
2nF2
U6A
SFH617A-2
U6B
SFH617A-2
Vs
+24V
R67
1k0
SSFB
Vdd D
D
D
D
U5
VIPER-12A
R82
1k5
R79
1k0
D15
1N5822
D16
1N5821
D20
BAV103
C56
100nF
C45
1000uF/10V
C47
1000uF/10V
C50
10uF/50V
C46
100uF/10V
C49
100uF/10V
Q9
BC857C
C48
10uF/50V
R76
1k5
U8A
SFH617A-2
R75
0R
U8B
SFH617A-2
Vdc
R74
10k
R77
4k7
D19
C-30V
St-By
D18
B-10V
+400V
C57
1nF0
R69
0R
Vdc
C52
47nF
R68
22k
C51
100nF
R71
10k
1
2
4
5
9 - 10
8
7
6
R83
1M0
Q8
BC847C
R84
150k
R72
10k
C55
10uF/50V
R66
1k0
C58
10nF
T2
T-FLY-AUX-E20
+400V
Q11
BC557C
R70
22R
+5Vst-by
+3V3
+5Vst-by
Q7
BC547C
D22
C-15V
L7
33uH
L8
33uH
D17
LL4148
123456789
10
J3
CON10
+12V
D21
B-27V
D23
B-15V
R62
47
R64
1k6
Vs
Vaux
+5Vst-by
D14
PKC-136
St-By
R73
8k2
Figure 4.Auxiliary converter electrical diagram
8/35
AN2393Electrical test results
2 Electrical test results
2.1 Efficiency measurements
Ta bl e 1 and Tabl e 2 show the output voltage measurements at the nominal mains voltages
of 115 V
and at light load operation, the input power is measured using a Yokogawa WT-210 digital
power meter.
Particular attention has to be paid when measuring input power at full load in order to avoid
measurement errors due to the voltage drop on cables and connections. Therefore please
connect the WT210 voltmeter termination to the board input connector. For the same reason
please measure the output voltage at the output connector or use the remote sense option
of your active load for a correct output voltage measurement.
In Ta bl e 1 , Table 2 and Figure 5 the overall circuit efficiency is measured at each load
condition, at both nominal input mains voltages of 115 V
and 230 Vac. The values were
ac
measured after 30 minutes of warm-up at maximum load. The high efficiency of the PFC
pre-regulator working in FOT mode and the very high efficiency of the resonant stage
working in ZVS (i.e. with negligible switching losses), provides for an overall efficiency better
than 88%. This is a significant high value for a two-stage converter with two output voltages
delivering an output current in excess of 5 amps, especially at low input mains voltage
where the PFC conduction losses increase. Even at lower loads, the efficiency still remains
high.
The global efficiency at full load has been measured even at the limits of the input voltage
range, with good results:
●At V
●At V
= 90 Vac - full load, the efficiency is 86.88% (P
IN
= 264 Vac - full load, the efficiency is 90.90% (P
IN
= 208.8 W and PIN = 240.3 W)
OUT
= 208.7 W and PIN =
OUT
229.6 W)
9/35
Electrical test resultsAN2393
%
Also at light load, at an output power of about 10% of the maximum level, the overall
efficiency is very good, reaching a value better than 79% over the entire input mains voltage
range.
Figure 6 shows the efficiency measured at various input voltages versus output power.
Figure 5.Overall efficiency versus output power at nominal mains voltages
Figure 7 shows some waveforms during steady state operation of the resonant circuit at full
load. The Ch3 waveform is the half-bridge square voltage on Pin 14 of L6599, driving the
resonant circuit. In the picture it is not evident, but the switching frequency is normally
slightly modulated following the PFC pre-regulator 100-Hz ripple that is rejected by the
resonant control circuitry. The switching frequency has been selected approximately at
95-kHz in order to have a good trade off between transformer losses and dimensions.
The Ch4 waveform represents the transformer primary current flowing into the resonant
tank. As shown, it is almost sinusoidal because the operating frequency is close to the
resonance of the leakage inductance of the transformer and the resonant capacitor (C28).
In this condition, the circuit has a good margin for ZVS operation, providing good efficiency,
11/35
Electrical test resultsAN2393
while the almost sinusoidal current waveform just allows for an extremely low EMI
generation.
Figure 8 shows the same waveforms of previous figure, when both the outputs are not
loaded. This picture demonstrates the ability of the converter to operate down to zero load,
with the output voltages still within regulation. The resonant tank current has obviously a
triangular shape and represents the magnetizing current flowing into the transformer
primary side.
Figure 7.Resonant circuit primary side waveforms at full load
Ch3: half-bridge square voltage
Ch4: resonant tank current
Figure 8.Resonant circuit primary side waveforms at no-load condition
Ch1: +12V output voltage
Ch2: +24V output voltage
Ch3: half-bridge square voltage
Ch4: resonant tank current
In Figure 9 and Figure 10, waveforms relevant to the secondary side are represented: the
rectifiers reverse voltage is measured by CH1 (for both +24 V and +12 V outputs) and the
peak to peak value is indicated on the right side of the figure. It is a bit higher than the
theoretical value that would be 2(V
OUT+VF
): it is possible to observe a small ringing on the
bottom side of the waveform, responsible for this difference.
12/35
AN2393Electrical test results
Waveform CH3 shows the current flowing into one of the two output diodes for each output
voltage (respectively D8A and D10A). Also this current shape is almost a sine wave, whose
average value is one half the output current.
The ripple and noise on the output voltage is shown on CH2. Thanks to the advantages of
the resonant converter, the high frequency noise of the output voltages is less than 50 mV,
while the residual ripple at twice the mains frequency is lower than 75 mV at maximum load
and any line condition, as shown in Figure 11.
Figure 9.Resonant circuit secondary side
+24 V output waveforms:
Ch1: +24 V diode reverse voltage
Ch2: high freq. ripple on +24 V output voltage
Ch3: diode D8A current
waveforms: +24 V output
Figure 11. Low frequency (100 Hz) ripple voltage on the output voltages
Figure 10. Resonant circuit secondary side
waveforms: +12 V output
+12 V output waveforms:
Ch1: +12V diode reverse voltage
Ch2: high freq. ripple on +12 V output voltage
Ch3: diode D10A current
Ch1: 100 Hz ripple voltage on +12 V
Ch2: 100 Hz ripple voltage on +24 V
Figure 12 shows the dynamic behavior of the converterduring a load variation from 0 to
100% on one output, with the other output at maximum load. This figure also highlights the
induced effect of this load change on the PFC pre-regulator output voltage (+400 V on Ch3
track). Both the transitions (from 0 to 100% and from 100% to 0) are clean and do not show
any problem for the output voltage regulation.
13/35
Electrical test resultsAN2393
Thus, it is clear that the proposed architecture is really suitable for power supplies operating
with strong load variation without any problem related to the load regulation.
Figure 12. Load transition (0 - 100%) on +24 V
output voltage
Figure 13. Load transition (0 - 100%) on +12 V
output voltage
Dynamic load +24 V @0-6 A - 12 V @ max load (5 A)
Ch1: +24 V output voltage
Ch2: +12 V output current
Ch3: PFC output voltage (400 V)
Ch4: + 24 V output current
Dynamic load +12 V @0-5 A - 24 V @ max load (6 A)
Ch1: +24 V output voltage
Ch2: +12 V output current
Ch3: PFC output voltage (400 V)
Ch4: + 12 V output current
2.3 Stand-by and no load power consumption
The board is specifically designed for light load and zero load operation, as during Stand-by
or Power-off operation, when no power is requested from the +24 V and +12 V outputs.
Though the resonant converter can operate down to zero load, some tricks are required to
keep very low the input power drawn from the mains when the system is in this load
condition. Thus, when entering this power management mode, the ST-BY signal needs to be
set high (by the microcontroller of the system). This forces the PFC pre-regulator and the
resonant stage to switch off (because the supply voltage of the two control ICs is no longer
present (Figure 4) and only the auxiliary flyback converter continues working just to supply
the microprocessor circuitry.
Ta bl e 3 and Tabl e 4 show the measurements of the input power in several light load
conditions at 115 and 230 V
These tables show that at no load the input power is lower than 0.5 W.
ac
.
14/35
AN2393Electrical test results
Table 3.Stand-by consumption at V
+5 V(V) @load(A)+3.3 V(V) @load(A)P
5.08 - 0.0183.35 - 0.1020.430.863
5.04 - 0.0183.35 - 0.0790.360.751
4.98 - 0.0183.35 - 0.0460.240.582
4.92 - 0.0183.35 - 0.0230.170.445
4.47 - 0.0003.35 - 0.0000.000.221
Table 4.Stand-by consumption at VIN = 230 V
+5 V(V) @load(A)+3.3 V(V) @load(A)P
5.08 - 0.0183.35 - 0.1020.431.138
5.04 - 0.0183.35 - 0.0790.361.022
4.98 - 0.0183.35 - 0.0460.240.857
4.92 - 0.0183.35 - 0.0230.170.740
4.47 - 0.0003.35 - 0.0000.000.470
= 115 V
IN
ac
ac
(W)PIN (W)
OUT
(W)PIN (W)
OUT
2.4 Short-circuit protection
The L6599 is equipped with a current sensing input (pin #6, ISEN) and a dedicated overcurrent management system. The current flowing in the circuit is detected (through the not
dissipative sensing circuit already mentioned in Section 1 and the signal is fed into the ISEN
pin. It is internally connected to the input of a first comparator, referenced to 0.8 V, and to
that of a second comparator referenced to 1.5 V. If the voltage externally applied to the pin
exceeds 0.8 V, the first comparator is tripped causing an internal switch to be turned on
discharging the soft-start capacitor CSS.
For output short-circuits, this operation results in a nearly constant peak primary current.
Using the L6599, the designer can externally program the maximum time (t
converter is allowed to run overloaded or under short-circuit conditions. Overloads or shortcircuits lasting less than t
immunity to short duration phenomena. If, instead, t
(OLP) procedure is activated that shuts down the L6599 and, in case of continuous
overload/short circuit, results in continuous intermittent operation with a user-defined duty
cycle. This function is realized with the pin DELAY (#2), by means of a capacitor C24 and
the parallel resistor R37 connected to ground. As the voltage on the ISEN pin exceeds 0.8 V,
the first OCP comparator, in addition to discharging CSS, turns on an internal current
generator that via the DELAY pin charges C24. As the voltage on C24 reaches 3.5 V, the
L6599 stops switching and the PFC_STOP pin is pulled low. Also the internal generator is
turned off, so that C24 will now be slowly discharged by R37. The IC will restart when the
voltage on C24 becomes less than 0.3 V. Additionally, if the voltage on the ISEN pin reaches
1.5 V for any reason (e.g. transformer saturation), the second comparator will be triggered,
the L6599 will shutdown and the operation will be resumed after an on-off cycle.
will not cause any other action, hence providing the system with
SH
) that the
SH
is exceeded, an overload protection
SH
Figure 14 and Figure 15 illustrate the L6599 short-circuit protection sequence described
above. The on-off operation is controlled by the voltage on pin #2 (DELAY), providing for the
hiccup mode of the circuit. Thanks to this control pin, the designer can select the hiccup
mode timing and thus keep the average output current at a safe level. Please note on the left
15/35
Electrical test resultsAN2393
side of the figure the very low mean current flowing in the shorted output which is less than
0.3 A.
Figure 14. +24 V output short-circuit
waveforms
Figure 15. +12 V output short-circuit
waveforms
Short circuit on +24 V output voltage
Ch1: +24 V output voltage
Ch2: L6599 pin 6 (ISEN)
Ch3: L6599 pin 2 (DELAY)
Ch4: +24 V output current
2.5 Overvoltage protection
Both the PFC pre-regulator and the resonant converter are equipped with their own overvoltage protection circuit. The PFC controller L6563 is internally equipped with a dynamic
and a static overvoltage protection circuit sensing the error amplifier via the voltage divider
dedicated to the feedback loop to sense the PFC output voltage. If an internal threshold is
exceeded, the IC limits the PFC output voltage to a programmable, safe value.
Moreover, in the L6563 there is an additional protection against loop failures using an
additional divider (R5, R7, R9, R16 and R25) connected to a dedicated pin (PFC_OK, Pin 7)
protecting the circuit in case of loop failures or disconnection or deviation from the nominal
value of the feedback loop divider. Hence the PFC output voltage is always under control
and in case a fault condition is detected the PFC_OK circuitry will latch the L6563
operations and, by means of the PWM_LATCH pin (Pin 8) it will latch the L6599 as well via
the DIS pin (Pin 8).
The OVP circuit (seeFigure 4) for the output voltages of the resonant converter uses two
zener diodes (D21 and D23) to sense the +24 V and+12 V. If one of the output voltages
exceeds the threshold imposed by these zener diodes plus the V
Q9 starts conducting and the optocoupler U8 opens Q7, so that the V
the controller ICs L6563 and L6599 is no longer available. This state is latched until a mains
voltage recycle occurs.
Short circuit on +12 V output voltage
Ch1: +12 V output voltage
Ch2: L6599 pin 6 (ISEN)
Ch3: L6599 pin 2 (DELAY)
Ch4: +12 V output current
of Q10, the transistor
BE
supply voltage of
AUX
16/35
AN2393Thermal tests
3 Thermal tests
In order to check the design reliability, a thermal mapping by means of an IR Camera was
performed. Figure 16 and Figure 17 show the thermal measurements of the board,
component side, at nominal input voltage. The correlation between measurement points and
components is indicated for both diagrams.
Figure 16. Thermal map @115 V
Figure 17. Thermal map at 230 V
- full load
ac
- full load
ac
17/35
Thermal testsAN2393
Table 5.Key components temperature at 115 Vac - full load
Ambient temperature: 25° C
ItemTemp (°C)
D244.9
Q253.7
D350.3
L147.0
L346.0
L4 (Fe)45.8
L4 (Cu)49.2
C837.3
R278.0
Q540.2
Q646.7
D8A56.2
D8B56.7
D10A42.1
D10B42.7
C2945.1
C3046.1
C3742.0
C3841.6
L571.2
L656.0
T151.7
T156.8
U581.4
D1474.2
D1557.6
D1655.3
T256.4
18/35
AN2393Thermal tests
Table 6.Key components temperature at 230 V
Ambient temperature: 25° C
ItemTemp (°C)
D237.1
Q246.6
D344.0
L133.6
L334.9
L4 (Fe)39.1
L4 (Cu)41.2
C837.1
R265.8
Q538.3
Q643.7
D8A56.4
D8B55.6
D10A42.1
- full load
AC
D10B43.8
C2948.2
C3047.4
C3744.3
C3844.5
L573.6
L657.3
T1 (Fe)51.3
T1 (Cu)58.8
U581.8
D1474.4
D1559.4
D1656.3
T256.8
All other board components work within the temperature limits, assuring a reliable long term
operation of the power supply.
Note that the temperatures of L4 and T1 have been measured both on the ferrite core (Fe)
and on the copper (Cu).
19/35
Conducted emission pre-compliance testAN2393
4 Conducted emission pre-compliance test
The limits indicated on both diagrams at 115 Vac and 230 Vac comply with EN55022 Class-B
specifications. The measurements have been taken in Quasi Peak detection mode.
Figure 18. CE quasi peak measurement at 115 V
Figure 19. CE quasi peak measurement at 230 V
and full load
ac
and full load
ac
20/35
AN2393Bill of materials
5 Bill of materials
Table 7.Bill of materials
ItemPart type/valueDescriptionSupplier
C2100 nF-X2275 V
C3330 nF-X2275 V
C4680 nF-X2275 V
X2 Safety Capacitor MKP R46Arcotronics
ac
X2 Safety Capacitor MKP R46Arcotronics
ac
X2 Safety Capacitor MKP R46Arcotronics
ac
C5330 nF/630 VPolypropylene Capacitor High Ripple MKP R71Arcotronics - Epcos
C6680 nF/630 VPolypropylene Capacitor High Ripple MKP R71Arcotronics - Epcos
C7470 nF/630 VPolypropylene Capacitor High Ripple MKP R71Arcotronics - Epcos
C8220 µF/450 VAluminium ELCAP USC Series 85 DEG SNAP-INRubycon
C92nF2-Y1400 V
C102nF2-Y1250 V
C112nF2-Y1250 V
Y1 Safety Ceramic Disk CapacitorMurata
ac
Y1 Safety Ceramic Disk CapacitorMurata
ac
Y1 Safety Ceramic Disk CapacitorMurata
ac
C12100 nF50 V 1206 SMD Cercap General PurposeBC Components
C1310 µF/50 VAluminium ELCAP General Purpose 85 DEGRubycon
C14100 nF50 V 1206 SMD Cercap General PurposeBC Components
C15100 pF100 V 0805 SMD Cercap General PurposeBC Components
C161 µF25 V 1206 SMD Cercap General PurposeBC Components
C17220 pF100 V 0805 SMD Cercap General PurposeBC Components
C18330 pF100 V 0805 SMD Cercap General PurposeBC Components
C1910 nF100 V 0805 SMD Cercap General PurposeBC Components
C20470 nF50 V 1206 SMD Cercap General PurposeBC Components
C212nF2100 V 1206 SMD Cercap General PurposeBC Components
C2210 nF100 V 0805 SMD Cercap General PurposeBC Components
C232 µF225 V 1206 SMD Cercap General PurposeBC Components
C24470 nF25 V 1206 SMD Cercap General PurposeBC Components
C25470 µF/35 VAluminium ELCAP YXF Series 105 DEGRubycon
C26270 pF100 V 0805 SMD Cercap General PurposeBC Components
C27100 nF50 V 1206 SMD Cercap General PurposeBC Components
C2822 nF/630 V/400 V
Polypropylene Capacitor High Ripple PHE450RIFA-EVOX
ac
C292200 µF/35 VAluminium ELCAP YXF Series 105 DEGRubycon
C302200 µF/35 VAluminium ELCAP YXF Series 105 DEGRubycon
C3110 µF/50 VAluminium ELCAP General Purpose 85 DEGRubycon
C32100 nF50 V 1206 SMD Cercap General PurposeBC Components
C334 nF7100 V 1206 SMD Cercap General PurposeBC Components
21/35
Bill of materialsAN2393
Table 7.Bill of materials (continued)
ItemPart type/valueDescriptionSupplier
C34220 pF/630 VPolypropylene Capacitor High Ripple PFRRIFA-EVOX
C35470 µF/25 VAluminium ELCAP YXF Series 105 DEGRubycon
C372200 µF/25 VAluminium ELCAP YXF Series 105 DEGRubycon
C382200 µF/25 VAluminium ELCAP YXF Series 105 DEGRubycon
C39220 nF50 V 1206 SMD Cercap General PurposeBC Components
C4010 nF100 V 1206 SMD Cercap General PurposeBC Components
C4110 µF/50 VAluminium ELCAP General Purpose 85 DEGRubycon
C4447 nF100 V 1206 SMD Cercap General PurposeBC Components
C451000 µF/10 VAluminium ELCAP YXF Series 105 DEGRubycon
C46100 µF/10 VAluminium ELCAP YXF Series 105 DEGRubycon
C471000 µF/10 VAluminium ELCAP YXF Series 105 DEGRubycon
C4810 µF/50 VAluminium ELCAP General Purpose 85 DEGRubycon
C49100 µF/10 VAluminium ELCAP YXF Series 105 DEGRubycon
C5010 µF/50 VAluminium ELCAP General Purpose 85 DEGRubycon
C51100 nF100 V 0805 SMD Cercap General PurposeBC Components
C5247 nF100 V 0805 SMD Cercap General PurposeBC Components
C532 nF2100 V 0805 SMD Cercap General PurposeBC Components
C54100 nF50 V 1206 SMD Cercap General PurposeBC Components
C5510 µF/50 VAluminium ELCAP General Purpose 85 DEGRubycon
C56100 nF50 V 1206 SMD Cercap General PurposeBC Components
C571nF0100 V 0805 SMD Cercap General PurposeBC Components
C5810 nF50 V X7R Standard Ceramic CapacitorBC Components
D11N5406General Purpose RectifierVishay
D2D15XB60Single Phase Bridge RectifierShindengen
D3STTH8R06TO220FP Ultrafast High Voltage RectifierSTMicroelectronics
D5LL4148MINIMELF Fast Switching DiodeVishay
D6LL4148MINIMELF Fast Switching DiodeVishay
D7LL4148MINIMELF Fast Switching DiodeVishay
D8A-B STPS20H100CFTO220FP Power Schottky RectifierSTMicroelectronics
D9LL4148MINIMELF Fast Switching DiodeVishay
D10A-BSTPS20L40CFTO220FP Power Schottky RectifierSTMicroelectronics
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