ST AN2123 APPLICATION NOTE

AN2123

Application Note

TD351 Advanced IGBT Driver

Principles of operation and application

1 Introduction

The TD351 is an advanced IGBT driver with integrated control and protection functions. It is a simplified version of the TD350, available in an SO8 or DIP8 package. The TD35x family (including the TD350, TD351 and TD352) provides a wide range of drivers specially adapted to drive 1200 V IGBTs with current ratings of 15 to 75 A in Econopak-like modules (see Figure 2).

The main features of the TD351 are:

-1 A sink/0.75 A source peak output current minimum over the full temperature range (-20°C to 125°C),

-active Miller clamp function to reduce the risk of induced turn-on in high dV/dt conditions, and in most cases, without requiring a negative gate drive,

-optional 2-step turn-off sequence to reduce over-voltage in case of an over-current or a shortcircuit situation; a feature that protects the IGBT and avoids RBSOA problems,

-input stage compatible with both an optocoupler and a pulse transformer.

Applications include three-phase full-bridge inverters such as in motor speed control and UPS systems

(see Figure 1).
Figure 1. TD351 in 3-phase inverter application (1200 V IGBTs)
				V+ DCbus
High-side power
supply	TD351	TD351	TD351
or
Bootstrap
Circuitry
		Phase 1	Phase 2	Phase 3
Low-side power	TD351	TD351	TD351
supply
				V- DCbus
AN2123/0205		Revision 1		1/15

TD351 application example

AN2123

Figure 2. IGBT modules

2 TD351 application example

A TD351 application example is shown in Figure 3. In this example the device is supplied by a +16V isolated voltage source. An optocoupler is used for input signal galvanic isolation. The IGBT is driven by 44Ω for turn-on and 22Ω for turn-off thanks to the use of two gate resistors and one diode: sink and source currents can therefore be tuned independently to help and solve EMI issues. Power switch drivers are used in very noisy environment and decoupling of the supplies should be cared. In the application example the decoupling is made by a 100nF ceramic capacitor located as close as possible to the TD351 in parallel with a bigger electrolytic capacitor.

Figure 3. TD351 application example

						16V
			TD351		100nF
		1	IN	VH	8
		2	VREF	OUT	7
4K7		3	CD	VL	22Ω	22Ω
	16K				6
100pF	10nF	4	LVOFF	CLAMP	5
	470pF
		VH
	10K
	11V
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Input stage

3 Input stage

The TD351 is compatible with the use of both pulse transformers or optocouplers. The schematics shown in Figure 4 can be considered as example of use with both solutions.

When using a pulse transformer, a 2.5 V reference point can be built from the 5 V VREF pin with a resistor bridge. The capacitor between the Vref and the bridge middle point provides decoupling of the 2.5 V reference, and also insures a high level on IN input at power-up, in order to start the TD351 in the OFF state.

When using an optocoupler, the IN pin can be pulled-up to Vref. The pull up resistor is to be chosen between 5 kΩ to 20 kΩ depending on the characteristics of the optocoupler. An optional filtering capacitor can be added in case of a highly noisy environment, although the TD351 already includes filtering on input signals and rejects signals smaller than 135 ns (tonmin specification).

Waveforms from the pulse transformer must comply with the tonmin and Vton/Vtoff specifications (see Figure 5). To turn TD351 output on, the input signal must be lower than 0.8 V for 220 ns minimum.

Conversely, the input signal must be higher that 4.2 V for 220 ns minimum in order to turn off TD351 output. A pulse width of about 500 ns at the threshold levels is recommended. In all cases, input signal at the IN pin must be between 0 and 5 V.

Figure 4. Application schematic (pulse transformer at left; optocoupler at right)
Pulse transformer		Optocoupler
	TD351			TD351
1	IN		1	IN
10K	VREF		2	VREF
2
		4K7
10K
		100pF	10nF
10nF
10K
Figure 5. Typical input signal waveforms with pulse transformer (left) or optocoupler (right)

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Output stage

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4 Output stage

The output stage is able to sink/source about 1.7 A / 1.3 A typical at 25°C with a voltage drop VOL/VOH of 6 V (see Figure 6). The minimum sink/source currents over the full temperature range (-20°C/+125°C) are 1 A sink and 0.75 A source. VOL and VOH voltage drops at 0.5 A are guaranteed to 2.5 V and 4 V maximum respectively, over the temperature range (see Figure 7). This current capability sets the limit of IGBT driving, and the IGBT gate resistor should not be lower than about 15Ω.

Figure 6. Typical Output stage current capability at 25°C (VH=16V)

OUT source current versus voltage (turn-on)

OUT sink current versus voltage (turn-off)

	2.5					2.5
	2.0					2.0
Iout (A)	1.5				Iout (A)	1.5
	1.0					1.0
	0.5					0.5
	0.0					0.0
	0	5	Vout (V)	10	15

0

5

10

15

Vout (V)

Figure 7. Typical VOL and VOH voltage variation with temperature

		High level output voltage vs. Temperature								Low level output voltage vs. Temperature
	4.0								3.0
	3.0
(V)					Iosource=500mA				2.0
									(V)
												Iosink=500mA
VH-VOH	2.0								VOL-VL
									1.0
	1.0			Iosource=20mA
														Iosink=20mA
	0.0								0.0
	-50	-25	0	25	50	75	100	125
									-50	-25	0	25	50	75	100	125
				Temp (°C)
												Temp (°C)

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AN2123

Active Miller clamp

5 Active Miller clamp

The TD351 offers an alternative solution to the problem of Miller current in IGBT switching applications.

Traditional solutions to the Miller current problem are:

to drive the IGBT gate to a negative voltage in OFF-state in order to increase the safety margin

or, to implement an additional capacitor between the IGBT gate and collector as described in the lefthand schematic in Figure 8)

The solution proposed by the TD351 uses a dedicated CLAMP pin to control the Miller current. When the IGBT is off, a low impedance path is established between IGBT gate and emitter to carry the Miller current, and the voltage spike on the IGBT gate is greatly reduced (see the right-hand schematic in Figure 8). The CLAMP switch is open when the input is activated and is closed when the actual gate voltage goes close to the ground level. In this way, the CLAMP function doesn’t affect the turn-off characteristics, but simply keeps the gate at a low level during the entire off-time.

The main benefit is that negative supply voltage can be avoided in most cases, allowing for the use of a bootstrap technique for the high-side driver supply, and a consistent cost reduction for the application.

In addition, the use of the active Miller clamp feature avoids the need to implement any additional capacitors between the IGBT gate and the collector. Such capacitors would negatively affect the ability of the driver to control turn-on and turn-off.

Figure 8. Active Miller Clamp: principles of operation

High-side

driver

Miller current

Low-side

driver

10R

10nF

optional capacitor

implemented to

reduce voltage spike

High-side

TD351

	Miller current
	Low-side
high dV/dt !	TD351
	10R
	active clamp
	no need for
	additional
	capacitor

voltage spike on IGBT gate !

high dV/dt !

reduced voltage spike

The test results shown in Figure 9 prove how the active Miller clamp results in a consistent reduction of the voltage spike on IGBT gate.

The left-hand waveform shows the result of a 400 V switching with a 10 nF additional Gate to Emitter capacitor to control the voltage spike on gate.

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