ST AN1453 APPLICATION NOTE

AN1453

®

NEW FAMILY OF 150V POWER SCHOTTKY

INTRODUCTION

Nowadays, the Switch Mode Power Supply
(SMPS) is becoming more widespread as a result
of computer, telecom and consumer applications.

The constant increase in services (more
peripherals) and performance, which offers us
these applications, tends to move conversion
systems towards higher output power.

In addition to these developments dictated by the
market, SMPS manufacturers are in competition,
their battlefield being the criteria of power density,
efficiency, reliability andcost, this last being factor
very critical.

Today, SMPS designers of 12V-24V output have
practically the choice between a 100V Schottky or
a 200V bipolar diode.
The availability of an intermediate voltage has
become necessary to gain in design optimization.

APPLICATION NOTE

By F. GAUTIER

In the following examples, the conduction losses
between a 150V Schottky and a 200V bipolar
diode in a Flybackand aForward converter willbe
compared.

The conduction losses in the diode are calculated
from the classical formula:

P=VI+RI

cond T0 F(AV) d IF(RMS)

V :threshold voltage with V = V +R .I

t0 F(@IF) T0 d F

R : dynamic resistance with R = V / I

ddFF

where V

and Rdare calculated from the current

T0

⋅⋅

range of current view by the diode (Fig. 1), for
better accuracy.

Figure 1 shows also, the typical current through
the rectification diode and thecorresponding I

2

and

I

IF(RMS)

:

Fig. 1: Typical current through a rectification diode

2

∆∆

F(AV)

This is why STMicroelectronics is introducing a
new family of 150V POWER SCHOTTKY diodes,
intendedfor12Vandmoresecondaryrectification,
in applications such as desktops, file servers or
adaptors for notebook.

Consequently, this application note will underline
the advantages of a 150V Schottky technology
compared to a 200V ultra fast diode.

In order to do this, the example of a Flyback
converter will be used, and thestatic and dynamic
parameters of the 150V Schottky will be detailed,
as well as their influence in this converter.

1.CONDUCTIONLOSSES&EFFICIENCY GAIN

Schottky diodes are mainly used for output
rectification. In a typical SMPS working with a
switching frequency lower than 100kHz,
conduction losses are generally the main lossesin
the diode. They are directly linked to the curve of
forward voltage (V

) versus forward current (IF),

F

and obviously the best gain in efficiency will be
obtained with the lowest V

July 2001

.

F

I

D

I

ma x

I

min

0

α

I=(I I)

F(AV)

2

I(III

F(RMS)

R=

d

ID

+

max min

2

α

2

ID

=++

VV

max min2m

3

−

F(@Imax) F(@Imin)

II

−

max min

αI.T

D

I

⋅

)

ax min

VV

=

T0 F

NB:

-In the datasheet, the V
values given for I

and2IFat 125°C.

F

-In discontinuous mode I

and Rdare maximum

T0

=0.

min

T

(@imax) d max

RI−⋅

1/9

t

APPLICATION NOTE

1.1. Example 1: FLYBACK

The first example is a 24V/48W Flyback converter
working in continuous mode (Vmains=90V) with
the following conditions:

0.4, I 6.66A, I 3.33A, I 2A

== = =

α

ID max min out

ID ID

Fig. 2: Rectification diode in a Flyback converter

I

V

I

in

o u t

D

Calculations per diode give:

I = 1A and I = 1.6A

F(AV)per diode F(RMS)

per diode

We can now calculate the efficiency gain (∆η(%)=

- η) for this Flyback converter which has a

η

ref

reference (ref) efficiency of 85% with
STPR1020CT:

Fig. 3: Example of efficiency gain in Flyback
converter

V

P
V

out
out

=48W
=24V

T0

typ(V)

1.5A, 3A,
125°C

R

mΩ

d

P

cond

(W)

∆P

(W)

η=85

%

∆η%

STPR102CT
2x5A / 200V

0.58 46.5 1.4 0 (ref) 0 (ref)

PN diode

1.2. Example 2: FORWARD

In the following example,the conduction losses in
a 12V/96W Forward converter are simulated:

Fig. 4: Rectification diode in a Forward converter

I

I

o u

L

V

D1

in

D2

α

0.3, I 9A, I 7A, I 8A====

D1 Lmax Lmin out

Calculations per diode give:

==

I 2.4A, I 4.39A

F(AV)D1 F(RMS)D1

=

I 5.6A, I

F(AV)D2 F(R

MS)D2

The difference of efficiency between a
STPR1620CT (2x8A, 200V Ultrafast) and a
STPS16150CT (2x8A, 150V Schottky) for a 12V
output, are given in table Fig. 5:

Fig. 5: Example of efficiency gain in Flyback
converter

V

R

T0

d

mΩ

P
V

out
out

=96W
=12V

typ(V)

7A, 9A,

125°C

STPR1620CT

0.8 20 6.48 Ref Ref

P

(W)

cond

6.71A=

η=85

∆P

%

(W)

∆η%

STPR162CT
2x8A / 200V
PN diode

STPS10150CT
2x5A / 150V
Schottky diode

STPS16150CT
2x8A / 150V
Schottky diode

2/9

0.54 46.5 1.32 -0.08 +0.12

0.50 43 1.22 -0.18 +0.27

0.47 40 1.14 -0.26 +0.39

STPS16150CT

0.68 20 5.60 -0.95 +0.72

These two examples show that whatever the type
of converter, a significant efficiency gain can be
achievedonly by replacing a200V bipolar diode by
a 150V Schottky.

APPLICATION NOTE

2. REVERSE LOSSES AND T

JMAX

2.1. Reverse losses: Prev

The reverse losses can be determined by:

=⋅⋅−α)

PVI(1

rev R R

with:

): duty cycle when the reverse voltage (VR)is

(1-

applied
IR: leakage current versus VRand operating

junction temperature (T

V

: reapplied voltage accross the diode

R

)

j

Fig. 6 shows an example of reverse losses in a
Flyback converter with the following conditions:

()1−= = = °α0.4, V 80V, T 125 C

Rj

Fig. 6: Example of reverse losses in a Flyback
converter

STPS10150CT

per diode

I

Rtyp

100V, 125°C

130µA 4.2mW

P

rev

per diode

Thus, the reverse losses are very low due to the
low value of the leakage current.

The following paragraph will show that due to
these low values of reverse current, the thermal
runaway limit is only reached for high junction
temperature.

2.2. T

before thermal instability is reached

jmax

Remembering that the stability criterion is given
by:

dP

rev

<

dT1R

j th(j a)

−

with:

=−α)

P V .I .(1

rev R R(VR,Tjmax)

The above formulae give the critical value of the
leakage currentbefore the thermal runaway limit is
reached:

I

R(VR,Tjmax)

=

Vc.R

R th(j a)

The evolution of the leakagecurrent versus T

is given by:

V

R

IIexp

=

R(V ,Tj) R(V ,125)

RR

1

⋅⋅−

1()α

−

−

c(Tj 125)

and

j

From these physical laws, it can be deduced that:

Example:
Flyback converter with 2 diodes in parallel

()1−= = =α0.4, c 0.069, V 80V

R 2.4 C / W, R 7.6 C / W

th(j c)total th(c a)−−

Fig. 7: Example of T

For a dual
diode

STPS10150CT

=° =°

with STPS10150CT

jmax

I

Rmax

(80V,125°C)

1.3mA 45.28mA 176.5°C

R

I

R(VR,Tjmax)Tjmax

This example shows that in a typicalapplication, a
150V Schottky can be used up to 175°C.
STMicroelectronics specifies in the datasheet

at 175°C.

T

jmax

3. SWITCHING BEHAVIOUR

3.1. Turn-on behaviour

The behaviour at turn-on is characterized bya low
value of peak forward voltage (V
reverse recovery time (t

) (Fig. 8).

fr

) and forward

FP

Fig. 8: VFPand tfrfor STPS16150CT

=16A

I

F

/dt=100A/µs

dI

F

=25°C

T

j

t

fr

(ns)

V

(V)

FP

Per diode

STPS16150CT

These values depends mainly on the dI

100 2.2

/dt. The

F

switching losses at turn-on are always negligible.

3.2. Turn-off behaviour

The turn-off behaviour isa transitoryphenomenon
(ns), but repetitive depending on the switching
frequency. It is a source of spike voltage, noise
and for high switching frequency, ofnon-negligible
switching losses.

Inorder to illustrate thisphenomenon, the example
of a Flyback converter will be used once again.

The difference in behaviour between a 150V
Schottky and 200V bipolar diode will be compared
for the three following points: spike voltage, EMC
and switching losses.

T 125

=+⋅

jmax

1

In

c

I

R (V ,125 C)datashee

max R

I

R(V ,Tjmax)

R

° t

3/9

APPLICATION NOTE

3.2.1. Difference of spike voltage between a
150V Schottky and 200V PN diode

In a Flyback converter, the reverse voltage (V
used in §2) across the diode will be maximum, for
the maximum mains voltage (V

n

VV

=⋅+

R INmax

s

n

p

V

out

In addition to this nominal reverse voltage (V

):

INmax

(cf Fig. 9)

R

generallyan overvoltage spike at the turn-off ofthe
diode is observed (Fig. 9). It can be shown that
with a conventional bipolar diode, this spike is
more important for a Flyback converter working in
continuous mode than in a discontinuous mode.

In the case of a high spike voltage, the Maximum
Repetive Reverse Voltage (V
to be oversized, compared with the real need (V

) of the diode has

RRM

R

defined in Fig. 9.
To limit this peak and to preserve a "guard band"

with the V

(in order to avoid reaching the

RRM

breakdown voltage), the designer places a
snubber circuit (R

) in parallel with the diode.

S,CS

Generally, the "guard band" is such that the
maximum voltage reapplied to the diode does not
exceed 80% of the V

RRM

.

3.2.1.1 Turn-off behaviour for a PN diode

In the datasheet are specified the main turn-off
parameters (Q

R

rr,IRM,trr

…). These parameters are

represented in Fig. 10:

Fig. 10: Key parameters at turn-off for a bipolar
diode without snubber

Q = Q + Q
t = t + t

I R

Q

rrb

rrab

S =

t

V

R

V

Rma x

),

I

V

)

d /dt

I F

Q

rra

I

RM

t

atb

d /dt

The following oscillogram shows the turn-off
behaviour for a bipolar diode (STPR1620CT) with
snubber and without snubber, in a 24V/45W
Flyback working in continuous mode.

rr rra rrb

t

b

t

a

Fig.9: Spike voltageacross the rectificationdiode

R

C

s

s

Turn-off diode

V

V

npn

in

V

P

D

s

I

D

V

s

V=V

RIN+Vout

I

D

V

o u t

V

D

n

s

n

p

V

Rmax

I

RM

V

RRM

Thisspike voltage isdue to theleakage inductance
of the transformer (L

) and to the nature of the

f

recovery charge of the diode,which itselfdepends
on the diode technology: bipolar diode or Schottky
diode.

To observe the phenomenon correctly, it is
necessary to compensate the delay time between
the voltage and the current, (by temporal shift) due
to the measuring equipment Fig. 11.

Fig. 11: Switching behaviour of a 200V bipolar
diode

dI /dt=130A/µs

F

I =4A

RM

delay time

t0

t0

Tj=100°C

dI /dt=600A/µs

R

Compensative curve

V =250V

Rmax

dV/dt

V =90V

Rmax

I

V =42V

R

I

R =22ohms

s

V

V

20V/div
2A/div
50ns/div

V

C =2.2nF

s

V

I

I

4/9

APPLICATION NOTE

Without a snubber, in this example the diode is
repeatedly in conduction because the oscillationis
very strong. Furthermore, the voltage is close to
the breakdown voltage. This means that the
systemis no longer reliableand a snubbercircuit is
necessary.

Onthese 2 oscillograms, we cansee that the value
of the maximum reverse current (I

) is defined

RM

when the reverse voltage rises (typical behaviour
of a bipolar diode). At this time the voltage is not
fixed by the diode.

The curve Q

versus dIF/dt and Tjis given in

rr,IRR

the datasheet. For example in Fig. 12, the
evolution of I

versus dIF/dt for a STPR1620CT

RM

can be observed.

Fig. 12: Peak reverse recovery current versus

/dt (per diode)

dI

F

IRM(A)

20

IF=IF(av) 90% confidence

10

Tj=125°C

STPR1620CG/CT

Fig.13: Equivalent modelat t0fora bipolar diode

R

C

s

s

L

f

I=I

V

n

p

L

P

D

n

s

L

s

V

s

V

o u t

V=V+V

R S o u t

L f RM

C

s

V

R

C

Qrrb

s

C

j

Where:

n

V

: secondary voltage

s

:leakageinductance of the transformer

L

f

C

: junction capacitance

j

C

: equivalent capacitance modeling the

Qrrb

V

=⋅

S

s

V

IN

n

p

reversecharge, necessary forthe establishment of
the potential barrier, which supports the reverse
voltage.

: output voltage

V

out

D

1

10 20 50 100 200 500

It can be also noticed, that the parameter I

dIF/dt(A/µs)

RM

significantly increases with the temperature.
In continuous mode the dI

/dt (few hundred A/µs)

F

is fixed by theleakage inductanceand thereverse
voltage (V

):

R

dIdtV

FR

with V

==⋅+

L

f

n

R

n

s

VV

IN out

p

It is many time higher than in discontinuous mode
(lower than 1A/µs):

dI

F out

=

dtVLL

: Secondary inductance)

(L

S

Sf

withL L

+

〉〉

Sf

Thus, with this curve we can see that, in
continuous mode (high dI

/dt), the bipolar diode

F

must evacuate a non-negligible charge, which
means a higher I

. This is verified onoscillogram

RM

Fig. 11.

With this value of I

, an equivalent model at t

RM

with a snubber circuit can be established:

With the following initial conditions at t=t

I I and V 0

=≈

LRM D

f bipolar

The equivalent schematic can be used to define

=

VV

DR

max

NB:

1) Without snubber, there is a L

(C=C

+C

j

) which lead to a second order

Qrrb

differential equation:

2

dV

dt

2

C

+⋅+⋅= = ⋅ωω ω

0

2

2

V V 0 and 1/ L C

CR f

0

with initial conditions at t=t0:

===

I I and V V 0

LRM C D

f0

In this equation, an approximation is made with C
constant, because in reality C

and C

j

the voltage applied.
The solution of the differential equation gives us:



VVVVI

==+ + ⋅

RDRR

max

Therefore we can see that the V
leakage inductance (L

). Thus,

0

(I

RM

temperature.

V

R

max

is very dependent on the

2



RM





) and on recovery charge

f

Rmax

:

0

, C circuit

f

2
0

vary with

Qrrb

2



L

f





C



depends the

5/9

APPLICATION NOTE

If C is a low value of capacitance, the expression
leads to a first order differential equation:

==+⋅

VVVLdI/dt

RDRfR

max

with:

dI /dt I /t and t Q / I /2
dI / dt I / 2 Q

==

R RM b b rrb RM

2

=⋅

RRM

()

b

rr

()

If the diode is of a snap-off type, we have
t0,Q 0

→→

b rrb

consequently
to reach the breakdown voltage of the diode. (It is

and dI
is considerable. There is a risk

V

R

max

/dt is very high,

R

not guaranteed by the manufacturer)

2) With a snubber, if it is supposed that

>> +

CCC

S j Qrrb

diode), we have a simple R

(more true for an ultrafast PN

s,Lf,CS

circuit to define

by the second order differential equation:

2

dV

C

SS

++⋅=ωωω

2

dt

2m

dV

0

C

dt

V.V

02R0

2

R

where:

R2C

m: the absorption coefficient

:natural frequency

0

ω

0

=

LC

m

=

1

⋅

fS

SS

L

f

There are 3 possible cases:

-m>1 behaviour withoutoscillation (over damping)

Fig. 14: Turn-off for a perfect Schottky diode
without snubber

I

D

V

Rma x

t

0

V

D

In this perfect case at t=t

I I 0 and V 0

== ≈

LRM D

f

, we have:

0

t

V

R

Without a snubber circuit, the new solution of the
differential equation gives:

VV 2V

==⋅

DR R

max

Unfortunately, it is difficult to realize a perfect high
voltage Schottky. The reason is the presence of a
parasitic bipolardiode in parallel with the Schottky.
If it is polarized, the recovery charge is added at
the turn-off.The phenomenon begins to appear for
a 100V technology.

In the same conditions as before, the
STPS16150CT is used:

- m<1 behaviour with oscillation damped (under

damping), where the frequency oscillation can be
determined by:

ωω

r

0

2

1m=−

-m=1 limit of behaviour without oscillation (critical

damping)

Like this, in this case,it ispossible tosuppress the
oscillation across the diode with m>1.

3.2.1.2 Turn-off behaviour for a Schottky diode

For an ideal Schottky diode, there is no recovery
charge (Q

=0). Therefore, it is said that the diode

rr

has a capacitive type recovery Fig. 14.

6/4

Fig. 15: Switching behaviour of a STPS16150CT

dI /dt=130A/µs

F

I =1.6A

RM

V =58V

Rmax

t0

t0

Tcase=100°C

dI /dt=250A/µs

R

V =120V

Rmax

dV/dt

V =42V

R

I

I

R =22ohms

s

V

V

C =2.2nF

s

20V/div
2A/div
50ns/div

I

V

V

I

APPLICATION NOTE

It can be observed that this is not an ideal
Schottky. In fact, when the voltage rises at , we
have a value of I

. The charge Q

RM

is not easily

rrB

identifiable because it is embedded in the
capacitive current.

However, the slope dI

/dt can be observed.

R

Unlike a PN diode, we can see that with the 150V
Schottky the maximum reverse voltage (V
and the maximum reverse current (I

RM

Rmax

) are

distinctly lower.

The equivalent model at t

for a STPR1620CT and

0

a STPS16150CT is the same, with thelower initial
conditions at t

In Fig. 16, we can see the curve C

:

0

I I and V 0

=≈

LRM D

f schottky

versus VRfor

j

the 200V bipolar and 150V Schottky diode.
Whatever the reverse voltage, the junction
capacitance of the 150V Schottkyis alwayshigher
than for a PN diode. This justifies, the lower
observed with the Schottky diode.

Fig. 16: Cjversus V

F=1MHz,Tj=125°C

300.0

250.0

R

STPR1620CT
STPS16150CT

Thus, a different efficiency of the snubber circuit
with a bipolar diode and a Schottky diode is
observed.In most cases, we cansay that the 150V
Schottky behaves better at turn-off, due to its
larger capacity and its softness recovery.

Model showed in Fig. 13can beused todefine the
snubber circuit.

NB:

)

In the case of a Forward converter with multiple
outputs (12V, 5V, 3.3V…) and cross regulation
with coupled inductor, the poor behaviour at the
turn-off with a bipolar diode on 12V output, will be
reflected on the other coupled outputs (that means
an overvoltage on rectification diode of 5Voutput).
A150V Schottky willdecrease the coupledeffects.

3.2.2. EMC Comparison between a 150V
Schottky and a 200V bipolar diode

The better the behaviour at turn-off of the 150V
Schottky in comparisonwith a 200V bipolar diode,
the better performance in the EMC.

Fig. 16 shows the comparison of electromagnetic
disturbance conducted in a 45W/24V Flyback
converter (in continuous mode) between a
STPR1620CT and a STPS16150CT with a
snubber circuit.

Fig. 17: Electromagnetic disturbances conducted
between a STPR1620CT and a STPS16150CT

200.0

F)

p

150.0

C(

100.0

50.0

0.0
1 10 100 1000

VR(V)

In summary, when we compare the different
parameters with those of the bipolar diode, we
have:

II

>

RM RM

bipolar schottky

CC

<

jj

bipolar schottky

dI / dt

()

R

dV / dt dV / dt

()()

V

Rma

polar

bi

bipolar schottky

x Rmax

bipolar schottky

dI / dt

>

()

R

>

V>

schottky

STPR1620CT

STPS16150CT

We can see near30MHz, thatthere is a difference
of -10dB. This difference is partially explained by
the higher dV/dt with a bipolar diode at turn-off
than with a 150V Schottky. In fact, the lower
capacitance junction of the PN diode favors the
high dV/dt at t
mode. (

, and therefore thecommon current

0

i C dV / dt, C

=⋅

CM

equivalent capacitance

junction-heatsink)
The other high dV/dt, which could take place due

to the strong oscillation, are suppressed bya good
choice of the snubber circuit.

7/9

APPLICATION NOTE

In the case of anEMC problem,the firstsolution is
to reduce the current slope (dI
gate resistance (R

and dIR/dt decrease as well as the

I

RM

about 10 ohms). In this way,

G

/dt) by adding a

F

.

V

R

max

3.2.3. Switching losses

We have evaluated the consequences of poor
behaviour at the turn-off: spike reverse voltage,
possible oscillations and EMC problem. For these
reasons, the designer may wish to use a soft
recovery PN diode, but which, in return, will
increase the switching time and particularly the
switching losses at turn-off.

Switching losses at turn-off due to the diode are
the sum of losses inside the diode and the energy
dissipated in the other elements of the circuit. In
fact, during this time the I

current, due to the

RM

recovery charge, flows through the transformer,
the power MOS transistor and the primary bulk
capacitor. Thus, there are additional losses. The
distribution of power losses at turn-off can be
detailed:

- Losses inside the diode:
P

turn off b RM R

−

1

tI VF

=⋅⋅ ⋅⋅

diode

2

4. RESULT OF EXPERIMENTS

Experimental measurements (Fig. 18) were
carried out in a 45W/24V Flyback converter
working in the following conditions:

V

= 90V P

IN

F

= 100kHz Tc= 100°C

s

= 45W V

out

out

= 24V

These experiments confirm the interest in a 150V
Schottky in comparison with a 200V bipolar diode.

Fig. 18: Experiments of efficiency in a Flyback
converter

η% ∆P(W)
STPR1620CT (2x8A)
STPS16150CT (2x8A)
STPS20150CT (2x10A)

84.04 0.41

84.4 0.18

84.69 Ref

CONCLUSION

We have been able tohighlight that whenwe have
the choice between 150V Schottky and 200V PN
diode, the 150V Schottky is the best choicefor the
safety of the component and the environment, the
limitation of parasitic effects and for the efficiency
of the converter.

- Losses due to the energy store in the leakage
inductance:

1
2

2

LI

RM

W

=⋅⋅

Lf f

which is mainly dissipated in the snubber resistor

).

(R

S

- Losses due to the eddy current in the transformer
(view AN1262)

- Losses dueto the all additional resistor of circuit,
defined by:

2

PI R

=⋅

turn off

−

R

RMS

(IRM)

∑

As described before (in §1), in a typical converter
working with a switch frequency lower than
100kHz, these different losses can be considered
negligible compared to the conduction losses.

However in applications such as the DC/DC
converter (12V-48V) working with a switching
frequency around 300kHz, these losses can be
predominant, and a 150V Schottky can be very
interesting to reduce the switching losses.

In fact, in addition to the low V

, the 150V Schottky

F

has a better switching behaviour, due to its
essentially capacitive recovery (less sensibility to
the temperature). We have the advantage ofa soft
recovery diode in terms of EMC and the Schottky
is preferable to a fast recovery diode in terms of
losses. The 150V Schottky diode is the better
choice versus the 200V bipolar as for EMC and
losses at turn-off are concerned. Experimental
measurements confirm this.

Moreover, future advancements will mean that this
product will be developed.

In fact, with the arrival of theEN6100-3-2 standard
andthe introduction of thePFC, whatever the input
voltageis, there will be acontinuous voltage on the
primary. This will lead to a reduction of the
transformation ratio, and in the same time, the
reverse voltage of the diode.

Consequently, a lower breakdown voltage diode
will be needed in the future to replace a 200V PN
diode used today.

Also, the tendency is for the output power of
adaptors to increase. This involves an increase in
the output voltages. The voltage requirements of
the diode in thiscase will be higher than 100Vand
a 150V diode is likely to be the appropriate
component.

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