ST AN1300 Application note

AN1300

APPLICATION NOTE

L6598 BASED 12V/3A RESONANT APPLICATION

by Eric Danstrom

Off-line switching power supplies have two major problems: high switching losses, and an operating en­vironment, which is very sensitive to the radiation of Electromagnetic Interference (EMI) and Radio Fre­quency Interference (RFI). The switching losses are a largest creator of EMI (conducted) and RFI
(radiated), sotheir control and reduction isamajor benefit to the power supplydesigner. Resonanttech­niques offer a hope for greatly reducing the switching losses and hence the factors contributing to the
EMI and RFI. This paper will present and organized approach to the design of a 36-watt, full-resonant,
off-line converter that can be used for portable PC applications.

Operation of the Resonant Converters

Full-resonant converters are based upon the half-bridge and full-bridge PWM topologies. They are driven with
a symmetrical w aveformwhose frequency is changed to control the output voltage. There is a deadtime provid­ed between the conduction of the upper and lower power switches to turn-on with zero voltage witching. Here
the drain voltage of the MOSFET turning off immediately swings to the opposite voltage and causes the oppos­ing antiparallel diode to conduct The opposing MOSFET can then turn-on with a very low drain-to-source volt­age.

There are two major types of resonant converters: the parallel- resonant converter shown in figure 1a, and the
series-resonant converter shown in figure 1b.

Figure 1. The Two Types of Resonant Converters

October 2000

(a) Parallel

(b) Series

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AN1300 APPLICATION NOTE

In the parallel-resonant converter, the load is placed in parallel with the resonant tank capacitor. This is done
through a forward-mode transformer where the output rectifiers conduct simultaneously with the positive and
negative excursions of voltage on the primary winding. The load though, must represent as high an impedance
as possible so it does not load the tank capacitor too much which would excessively lower the tank circuit's "Q".
This is done by the use of a choke-input (L-C) filter following the output rectifiers. The choke input filter is high
impedance when viewed above its filter pole frequency. The output voltage is mathematically the area under
the curve of the output voltage waveform. The number of turns in the secondary winding is therefore higher
than in a series resonant converter.

In the series resonant converter, which is proposed here, the load is placed in series with the resonant induc­tance and capacitance. Here the load must be low impedance. This is best done by the use of a capacitor-input
filter on the transformer's secondary, where the tank views the low impedance of the output capacitor(s) reflect­ed to the primary winding of the transformer.

Bothresonanttopologies are driven with a symmetrical waveformabovethat tank circuit's resonance frequency.
The output power is controlled by where on the tank circuit's gain Vs frequency curve the converter is operating.
The closer, to the tank's resonance, the higher is the output power.

The series-resonant half-bridge causes the current to resonate in sinusoidal fashion when the tank circuit's volt­age is excited at its resonance frequency. The tank circuit is made-up primarily of the resonant inductor (L
the Capacitor (C

), but also incorporates the transformer parasitic elements which also include to varying de-

r

grees, the primary and secondary leakage inductances and the inter-turn and inter winding capacitances. The
load impedance on the secondary of the transformer is also reflected to the primary circuit and becomes a por­tion of the tank circuit. By incorporating these parasitic elements into the tank circuit, the noise typically gener­ated by their unpredictable behavior within PWM converters is harnessed for real work by the converter.

The half-bridge capacitors (C1, C2) are in series with the tank circuit. Traditionally these are high enough in val­ue such that the center node stays at a fixed voltage of approximately one-half the DC input voltage. These ca­pacitors though could be reduced in value and used as the resonant capacitor itself, thus eliminating the
resonant capacitor. Going yet another step further, these capacitors are electrically in parallel within the reso­nant circuit, and one of them can be eliminated with no detriment to the converter's operation. The upper ca­pacitor would be the logical choice to eliminate from a reliability standpoint.

The external resonant inductor can also be eliminated, if one realizes that the leakage inductance is in series
with the primary winding, as is the external resonant capacitor. If one purposely makes a transformer with high
primary leakage inductance,one can then eliminate the external resonant inductor in some cases. For the pow­er systems (220-240 VAC), where the transformer has more turns on the primary winding, developing a leakage
inductance of around 100

µ

H is possible. This is done by using a 2-sectioned bobbin where the primary winding
is placed in one-half of the bobbin and the secondary in the other half. This lowering of the coupling between
the primary and secondary and the core, raises the leakage inductance. For the low voltage AC power systems
(100 - 120 VAC), it can be more difficult to create the large leakage inductance, so an external inductor may still
need to be placed within the tank circuit.

The tank circuit's approximate resonance frequency, neglecting the dead time period, can then be then calcu­lated by:

) and

r

f

------------------------------ -=

o

2π L'rC'r

1

[Eq. 1]

Where: Lord is the series combination of Lelk + Lr(ext) if used
C'r is the AC value of the bridge capacitor(s)
This frequency will be the primary point of reference for the converter. The control circuit must always stay

above this point to maintain the advantage of zero current switching in the semiconductors.
The output load determines the "Q" of the power stage. Its equivalent resistance is reflected through the trans-

former. This is given by equation 2.

L

r

----- -

C

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---------------------------- -=

Q

R

r

+

priRrefl

[Eq. 2]

AN1300 APPLICATION NOTE

Figure 2. The Control-Transfer Function of the Series-Resonant Converter

The damping resistance is the sum of the resistances within the primary circuits and the reflected load resis­tance. The equivalent resistance is reflected through the transformer. This is given by equation 3.

2

N1



R

refl

The typical transfer function of the resonant current within the tank circuit is shown in figure 2. It shows when
the converter provides more output power by lowering its frequency. The amplitude of the current grows signif­icantly. Conversely, whenthe load lightens, the control loop increases the frequency away from the tank circuit's
resonance. Simultaneously, the Q of the tank circuit decreases slightly.

Figure 3 shows the converter operating at 110% of rated load and the operating frequency is permitted to go
under the resonant frequency of the tank. This condition may occur when the input voltage is at its minimum
rating and the output is at rated or slightly over-rated load and the frequency lower limit is not set properly.

Period 1 is the deadtime of the power switches when the center node voltage between the MOSFETs swings
from one voltage rail to the opposite rail. After the voltage has transitioned, the primary current flows through
the opposing MOSFET’s antiparallel diode.

The rate of change in the voltage waveform is dictated by the sum of the parasitic capacitances on the node.
These would be primarily the output capacitances of the MOSFETs.

Period 2 begins when the opposing MOSFET turns-on. Current is already flowing through the antiparallel diode
when the MOSFET turns ON. The current will continue its decline driven by the tank circuit and then reverse
direction. During periods 1 and 2, the secondary voltage on the forwardwinding transitions to the output voltage
and the winding’s current begins a sinusoid-like increase. the peak and decline again towards zero where an­other deadtime is encoundered and the period repeats in the opposing direction.

When the input voltage is low and the output power draw is at or greater than its rated putput, one begins to
notice a current valley begin to appear just past the peaks of the primary current. The supply may be operating
below resonance. This "inflection" in the primary current waveform increases as the input voltage is further low­ered or the output load is increased.

Waveforms at light load can be seen in figure 4. Here, the two normal operating periods (1 & 2) are seen.

------- -



N2

R load()=

[Eq. 3]

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