ST AN1061 Application note

AN1061

APPLICATION NOTE

DESIGNING WITH L4978,

2A HIGH EFFICIENCY DC-DC CONVERTERY

1 INTRODUCTION

The L4978 is a 2A monolithic dc-dc converter, step- down , operating at fix frequency contin­uous mode.

It is realised in BCD60 II technology, and it’s available in two plastic packages, MINIDIP and
SO16L.

One direct fixed output voltage at 3.3V ±1% is available, adjustable for higher output voltage
values, till 40V, by an external voltage divider.

The operating input supply voltage ranges from 8V to 55V, while the absolute value, with no
load, is 60V.

New internal design solutions and superior technology performance allow to generate a device
with improved efficiency in all the operating conditions and with reduced EMI due to an inno­vative internal driving circuit, and reduced external component counts.

While internal limiting current and thermal shutdown are today considered standard protection
functions, mandatory for a safe load supply, oscillator with voltage feedforward improves line
regulation and overall control loop.

Soft-start avoids output overvoltages at turn-on, while, shorting this pin to ground, the device
is completely disabled, going into zero consumption state.

Figure 1.

AN1061/0505

Rev. 9

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AN1061 APPLICATION NOTE

2 DEVICE DESCRIPTION

For a better understanding of the device and its working principles, a short description of the
main building blocks is given here below, with packaging options and complete block diagram.

Figure 1 shows the two packaging options, with the pin function assignments.

Figure 2. Pins Connection.

GND

SS_INH

OSC

OUT

1

2

3

4 VCC

D97IN595

Figure 3. Block Diagram

THERMAL

SHUTDOWN

FB

2

INHIBIT SOFTSTART

7

8

3.3V

E/A

OSCILLATOR

SS_INH

COMP

7

6

5

VOLTAGES

MONITOR

PWM

FB8

COMP

BOOT

3.3V

INTERNAL

REFERENCE

R

Q

S

N.C.

SUPPLY

2

3

4

5

6

7

8

D97IN596

5.1V

DRIVE

GND

SS_INH

OSC

OUT

OUT

N.C.

N.C. N.C.

INTERNAL

16

15

14

13

12

11

10

9

VCC

5

CBOOT

CHARGE

CBOOT

CHARGE

AT LIGHT

LOADS

N.C.1

N.C.

FB

COMP

BOOT

VCC

N.C.

6

BOOT

3

OSC GND OUT

1

4

D97IN594

3 POWER SUPPLY & VOLTAGE REFERENCE

The device is provided with an internal stabilised power supply (of about 12V typ. ) that powers
the analog and digital control blocks and the bootstrap section.

From this preregulator, a 3.3V reference voltage ±2%, is internally available.
Oscillator and voltage feedforward.
Just one pin is necessary to implement the oscillator function, with inherent voltage feedfor-

ward.

2/21

Figure 4. Oscillator Internal Circuit.

V

CC

AN1061 APPLICATION NOTE

R

OSC

C

OSC

TO PWM

COMPARATOR

Osc

Q

1

5R

-

+

Q

R

2

1V

CLOCK

D97IN655A

A resistor Rosc and a capacitor Cosc connected as shown in Figure 4, allow the setting of the
desired switching frequency in agreement with the below formula:

1




6

-- -

5

100 C

⋅+

osc

Where F

is in kHz, R

sw

in KΩ and C

osc

SW

--------------------------------------------------------------------------------------=

R

oascCosc

in nF.

osc

⋅()In

The oscillator capacitor, Cosc, is discharged by an internal mos transistor with 100W of Rdson
(Q1) and during this period the internal threshold is set at 1V by a second mos, Q2 . When the
oscillator voltage capacitor reaches the 1V threshold, the output comparator turns off the mos
Q1 and turns on the mos Q2, restarting the Cosc charge.

The oscillator block, shown in figure 5, generates a sawtooth wave signal that sets the switch­ing frequency of the system.

Figure 5. Switching frequency vs. Rosc and Cosc.

fsw

(KHz)

500

200

100

50

20

10

5

0 20 40 60 80 R2(KΩ)

0.82nF

1.2nF

2.2nF

3.3nF

5.6nF

D97IN630

Tamb=25˚C

4.7nF

This signal, compared with the output of the error amplifier, generates the PWM signal that will
modulate the conduction time of the power output stage.

The way the oscillator has been integrated,does not require additional external components to
benefit of the voltage feedforward function.

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AN1061 APPLICATION NOTE

The oscillator peak-to-valley voltage is proportional to the supply voltage, and the voltage
feedforward is operative from 8V to 55V of input supply.

osc

VCC1–

--------------------=

6

∆V

Also the ∆V/∆t of the sawtooth is directly proportional to the supply voltage. As Vcc increases,
the Ton time of the power transistor decreases in such a way to provide to the chocke, and
finally also the load, the product Voltxsec constant.

Figure 6 shows how the duty cycle varies as a result of the change on the ∆V/∆t of the sawtooth
with the Vcc.

Figure 6. Voltage Feedforward Function

V1

V2-3

D97IN684

Vi=30V

Vi=15V

Vc

t

Vi=30V

Vi=15V

t

The output of the error amplifier doesn’t change in order to maintain the output voltage con­stant and in regulation.

With this function on board, the output response time is greatly reduced in presence of an
abrupt change on the supply voltage, and the output ripple voltage at the mains frequency is
greatly reduced too.

In fact, the slope of the ramp is modulated by the input ripple voltage, generally present in the
order of some tens of Volt, for both off-line and dc-dc converters using mains transformers.

The charge and discharge time are approximable to:

6

In



-- -



5

T

ch

⋅⋅=

R

oscCosc

T

100 C

dis

⋅=

osc

The maximum duty cycle is a function of Tch, Tdis and an internal delay and is expressed by
the equation:

Dmax

R

oscCosc

----------------------------------------------------------------------------------=

R

oscCosc

In

In

6



-- -



5

6



-- -



5

100 C

80 10

⋅–⋅⋅

⋅+⋅⋅

9–

osc

and is represented in figure 7.

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AN1061 APPLICATION NOTE

Figure 7. Maximum Duty Cycle vs Rosc and Cosc as Parameter

D

max

0.90

0.80

0.70

0.60

5.3nF

4.7nF

2.2nF

1.2nF

0.8nF

0 4 8 121620242832R

D97IN685

OSC

(KΩ)

3.1 Current Protection

The L4978 has two current limit levels, pulse by pulse and hiccup modes.
Increasing the output current till the pulse by pulse limiting current threshold (Ith1 typ. value of

3A) the controller reduces the on-time till the value of TB = 300ns that is a blanking time in
which the current limit protection does not trigger. This minimun time is necessary to avoid un­desirable intervention of the protection due to the spike current generated during the recovery
time of the freewheeling diode.

In this condition, because of this fixed balnking time, the output current is given by:

I

max

----------------------------------------------------------------------------------------------------------------------------------------------------=

R

ORDRL

VCCTBF

SWVf

+()1TBFSW⋅–()R

1TBFSW⋅–()⋅–⋅⋅[]

+()TBFSW⋅++[]

dsonRL

Where Ro is the load resistance, Vf is the diode forward voltage. RD and RL are the series
resistance of,

respectively, the freewheeling diode and the choke.
Typical output characteristics are represented in figure 8 and 9.
In fig 8, the pulse by pulse protection is sufficient to limit the current.
In fig 9 the pulse by pulse protection is no more effective to limit the current due to the minimun

Ton fixed by the blanking time TB, and the hiccup protection intervenes because the output
peak current reachs the relative threshold.

Figure 8. Output Characteristic Figure 9. Output Characteristic

V

O

D98IN909A

A

B

C

3.6A

3A

I

O

V

O

D99IN1077

3A

A

I

3.6A

O

5/21

AN1061 APPLICATION NOTE

At the pulse by pulse intervention (point A) the output voltage drops because of the Ton reduc­tion, and the current is almost constant. Going versus the short circuit condition, the current is
only limited by the series resistances RD and RL (see relation above) and could reach the hic­cup threshold (point B), set 20% higher than the pulse by pulse. Once the hiccup limiting cur­rent is operating, in output short circuit

condition, the delivered average output current

decreases dramatically at very low values (point C).

Figure 10. Current Limit internal schematic circuit.

Q

S

OSC

R

V

Th1

+

-

V

Th2

+

-

V

CC

12V

OUT

C

SS

OSC

VFB

PWM

+-

VREF

+-

HICCUP

THERMAL

UNDERVOLTAGE

D97IN658

SOFT START

LATCH

Q

S

R

+

-

0.4

Figure 10 shows the internal current limiting circuitry. Vth1 is the pulse by pulse while Vth2 is
the hiccup threshold.

The sense resistor is in series with a small mos realised as a partition of the main DMOS.
The Vth2 comparator (20% higher than Vth1) sets the soft start latch, initialising the discharge

of the soft start capacitor with a constant current (about 22µA). Reaching about 0.4V, the valley
comparator resets the soft start latch, restarting a new recharge cycle.

Figure 11 Shows the typical waveforms of the current in the output inductor and the soft start
voltage (pin 2).

Figure 11. Output current and soft-start voltage

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AN1061 APPLICATION NOTE

Figure 12. Maximum Soft Start

Capacitance with f

L

(µH)

fsw=100KHz

400

300

200

100

0

15 20 25 30 35 40 45 50 V

680nF

SW

= 100kHz

D97IN745

470nF

330nF

220nF

100nF

(V)

CCmax

Figure 13. Maximum Soft Start

Capacitance with fSW = 200kHz

L

(µH)

300

200

100

0

15 20 25 30 35 40 45 50 VCCmax(V)

fsw=200KHz

D97IN746

56nF

47nF

33nF

22nF

During the recharging of the soft start capacitor, the Ton increases gradually and, if the short
circuit is still present, when Ton>T

and the output peak current reachs the threshold, the hic-

B

cup protection intervenes again. So, the value of the soft start capacitor must not be too high
(in this case the Ton increases slowly thus taking much time to reach the T

value) to avoid

B

that during the soft start slope the current exceeds the limit before the protection activation.
The folllowing diagrams of Figure 12 and Figure 13 show the maximum allowed soft-start ca-

pacitor as a function of the input voltage, inductor value and switching frequency. A minimun
value of the soft start capacitance is necessary to guarantee, in short circuit condition, the
functionality of the limiting current circuitry. Infact, with a capacitor too small, the frequency of
the current peaks (see figure 11) is high and the mean current value in short circuit increases.

3.2 Soft Start and Inhibit functions.

The soft start and the inhibit functions are realised using one pin only, pin2. Soft-start is re­quested to inizialise all internal functions with a correct start-up of the system without over­stressing the power stage, avoiding the intervention of the current protection, and having an
output voltage rising smoothly without output overshoots.

At Vcc Turn-on or having had an intervention of inhibit function, an initial 5µA internal current
generator starts to charge the soft-start capacitor, from 0V to about 1.8V. From this hysteretic
threshold, a 40µA current generator is activated, putting in off state the previous generator.

At this point, the output PWM starts, initiating the rising phase of the output voltage.
The soft-start capacitor is quickly discharged in case of:

■ Thermal protection intervention

■ Hiccup limiting current condition

■ Supply voltage lower than UVLO off threshold.

The soft-start and inhibit schematic diagram is shown in figure 14.

7/21