ST AN1061 Application note

AN1061

APPLICATION NOTE

DESIGNING WITH L4978, 2A HIGH EFFICIENCY DC-DC CONVERTERY

1 INTRODUCTION

The L4978 is a 2A monolithic dc-dc converter, stepdown , operating at fix frequency continuous mode.

It is realised in BCD60 II technology, and it’s available in two plastic packages, MINIDIP and SO16L.

One direct fixed output voltage at 3.3V ±1% is available, adjustable for higher output voltage values, till 40V, by an external voltage divider.

The operating input supply voltage ranges from 8V to 55V, while the absolute value, with no load, is 60V.

New internal design solutions and superior technology performance allow to generate a device with improved efficiency in all the operating conditions and with reduced EMI due to an innovative internal driving circuit, and reduced external component counts.

While internal limiting current and thermal shutdown are today considered standard protection functions, mandatory for a safe load supply, oscillator with voltage feedforward improves line regulation and overall control loop.

Soft-start avoids output overvoltages at turn-on, while, shorting this pin to ground, the device is completely disabled, going into zero consumption state.

Figure 1.

	Rev. 9
AN1061/0505	1/21

AN1061 APPLICATION NOTE

2 DEVICE DESCRIPTION

For a better understanding of the device and its working principles, a short description of the main building blocks is given here below, with packaging options and complete block diagram. Figure 1 shows the two packaging options, with the pin function assignments.

Figure 2. Pins Connection.

				N.C.	1	16	N.C.
				GND	2	15	N.C.
GND	1	8	FB	SS_INH	3	14	FB
SS_INH	2	7	COMP	OSC	4	13	COMP
OSC	3	6	BOOT	OUT	5	12	BOOT
OUT	4	5	VCC	OUT	6	11	VCC
		D97IN595		N.C.	7	10	N.C.
				N.C.	8	9	N.C.
						D97IN596

Figure 3. Block Diagram

					VCC
					5
	THERMAL	VOLTAGES
	SHUTDOWN	MONITOR
						CBOOT
2						CHARGE
			INTERNAL	INTERNAL
SS_INH	INHIBIT	SOFTSTART
			REFERENCE	SUPPLY	5.1V
		3.3V
7						6
COMP
8	E/A					BOOT
		PWM
FB			R
3.3V			Q			CBOOT
			S			CHARGE
					DRIVE	AT LIGHT
	OSCILLATOR					LOADS
	3		1		4
	OSC		GND		OUT	D97IN594

3 POWER SUPPLY & VOLTAGE REFERENCE

The device is provided with an internal stabilised power supply (of about 12V typ. ) that powers the analog and digital control blocks and the bootstrap section.

From this preregulator, a 3.3V reference voltage ±2%, is internally available. Oscillator and voltage feedforward.

Just one pin is necessary to implement the oscillator function, with inherent voltage feedforward.

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AN1061 APPLICATION NOTE

Figure 4. Oscillator Internal Circuit.

	VCC
ROSC	TO PWM	5R
	COMPARATOR
	Osc	-	CLOCK
		+
	Q1	R	Q2
COSC

1V

D97IN655A

A resistor Rosc and a capacitor Cosc connected as shown in Figure 4, allow the setting of the desired switching frequency in agreement with the below formula:

FSW	=	1
		-------------------------------------------------------------------------------------	6	+ 100 Cosc
		(Roasc Cosc)In	--
			5

Where Fsw is in kHz, Rosc in KΩ and Cosc in nF.

The oscillator capacitor, Cosc, is discharged by an internal mos transistor with 100W of Rdson (Q1) and during this period the internal threshold is set at 1V by a second mos, Q2 . When the oscillator voltage capacitor reaches the 1V threshold, the output comparator turns off the mos Q1 and turns on the mos Q2, restarting the Cosc charge.

The oscillator block, shown in figure 5, generates a sawtooth wave signal that sets the switching frequency of the system.

Figure 5. Switching frequency vs. Rosc and Cosc.

fsw					D97IN630
(KHz)
500				Tamb=25˚C
		0.
200		82nF
		1.
100			2nF
			2.
			2nF
50			3.
			3nF
				4.
20				7nF
			5.
			6nF
10
5	20	40	60	80	R2(KΩ)
0

This signal, compared with the output of the error amplifier, generates the PWM signal that will modulate the conduction time of the power output stage.

The way the oscillator has been integrated,does not require additional external components to benefit of the voltage feedforward function.

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AN1061 APPLICATION NOTE

The oscillator peak-to-valley voltage is proportional to the supply voltage, and the voltage feedforward is operative from 8V to 55V of input supply.

∆ VCC – 1

Vosc = --------------------

6

Also the ∆V/∆t of the sawtooth is directly proportional to the supply voltage. As Vcc increases, the Ton time of the power transistor decreases in such a way to provide to the chocke, and finally also the load, the product Voltxsec constant.

Figure 6 shows how the duty cycle varies as a result of the change on the ∆V/∆t of the sawtooth with the Vcc.

Figure 6. Voltage Feedforward Function

V1	Vi=30V
	Vi=15V
	Vc
	t
V2-3
	Vi=30V
	Vi=15V
	t
	D97IN684

The output of the error amplifier doesn’t change in order to maintain the output voltage constant and in regulation.

With this function on board, the output response time is greatly reduced in presence of an abrupt change on the supply voltage, and the output ripple voltage at the mains frequency is greatly reduced too.

In fact, the slope of the ramp is modulated by the input ripple voltage, generally present in the order of some tens of Volt, for both off-line and dc-dc converters using mains transformers. The charge and discharge time are approximable to:

Tch =	Rosc Cosc	6
		In --
		5

Tdis = 100 Cosc

The maximum duty cycle is a function of Tch, Tdis and an internal delay and is expressed by the equation:

6 –9

Rosc Cosc In -- – 80 10 5

Dmax = ----------------------------------------------------------------------------------

6

Rosc Cosc In --5 + 100 Cosc

and is represented in figure 7.

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AN1061 APPLICATION NOTE

Figure 7. Maximum Duty Cycle vs Rosc and Cosc as Parameter

Dmax							D97IN685
		5.3nF
0.90			4.7nF
			2.2nF
			1.2nF
			0.8nF
0.80
0.70
0.60							32 ROSC(KΩ)
0	4	8	12 16	20	24	28

3.1 Current Protection

The L4978 has two current limit levels, pulse by pulse and hiccup modes.

Increasing the output current till the pulse by pulse limiting current threshold (Ith1 typ. value of 3A) the controller reduces the on-time till the value of TB = 300ns that is a blanking time in which the current limit protection does not trigger. This minimun time is necessary to avoid undesirable intervention of the protection due to the spike current generated during the recovery time of the freewheeling diode.

In this condition, because of this fixed balnking time, the output current is given by:

[VCC TB FSW – Vf (1 – TB FSW)]

I = ---------------------------------------------------------------------------------------------------------------------------------------------------

max [RO + (RD + RL)(1 – TB FSW) + (Rdson + RL)TB FSW]

Where Ro is the load resistance, Vf is the diode forward voltage. RD and RL are the series resistance of,

respectively, the freewheeling diode and the choke.

Typical output characteristics are represented in figure 8 and 9.

In fig 8, the pulse by pulse protection is sufficient to limit the current.

In fig 9 the pulse by pulse protection is no more effective to limit the current due to the minimun Ton fixed by the blanking time TB, and the hiccup protection intervenes because the output peak current reachs the relative threshold.

Figure 8. Output Characteristic	Figure 9. Output Characteristic
VO	VO
A	A

		B
	C	IO		IO
	D98IN909A		D99IN1077
		3A 3.6A		3A 3.6A

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AN1061 APPLICATION NOTE

At the pulse by pulse intervention (point A) the output voltage drops because of the Ton reduction, and the current is almost constant. Going versus the short circuit condition, the current is only limited by the series resistances RD and RL (see relation above) and could reach the hic-

cup threshold (point B), set 20% higher than the pulse by pulse. Once the hiccup limiting current is operating, in output short circuit condition, the delivered average output current decreases dramatically at very low values (point C).

Figure 10. Current Limit internal schematic circuit.

OSC	S Q
	R	VCC
	VTh1	VTh2
		+
	+	-
	-

OUT

PWM

12V

+-

OSC	HICCUP	SOFT START
		LATCH
	THERMAL	S Q
	UNDERVOLTAGE	R

-+

VFB

VREF

	+	0.4
D97IN658	-

CSS

Figure 10 shows the internal current limiting circuitry. Vth1 is the pulse by pulse while Vth2 is the hiccup threshold.

The sense resistor is in series with a small mos realised as a partition of the main DMOS. The Vth2 comparator (20% higher than Vth1) sets the soft start latch, initialising the discharge of the soft start capacitor with a constant current (about 22µA). Reaching about 0.4V, the valley comparator resets the soft start latch, restarting a new recharge cycle.

Figure 11 Shows the typical waveforms of the current in the output inductor and the soft start voltage (pin 2).

Figure 11. Output current and soft-start voltage

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	AN1061 APPLICATION NOTE
Figure 12. Maximum Soft Start	Figure 13. Maximum Soft Start
Capacitance with fSW = 100kHz	Capacitance with fSW = 200kHz

L								D97IN745
					680nF
( H)
								470nF
		fsw=100KHz
400
								330nF
300
200							220nF
100								100nF
0
15	20	25	30	35	40	45	50	VCCmax(V)

L

D97IN746

( H)

fsw=200KHz

56nF

300

47nF

200

33nF

22nF

100

0

15 20 25 30 35 40

45 50 VCCmax(V)

During the recharging of the soft start capacitor, the Ton increases gradually and, if the short circuit is still present, when Ton>TB and the output peak current reachs the threshold, the hiccup protection intervenes again. So, the value of the soft start capacitor must not be too high (in this case the Ton increases slowly thus taking much time to reach the TB value) to avoid that during the soft start slope the current exceeds the limit before the protection activation.

The folllowing diagrams of Figure 12 and Figure 13 show the maximum allowed soft-start capacitor as a function of the input voltage, inductor value and switching frequency. A minimun value of the soft start capacitance is necessary to guarantee, in short circuit condition, the functionality of the limiting current circuitry. Infact, with a capacitor too small, the frequency of the current peaks (see figure 11) is high and the mean current value in short circuit increases.

3.2 Soft Start and Inhibit functions.

The soft start and the inhibit functions are realised using one pin only, pin2. Soft-start is requested to inizialise all internal functions with a correct start-up of the system without overstressing the power stage, avoiding the intervention of the current protection, and having an output voltage rising smoothly without output overshoots.

At Vcc Turn-on or having had an intervention of inhibit function, an initial 5µA internal current generator starts to charge the soft-start capacitor, from 0V to about 1.8V. From this hysteretic threshold, a 40µA current generator is activated, putting in off state the previous generator.

At this point, the output PWM starts, initiating the rising phase of the output voltage. The soft-start capacitor is quickly discharged in case of:

■Thermal protection intervention

■Hiccup limiting current condition

■Supply voltage lower than UVLO off threshold.

The soft-start and inhibit schematic diagram is shown in figure 14.

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