SGS Thomson Microelectronics TS615IPWT, TS615 Datasheet

TS615

DUAL WIDE BAND OPERATIONAL AMPLIFIER

WITH HIGH OUTPUT CURRENT

■ LOW NOISE : 2.5nV/√Hz

■ HIGH OUTPUT CURRENT : 420mA

■ VERY LOW HARMONIC AND INTERMODU-

LATION D I S TORTIO N

■ HIGH SLEW RA TE : 410V/µs

■ -3dB BANDWIDTH : 40MHz@gain=12dB on

25Ω load single ended.

■ 21.2Vp-p DIFFERENTIAL OUTPUT SWING

on 50Ω load, 12V power supply

■ CURRENT FEEDBACK STRUCTURE

■ 5V to 12V POWER SUPPLY

■ SPECIFIED FOR 20Ω and 50Ω DIFFER EN-

TIAL LOAD

■ POWER DOWN FUNCTION WITH A SHORT

CIRCUITED OUTPUT to keep the matching with the line in sleep mode

DESCRIPTION

The TS615 is a dual operational am plifier featuring a high output current 410mA. These drivers can be configured differentially for driving signals in telecommunication systems using multiple carriers. The TS615 is ideally suited for xDSL (High Speed Asymmetrical Digital Subscriber Line) applications. This circuit is c apable of driving a 10 Ω or 25Ω load at ±2.5V, 5V, ±6V or +12V power supply. The TS615 will be able to reach a -3dB bandwidth of 40MHz on 25Ω load with a 12dB gain. This device is designed for the high slew rates to support low harmonic distortion and intermodulation. The TS615 is fitted out with Power Down function to decrease the consumption. During this sleep state the device displays a short circuit output in order to keep the impedance matching with the line. The TS615 is housed in TSSOP14 Exposed-Pad plastic package for a very low thermal resistance.

P

TSSOP14 Exposed-Pad

(Plastic Micro package)

ORDER CODE

Part Number Temperature Range Package

TS615IPWT -40, +85°C PW

PW= Thin Shrink Small Outline Package with Exposed-Pad

(TSSOP Exposed-Pad) only available in Tape & Reel (PWT)

PIN CONNECTIONS (top view)

-VCC1

+VCC1

Non Inverting Input1

Inverting Input1

Power Down

NC

1

1 2

2 3

3

+ - - +

4

4 5

5 6

6 7

7

Top View

14

-VCC2

13

Output2Output1

12

+VCC2

Non Inverting Input2

11

11 10

10

Inverting Input2

Inverting Input2 NC

NC

9

NC

8

APPLICATION

■ Line driver for xDSL

■ Multiple Video Line Driver

December 2002

Cross Section V iew Showi ng Exposed-Pa d

This pad c an be con nected to a (-Vcc) copper ar ea on the PCB

1/27

TS615

ABSOLUTE MAXIMUM RATINGS

Symbol Parameter Value Unit

V

T

R R P ESD

except

pins 4, 5,

10, 11

ESD

only pins 4,

5, 10, 11

Supply voltage

CC

V

Differential Input Voltage

id

V

in Input Voltage Range

Operating Free Air Temperature Range -40 to + 85 °C

oper

Storage Temperature -65 to +150 °C

std

T

Maximum Junction Temperature 150 °C

j

Thermal Resistance Junction to Case 4 °C/W

thjc

Thermal Resistance Junction to Ambient Area 40 °C/W

thja

Maximum Power Dissipation (@25°C) 3.1 W

max.

CDM : Charged Device Model HBM : Human Body Model MM : Machine Model CDM : Charged Device Model HBM : Human Body Model MM : Machine Model Output Short Circuit

1. All voltage values, except differential voltage are with respect to network terminal.

2. Differential voltage are non-inverting input terminal with respect to the inverting input terminal.

3. The magnitude of input and output voltage must never exceed V

4. An output current limitation protects the circuit from transient currents. Short-circuits can cause excessive heating. Destructive dissipation can result from short circuit on amplifiers.

1)

2)

3)

±7 V ±2 V ±6 V

1.5 2

200

1 1

100

4)

+0.3V.

CC

kV kV

V kV kV

V

OPERATING CONDITIONS

Symbol Parameter Value Unit

V

Power Supply Voltage ±2.5 to ±6 V

CC

+1.5V to +VCC-1.5V

Common Mode Input Voltage

icm

-V

CC

TYPICAL APPLICATION:

Differential Line Driver for xDSL Applications

12

11

10

Vi

R1

R4

Vi Vo

5

4

Pw-Dwn

+

1/2TS615

_

14

R2

GND

R3

3

_

1/2TS615

+

1

6

+Vcc

-Vcc

+Vcc

-Vcc

13

2

12.5Ω

Vo

1:2

25Ω 100Ω

V

2/27

TS615

ELECTRICAL CHARACTERISTICS

V

= ±6Volts, Rfb=910Ω,T

CC

Note: as described on page 24 (table 71), the TS615 requires a 620Ω feedback res i st or for an optimis ed bandwidth wi t h a gai n of 12 B for a 12V power supply. Nevertheless, due to production test constraints, the TS615 is tested with the same feedback resistor for 12V and 5V power su ppl i es (910Ω).

Symbol Parameter Test Condition Min. Typ. Max. Unit

DC PERFORMANCE

V

Input Offset Voltage

io

V

∆

Z

C

CMR

SVR

Differential Input Offset Voltage

io

I

Positive Input Bias Current

ib+

I

Negative Input Bias Current

ib-

Input(+) Impedance 82 k

IN+

Z

Input(-) Impedance 54

IN-

Input(+) Capacitance 1 pF

IN+

Common Mode Rejection Ratio 20 log (∆V

/∆Vio)

ic

Supply Voltage Rejection Ratio 20 log (∆V

I

Total Supply Current per Operator No load 14 17 mA

CC

/∆Vio)

cc

DYNAMIC PERFORMANCE and OUTPUT CHARACTERISTIC

R

Open Loop Transimpedance

OL

-3dB Bandwidth

Full Power Bandwidth

BW

Gain Flatness @ 0.1dB

Tr Rise Time

Tf Fall Time

Ts Settling Time

SR Slew Rate

V

High Level Output Voltage

OH

V

Low Level Output Voltage

OL

Output Sink Current

I

out

Output Source Current

= 25°C (unless otherwise specified)

amb

T

amb

< T

T

min.

T

amb

T

amb

T

min.

T

amb

T

min.

V

∆

ic

T

min.

V

∆

cc

T

min.

V

out

T

min.

< T

amb

= 25°C

< T

amb

< T

amb

= ±4.5V

< T

amb

=±2.5V to ±6V < T

< T

amb

= 7Vp-p, RL = 25

< T

amb.

Small Signal V A

= 12dB, RL = 25

V

Large Signal V

= 12dB, RL = 25

A

V

Small Signal V

= 12dB, RL = 25

A

V

= 6Vp-p, AV = 12dB, RL

out

= 25

Ω

= 6Vp-p, AV = 12dB, RL

V

out

= 25

Ω

= 6Vp-p, AV = 12dB, RL

V

out

= 25

Ω

= 6Vp-p, AV = 12dB, RL

V

out

= 25

Ω

R

=25Ω Connected to GND

L

R

=25Ω Connected to GND

L

V

= -4Vp

out

< T

T

min.

V

out

T

min.

amb

= +4Vp

< T

amb

< T

max.

<20mVp

out

Ω

=3Vp

out

Ω

<20mVp

out

Ω

max.

1.25 3.5

2.1

mV

2.5 mV

630

7.8 315

3.2

A

µ

A

µ

Ω

58 63

61

72 79

78

Ω

521

8.9

dB

M

Ω

25 40

MHz

26

7 MHz

10.6 ns

12.2 ns

50 ns

330 410 V/µs

4.8 5.1 V

-5.5 -5.2 V

-350 -530

-440

330 420

mA

365

3/27

TS615

Note: as described on page 24 (table 71), the TS615 requires a 620Ω feedback res i st or for an optimis ed bandwidth wi t h a gai n of 12 B for

a 12V power supply. Nevertheless, due to production test constraints, the TS615 is tested with the same feedback resistor for 12V and 5V power su ppl i es (910Ω).

Symbol Parameter Test Condition Min. Typ. Max. Unit

NOISE AND DISTORTION

eN Equivalent Input Noise Voltage F = 100kHz 2.5 nV/√Hz iNp Equivalent Input Noise Current (+) F = 100kHz 15 pA/√Hz iNn Equivalent Input Noise Current (-) F = 100kHz 21 pA/√Hz

= 14Vp-p, AV = 12dB

HD2

HD3

IM2

IM3

2nd Harmonic distortion (differential configuration)

3rd Harmonic distortion (differential configuration)

2nd Order Intermodulation Product (differential configuration)

3rd Order Intermodulation Produ ct (differential configuration)

V

out

F= 110kHz, R

= 14Vp-p, AV = 12dB

V

out

F= 110kHz, R

= 50Ω diff.

L

= 50Ω diff.

L

F1= 100kHz, F2 = 110kHz

= 16Vp-p, AV = 12dB

V

out

= 50Ω diff.

R

L

F1= 370kHz, F2 = 400kHz

= 16Vp-p, AV = 12dB

V

out

R

= 50Ω diff.

L

F1 = 100kHz, F2 = 110kHz

= 16Vp-p, AV = 12dB

V

out

= 50Ω diff.

R

L

F1 = 370kHz, F2 = 400kHz

= 16Vp-p, AV = 12dB

V

out

= 50Ω diff.

R

L

-87 dBc

-83 dBc

-76 dBc

-75

-88 dBc

-87

4/27

TS615

ELECTRICAL CHARACTERISTICS

V

= ±2.5Volts, Rfb=910Ω,T

CC

Symbol Parameter Test Condition Min. Typ. Max. Unit

DC PERFORMANCE

V

Input Offset Voltage

io

V

∆

Z

C

CMR

SVR

Differential Input Offset Voltage

io

I

Positive Input Bias Current

ib+

I

Negative Input Bias Current

ib-

Input(+) Impedance 71 k

IN+

Z

Input(-) Impedance 62

IN-

Input(+) Capacitance 1.5 pF

IN+

Common Mode Rejection Ratio 20 log (∆V

/∆Vio)

ic

Supply Voltage Rejection Ratio 20 log (∆V

I

Total Supply Current per Operator No load 11.9 15 mA

CC

/∆Vio)

cc

DYNAMIC PERFORMANCE and OUTPUT CHARACTERISTICS

R

Open Loop Transimpedance

OL

-3dB Bandwidth

BW

Full Power Bandwidth

Gain Flatness @ 0.1dB

Tr Rise Time

Tf Fall Time

Ts Settling Time

SR Slew Rate

V

High Level Output Voltage

OH

V

Low Level Output Voltage

OL

Output Sink Current

I

out

Output Source Current

= 25°C (unless otherwise specified)

amb

T

amb

< T

T

min.

T

amb

T

amb

T

min.

T

amb

T

min.

V

∆

ic

T

min.

V

∆

cc

T

min.

V

out

T

min.

< T

amb

= 25°C

< T

amb

< T

amb

= ±1V

< T

amb.

=±2V to ±2.5V < T

< T

amb.

= 2Vp-p, RL = 10

< T

amb.

Small Signal V

= 12dB, RL = 10

A

V

Large Signal V

= 12dB, RL = 10

A

V

Small Signal V A

= 12dB, RL = 10

V

= 2.8Vp-p, AV = 12dB

out

= 10

R

Ω

L

V

= 2.8Vp-p, AV = 12dB

out

= 10

R

Ω

L

= 2.2Vp-p, AV = 12dB

V

out

= 10

R

Ω

L

= 2.2Vp-p, AV = 12dB

V

out

R

= 10

Ω

L

R

=10Ω Connected to GND

L

R

=10Ω Connected to GND

L

= -1.25Vp

V

out

< T

T

min.

V

out

T

min.

< T

amb

= +1.25Vp

< T

amb

max.

Ω

max.

<20mVp

out

Ω

= 1.4Vp

out

Ω

<20mVp

out

Ω

max.

0.5 2.5

1.2

2.5 mV 530 8

0.8 11

1.24

55 60

58

63 77

76

25.4

2.1

20 30

MHz

20

5.7 MHz

11 ns

11.5 ns

39 ns

100 130 V/µs

1.5 1.75 V

-2.05 -1.8 V

-350 -470

-450

200 270

245

mV

A

µ

A

µ

Ω

dB

M

Ω

mA

5/27

TS615

Symbol Parameter Test Condition Min. Typ. Max. Unit

NOISE AND DISTORTION

eN Equivalent Input Noise Voltage F = 100kHz 2.5 nV/√Hz iNp Equivalent Input Noise Current (+) F = 100kHz 15 pA/√Hz iNn Equivalent Input Noise Current (-) F = 100kHz 21 pA/√Hz

= 6Vp-p, AV = 12dB

HD2

HD3

IM2

IM3

2nd Harmonic distortion (differential configuration)

3rd Harmonic distortion (differential configuration)

2nd Order Intermodulation Product (differential configuration)

3rd Order Intermodulation Produ ct (differential configuration)

V

out

F= 110kHz, R

= 6Vp-p, AV = 12dB

V

out

F= 110kHz, R

= 20Ω diff.

L

= 20Ω diff.

L

F1= 100kHz, F2 = 110kHz

= 6Vp-p, AV = 12dB

V

out

R

= 20Ω diff.

L

F1= 370kHz, F2 = 400kHz V

= 6Vp-p, AV = 12dB

out

= 20Ω diff.

R

L

F1 = 100kHz, F2 = 110kHz

= 6Vp-p, AV = 12dB

V

out

= 20Ω diff.

R

L

F1 = 370kHz, F2 = 400kHz V

= 6Vp-p, AV = 12dB

out

R

= 20Ω diff.

L

-97 dBc

-98 dBc

-86

-88

-90

-85

dBc

POWER DOWN MODE FEA TURES (The Power Down command is a MOS input featuring a high input impedance)

V

= ±2.5Volts, 5Volts, ±6Volts or 12Volts, T

CC

Symbol Parameter Min. Typ. Max. Unit

Pin (6) Threshold Voltage for Power Down Mode

V

Low Level

pdw

High Level

Icc

R C

Power Down Mode Total Current Consumption@ VCC=5V

pdw

Power Down Mode Total Current Consumption@ V Power Down Mode Output Impedance @ VCC=5V

pdw

Power Down Mode Output Impedance @ V Power Down Mode Output Capacitance 63 pF

pdw

POWER DOWN CONTROL CIRCUIT STATUS

V

=Low Level

pdw

=High Level

V

pdw

= 25°C

amb

CC

=12V

CC

Active Standby

=12V

-V

CC

+2

-VCC+0.8 +V

69 80

148 180

19 23

15.3 19

CC

V

A

µ

A

µ

Ω Ω

6/27

TS615

Figure 1 : Load Configuration

Load: RL=25Ω, VCC=±6V

+6V

TS615

+6V

-6V

+

_

25Ω

50Ω

cable

49.9Ω

33Ω

33Ω 1W

1W

cable

Figure 2 : Closed Loop Gai n vs. Frequency

AV=+1

2

0

-2

-4

-6

-8

(gain (dB)

-10

-12

-14

(Vcc=±2.5V, Rfb=1.1kΩ, Rload=10Ω) (Vcc=±6V, Rfb=750

-16 100 1k 10k 100k 1M 10M 100M

gain

phase

Ω, Rload=25Ω)

Frequency (Hz)

(Vcc=±6V)

(Vcc=±2.5V)

(Vcc=±6V)

50Ω

40

20

0

-20

-40

-60

-80

-100

-120

Figure 4 : Load Configuration

Load: RL=10Ω, VCC=±2.5V

+2.5V

TS615

+2.5V

-2.5V

10

+

_

49.9

Ω

11

Ω

0.5W

Ω

Figure 5 : Closed Loop Gai n vs. Frequency

AV=-1

2

0

-2

)

°

Phase (

-4

-6

-8

(gain (dB))

-10

-12

-14

(Vcc=±2.5V, Rfb=1kΩ, Rin=1kΩ , Rload=10Ω) (Vcc=±6V, Rfb=680

-16 100 1k 10k 100k 1M 10M 100M

gain

phase

(Vcc=±2.5V)

(Vcc=±6V)

Ω, Rin=680Ω, Rload=25Ω)

Frequency (Hz)

cable

(Vcc=±2.5V)

(Vcc=±6V)

50

Ω

50

Ω

-140

-160

-180

-200

)

°

-220

Phase (

-240

-260

-280

-300

Figure 3 : Closed Loop Gai n vs. Frequency

AV=+2

8

6

4

2

0

-2

(gain (dB))

-4

-6

-8

-10 100 1k 10k 100k 1M 10M 100M

gain

phase

(Vcc=±2.5V)

Frequency (Hz)

(Vcc=±6V)

(Vcc=±2.5V)

(Vcc=±6V)

40

20

0

-20

-40

-60

-80

-100

-120

Figure 6 : Closed Loop Gai n vs. Frequency

AV=-2

8

6

4

)

°

Phase (

2

0

-2

(gain (dB))

-4

-6

(Vcc=±2.5V, Rfb=1kΩ, Rin=510Ω, Rload=10Ω)

-8

(Vcc=±6V, Rfb=680

-10 100 1k 10k 100k 1M 10M 100M

gain

phase

(Vcc=±2.5V)

(Vcc=±6V)

Ω, Rin=750//620Ω, Rload=25Ω)

Frequency (Hz)

(Vcc=±2.5V)

(Vcc=±6V)

-140

-160

-180

-200

-220

-240

-260

-280

-300

)

°

Phase (

7/27

TS615

Figure 7 : Closed Loop Gai n vs. Frequency

AV=+4

14

12

10

8

6

4

(gain (dB))

2

0

-2

(Vcc=±2.5V, Rfb=910Ω, Rg=300Ω, Rload=10Ω) (Vcc=±6V, Rfb=620

-4 100 1k 10k 100k 1M 10M 100M

gain

phase

(Vcc=±2.5V)

(Vcc=±6V)

Ω, R g =560//330Ω, Rload=25Ω)

Frequency (Hz)

(Vcc=±2.5V)

(Vcc=±6V)

Figure 8 : Closed Loop Gai n vs. Frequency

AV=+8

20

18

16

14

12

10

(gain (dB))

8

6

4

(Vcc=±2.5V, Rfb=680Ω, Rg=240//160Ω, Rload=10Ω) (Vcc=±6V, Rfb=510

2

100 1k 10k 100k 1M 10M 100M

gain

phase

(Vcc=±2.5V)

(Vcc=±6V)

Ω, Rg=270//100Ω, Rload=25Ω)

Frequency (Hz)

(Vcc=±2.5V)

(Vcc=±6V)

40

20

0

-20

-40

-60

-80

-100

-120

40

20

0

-20

-40

-60

-80

-100

-120

Figure 10 : Closed Loop Gain vs. Frequency

AV=-4

14

12

10

8

)

°

Phase (

6

4

(gain (dB))

2

0

(Vcc=±2.5V, Rfb=1kΩ, Rin=320//360Ω, Rload=10Ω)

-2

(Vcc=±6V, Rfb=620

-4 100 1k 10k 100k 1M 10M 100M

gain

phase

(Vcc=±2.5V)

(Vcc=±6V)

Ω, Rin=360//270Ω, Rload=25Ω)

Frequency (Hz)

Figure 11 : Closed Loop Gain vs. Frequency

AV=-8

)

°

Phase (

20

18

16

14

12

10

(gain (dB))

8

6

4

(Vcc=±2.5V, Rfb=680Ω, Rin=160//180Ω, Rload=10Ω) (Vcc=±6V, Rfb=510

2

100 1k 10k 100k 1M 10M 100M

gain

phase

(Vcc= ± 2. 5V)

(Vcc=±6V)

Ω, Rin=150//110Ω, Rload=25Ω)

Frequency (Hz)

(Vcc=±2.5V)

(Vcc=±6V)

(Vcc=±2.5V)

(Vcc= ± 6V )

-140

-160

-180

-200

-220

-240

-260

-280

-300

-140

-160

-180

-200

-220

-240

-260

-280

-300

)

°

Phase (

)

°

Phase (

Figure 9 : Bandwidth vs. Temperature

AV=+4, Rfb=910

50

45

40

35

Bw (MHz)

30

25

20

-40-200 20406080

8/27

Ω

Vcc=±6V Load=25

Vcc=±2.5V Load=10

Ω

Temperature (°C)

Figure 12 : Positive Slew Rate

AV=+4, Rfb=620

4

2

(V)

0

OUT

V

-2

-4

0.0 10.0n 20.0n 30.0n 40.0n 50.0n

, V

Ω

=±6V, RL=25

CC

Time (s)

Ω

TS615

Figure 13 : Positive Slew Rate

AV=+4, Rfb=910

2

1

(V)

0

OUT

V

-1

-2

0.0 10.0n 20.0n 30.0n 40.0n 50.0n

, V

Ω

CC

=±2.5V, RL=10

Time (s)

Ω

Figure 14 : Negative Slew Rate

AV=+4, Rfb=620Ω, VCC=±6V, RL=25

4

2

Ω

Figure 16 : Positive Slew Rate

AV= - 4, Rfb=620

4

2

(V)

0

OUT

V

-2

-4

0.0 10.0n 20.0n 30.0n 40.0n 50.0n

, V

Ω

=±6V, RL=25

CC

Time (s)

Ω

Figure 17 : Positive Slew Rate

AV= - 4, Rfb=910

2

1

, V

Ω

CC

=±2.5V, RL=10

Ω

(V)

0

OUT

V

-2

-4

0.0 10.0n 20.0n 30.0n 40.0n 50.0n

Time (s)

Figure 15 : Negative Slew Rate

AV=+4, Rfb=910

2

1

(V)

0

OUT

V

-1

-2

0.0 10.0n 20.0n 30.0n 40.0n 50.0n

, V

Ω

CC

=±2.5V, RL=10

Time (s)

Ω

(V)

0

OUT

V

-1

-2

0.0 10.0n 20.0n 30.0n 40.0n 50.0n

Time (s)

Figure 18 : Negative Slew Rate

AV= - 4, Rfb=620Ω, VCC=±6V, RL=25

4

2

(V)

0

OUT

V

-2

-4

0.0 10.0n 20.0n 30.0n 40.0n 50.0n

Time (s)

Ω

9/27

TS615

Figure 19 : Negative Slew Rate

AV= - 4, Rfb=910

2

(V)

0

OUT

V

-2

0.0 10.0n 20.0n 30.0n 40.0n 50.0n

, V

Ω

CC

=±2.5V, RL=10

Time (s)

Ω

Figure 20 : Slew Rate vs. Temperature

AV=+4, Rfb=910

200

150

100

s)

µ

50

0

−50

Slew Rate (V/

−100

−150

−200

−40 −200 20406080

, V

Ω

Positive SR

Negative SR

=±2.5V, RL=10

CC

Temperature (°C)

Ω

Figure 22 : Input Voltage Noise Level

AV=+92, Rfb=910Ω, Input+ connected to Gnd via 10

5.0

+ 6V

+

Output

4.5

Hz)

√

4.0

3.5

3.0

Input Voltage Noise (nV/

2.5

2.0 100 1k 10k 100k 1M

10

(Frequency (Hz)

Output

_

6VΩ-

- 6V

910Ω

910

910Ω

910

Ω

Figure 23 : Transimpedance vs. Temperature

Open Loop

)

Ω

(M R

30

25

Vcc=±6V

20

15

OL

10

Vcc=±2.5V

5

0

-40-200 20406080

Temperature (°C)

Ω

Figure 21 : Slew Rate vs. Temperature

AV=+4, Rfb=910

600 500 400 300 200

s)

µ

100

0

−100

−200

Slew Rate (V/

−300

−400

−500

−600

−40 −200 20406080

10/27

, V

=±6V, RL=25

Ω

CC

Positive&Negative SR

Ω

Rfb=620

Temperature (°C)

Ω

Positive&Negative SR

Ω

Rfb=910

Figure 24 : Icc vs. Power Supply

Open loop, no load

16 14 12 10

8

Icc(+)

6 4 2 0

(mA)

-2

CC

I

-4

-6 Icc(-)

-8

-10

-12

-14

-16 56789101112

VCC (V)

TS615

Figure 25 : Iib vs. Power Supply

Open loop, no load

7

IB+

6

5

4

A)

µ

(

B

3

I

IB-

2

1

0

5 6 7 8 9 10 11 12

Vcc (V)

Figure 26 : I

Open loop, no load

5

4

3

A)

µ

(

IB(-)

2

I

1

0

-40-200 20406080

ib(-) vs. Temperature

Vcc=±6V

Vcc=±2.5V

Temper ature (°C)

Figure 28 : I

ib(+) vs. Temper ature

Open loop, no load

8

7

6

5

4

A)

µ

(

3

IB(+)

I

2

1

0

-1

-40-200 20406080

Figure 29 : V

Open loop, RL=25

6 5 4 3 2

(V)

1

OL

0

& V

-1

OH

V

-2

-3

-4

-5

-6 56789101112

Vcc=±6V

Vcc=±2.5V

Temperature (° C)

oh & Vol vs. Power Supply

Ω

VOH

VOL

Vcc (V)

Figure 27 : I

cc vs. Temperature

Open loop, no load

14 12 10

Icc(+) for Vcc=±2.5V

8

Icc(+) for Vcc=±6V

6 4 2 0

(mA)

-2

CC

I

-4

-6

Icc(-) for Vcc=±6V

-8

-10

-12

-14

Icc(-) for Vcc=±2.5V

-40-200 20406080

Tem perature ( °C)

Figure 30 : V

oh vs. Temperature

Open loop

6

5

4

3

(V)

OH

V

2

1

0

-40-200 20406080

Vcc=±6vV

Ω

Load=25

Vcc=±2.5V

Ω

Load=10

Temperature (°C)

11/27

TS615

Figure 31 : Vol vs. Temperature

Open loop

0

Vcc=±2.5V Load=10

-1

-2

-3

(V)

OL

V

-4

-5

-6

-40-200 20406080

Figure 32 : Differential V

Open loop, no load

450

400

350

V)

µ

(

IO

V

∆

300

250

200

-40-200 20406080

Ω

Vcc=±6V Load=25

Ω

Temperature (°C)

Vcc=±6V

Temperature (°C)

io vs. Temperature

Vcc=±2.5V

Figure 34 : CMR vs. Temperature

Open loop, no load

70

68

66

64

62

60

58

CMR (dB)

56

54

52

50

-40-200 20406080

Vcc=±2.5V

Temperature (°C)

Vcc=±6V

Figure 35 : SVR vs. Temperature

Open loop, no load

84

82

80

SVR (dB)

78

76

Vcc=±2.5V

-40-200 20406080

Vcc=±6V

Temperature (°C)

Figure 33 : V

io vs. Temperature

Open loop, no load

2.0

Vcc=±6V

1.5

1.0

(mV)

IO

V

0.5

0.0

Vcc=±2.5V

-0.5

-40-20 0 20406080

Temperature (°C)

12/27

Figure 36 : Iout vs. Temperature

Open loop, VCC=±6V, RL=10Ω

300 250 200 150 100

Isource

50

0

-50

-100

-150

Iout (mA)

-200

-250

-300

-350

-400

-450

Isink

-40-200 20406080

Temperature (°C)

TS615

Figure 37 : Iout vs. Temperature

Open loop, VCC=±2.5V, RL=25Ω

300 250 200 150 100

50

Isource

0

-50

-100

-150

Iout (mA)

-200

-250

-300

-350

-400

-450

Isink

-40-200 20406080

Temperature (°C)

Figure 38 : Maximum Output Amplitude vs. Load

AV=+4, Rfb=620Ω, VCC=±6V

12

) (V

V

10

8

P-P

6

OUT-MAX

4

2

Vcc=±6V

Vcc=±2.5V

Figure 40 : Isource vs. Output Amplitude.

VCC=±2.5V, Open Loop, no Load

700

600

500

400

300

Isource (mA)

200

100

0

0.0 0.5 1.0 1.5 2.0 2.5

Vout (V)

Figure 41 : Isink vs. Output Amplitude

VCC=±6V, Open Loop, no Load

0

-100

-200

-300

-400

Isink (mA)

-500

-600

0

0 50 100 150 200

R

(Ω)

LOAD

Figure 39 : Isink vs. Output Amplitude.

VCC=±2.5V, Open Loop, no Load

0

-100

-200

-300

-400

Isink (mA)

-500

-600

-700

-2.5 -2.0 -1.5 -1.0 -0.5 0.0

Vou t (V)

-700

-6 -5 -4 -3 -2 -1 0

Vout (V)

Figure 42 : Isource vs. Output Amplitude

VCC=±6V, Open Loop, no Load

700

600

500

400

300

Isource (mA)

200

100

0

0123456

Vout (V)

13/27

TS615

Figure 43 : Icc (Power Down) vs. Temperature

No load, Open Loop

200

150

100

50

A)

µ

0

(

CC pdw

-50

I

-100

-150

-200

Vcc=±2.5V

-40-200 20406080

Vcc=±6V

Temperature (°C)

Figure 44 : Group Delay

VCC=±6V, VCC=±2.5V

100

90

80

70

60

50

Delay (ns)

40

30

20

10

300k

1M 10M

Frequency (Hz)

Av=4

Vcc=±6V, Rfb=620Ω, Load = 25Ω Vcc=±2.5V, R f b=910Ω, Load =10Ω IF Bw = 10Hz Smoothing=19.247MHz on 10ns/div scale

50M

14/27

TS615

C

t

INTERMODULATION DISTORTION PRODUCT

A non-ideal output of the amplifier can be described by the following development :

Vout C0C1VinC2V

++ +=

2

in

…

CnV

n

in

due to a non-linearity in the input-output amplitude transfer. In the case of the input is Vin=Asinωt, C is the DC component, C1(Vin) is the fundamental, C

is the amplitude of the harmonics of the output

n

signal V

out

.

A one-frequency (one-tone) input signal contributes to a harmonic distortion. A two-tones input signal contributes to a harmonic disto rtion and intermodulation product.

This intermodulation product or intermodulation distortion study of a two-tones input signal is the first step of the amplifier characterization of driving capability in the case of a multi-tone signal.

In this case :

C

+A

()

2

C

+

…

()

n

VinA

tsin B

ω

1

A

tsin B

ω

1

tsin B

ω

1

2

tsin+

ω

2

n

tsin+

ω

2

tsin+=

ω

2

In this expression, we recognize the second order intermodulation IM2 by the frequencies (ω and (ω IM3 by the frequencies (2ω

+2ω2) and (ω1+2ω2).

(−ω

1

) and the third order intermodulation

1+ω2

), (2ω1+ω2),

1-ω2

The measurement of the intermodulation product of the driver is achieved by using the driver as a

0

mixer by a summing amplifier configuration. By this way, the non-linearity problem of an external mixing device is avoided.

Figure 45 : Non-inve r t ing S umming Amp lif ier

1kΩ

49.9Ω

Vin1

50Ω

1:√2

North Hills

0315PB

400Ω

Vin2

50Ω

49.9Ω

1:√2

North Hills

0315PB

49.9Ω

400Ω

49.9Ω

1kΩ

11

+Vcc

+

1/2TS615

_

13

10

Rfb1

Rg1

Vout diff.

Rg2

Rfb2

_

1/2TS615

1kΩ

+

-Vcc

33

33Ω

49.9Ω

Ω

1-ω2

√2:1

100Ω 50Ω

North Hills

0315PB

)

V

outC0C1

+=

and :

C+

()

1

C

2



------- A

–

2+C2AB

3C3A2B

----------------------- -+2 2

3C3A2B

----------------------- -+ 2

3



A



tsi n B

ω

1

cos t B



2

()



C3A33

+





()





–

()



…

V

outC0C2

3

ω

A

()

A

ω

2

ω

1

–

ω1ω

()



3



ω

–

ω

1

ω

1

C

+

+=

t2A2B

2

tsin B

ω

1

tsin B

1

ω

2

t

cos– tcos

2

3

-------

¥+

4

tsin B33

1

tsin

-- -– 2

ω

2

1

–

2ω+

tsin

-- 2

2

n

V

()

n

in



A2B2+

---------------------



2



ω

tsin+

ω

2

tsin+

2

tcos+

ω

2

–

ω

()

1

2

tsin+

ω

2

+

ω1ω

()

ω

()

1

tsin 2AB

1

tsin

2

2ω+

tsin

2

ω

sin++sin+

2

The following graphs show the IM2 and the IM3 of the amplifier in different configuration. The two-tones input signal is achieved by the multisource generator Marconi 2026. Each tone has the same amplitude. The measurement is achieved by the spectrum analyzer HP3585A .

15/27

TS615

Figure 46 : Intermodulation vs. Output Amplitude

370kHz & 400kHz, AV=+1.5, Rfb=1kΩ, RL=14Ω diff.,VCC=±2.5V

IM2 30kHz

IM3 1140kHz, 1170kHz

-30

-40

-50

-60

IM3

-70

340kHz, 430kHz

-80

IM2 and IM3 (dBc)

-90

-100 012345678

IM2 770kHz

Differential Output Voltage (Vp-p)

Figure 47 : Intermodulation vs. Output Amplitude

370kHz & 400kHz, AV=+1.5, Rfb=1kΩ, RL=28Ω diff.,VCC=±2.5V

IM2 30kHz

IM2 770kHz

IM3 1140kHz, 1170kHz

-30

-40

-50

-60 IM3

340kHz, 430kHz

-70

-80

IM2 and IM3 (dBc)

-90

-100 012345678

Differential Output Voltage (Vp-p)

Figure 49 : Intermodulation vs. Load

370kHz & 400kHz, AV=+1.5, Rfb=1kΩ, Vout=6.5Vpp, VCC=±2.5V

-30

-40

-50

-60

-70

-80

IM2 and IM3 (dBc)

-90

-100

-110 0 20 40 60 80 100 120 140 160 180 200

IM3 340kHz, 430kHz, 1140kHz, 1170kHz

IM2 30kHz

IM2 770kHz

Differential Load (Ω)

Figure 50 : Intermodulation vs. Output Amplitude

100kHz & 110kHz, AV=+4, Rfb=620Ω, RL=200Ω diff.,VCC=±6V

IM3 320kHz

IM2 210kHz

-30

-40

-50

IM3 90kHz, 120kHz

-60

-70

-80

IM2 and IM3 (dBc)

-90

-100

-110 246810121416182022

IM3 310kHz

Differential Output Voltage (Vp-p)

Figure 48 : Intermodulation vs. Gain

370kHz & 400kHz, RL=20Ω diff., Vout=6Vpp, VCC=±2.5V

-30

-40

-50

-60

-70

-80

IM2 and IM3 (dBc)

-90

-100

-110

1.0 1.5 2.0 2.5 3.0 3.5 4.0

IM3 340kHz, 430kHz, 1140kHz, 1170kHz

IM2 30kHz

Closed Lo op Gain (Linear)

16/27

IM2 770kHz

Figure 51 : Intermodulation vs. Output Amplitude

100kHz & 110kHz, AV=+4, Rfb=620Ω, RL=50Ω diff., VCC=±6V

-30

-40

-50

-60

-70

-80

IM2 and IM3 (dBc)

-90

-100

-110 2 4 6 8 10121416182022

IM3 90kHz, 120kHz, 310kHz, 320kHz

Differential Output Voltage (Vp -p)

IM2 210kHz

TS615

Figure 52 : Intermodulation vs. Frequency Range

AV=+4, Rfb=620Ω, RL=50Ω diff., Vout=16Vp p, VCC=±6V

f1=400kHz f2=430kHz

f1=1MHz f2=1.1MHz

1.1M

-60

Quadratic Summation of all IM2 and IM3 components

generated by each two-tones signal

-65

-70 f1=100kHz

-75

f2=110kHz

-80

(dB)

-85

-90

-95

-100

f1=200kHz f2=230kHz

100k 200k 300k 400k 500k 600k 700k 800k 900k 1M 1M

Frequency (Hz)

Figure 53 : Intermodulation vs. Output Amplitude

370kHz & 400kHz, AV=+4, Rfb=620Ω, RL=200Ω diff.,VCC=±6V

-30

-40

-50

-60

-70 IM3

-80

340kHz, 430kHz

IM2 and IM3 (dBc)

-90

IM3 1140kHz, 1170kHz

IM2 770kHz

IM2 30kHz

Figure 54 : Intermodulation vs. Output Amplitude

370kHz & 400kHz, AV=+4, Rfb=620Ω, RL=50Ω diff., VCC=±6V

-30

-40

-50

-60

-70

-80

IM2 and IM3 (dBc)

-90

-100

-110 0246810121416182022

IM3 340kHz, 430kHz

Differential Output Voltage (Vp-p)

IM3 1140kHz, 1170kHz

IM2 30kHz

IM2 770kHz

-100

-110 0246810121416182022

Differential Output Voltage (Vp-p)

17/27

TS615

PRINTED CIRCUIT BOARD LAYOUT CONSIDERATIONS

In this range of frequency, printed circuit board parasites can affect the closed-loop performance.

The implementation of a proper ground plane in both sides of the PCB is mandatory to provide low inductance and low resistance common return. Most important for controlling the gain flatness and the bandwidth are stray capac itances at the output and inverting input. For minimizing the coupling, the space between signal lines and ground plane will be increased. Connec tions of the feedback component s mu st be as short as possible in order to decrease the associated inductance which affect high frequen cy gain errors. It is very important to choose external components as small as possible such as surface mounted devices, SMD, in orde r to minimize the size of all the DC and AC connections.

THERMAL INFORMATION

The TS615 is housed in an Exposed-Pad plastic package. As described on the figure 56, this package uses a lead frame upon which the dice is mounted. This lead frame is exposed as a thermal pad on the underside of the package. The thermal contact is direct with the di ce. This thermal path provide an excellent thermal performance.

Figure 55 : Exposed-Pad Package

DICE

Bottom Vie w

Side View

Bottom Vie w

DICE

Cross Section Vie w

Figure 56 : Evaluation Board

1

The thermal pad is electrically isolated from all pins in the package. It should be soldered to a copper area o f th e PC B underneath the pac kage. Through these thermal paths within this copper area, heat can b e conducted away from the pack age. In this case, the copp er area should be connected to (-

V

).

CC

18/27

Figure 57 : Schematic Diagram

J105

J106

J107

J108

J109

Differential Amplifier

J106

J109

R106

4

R107

R102 R1 01

R103

R108

R109

R104

R110

R105

R107

R102

R110

R105

4

+

1/2TS615

_

2

5

R114

R111

R112

R115

10

_

1/2TS615

13

+

11

R113

4

+

1/2TS615

_

2

5

R114

R111

R112

R115

10

_

1/2TS615

13

+

11

R113

TS615

Non-Inverting Amplifier

4

R107

J106

R118

R116

R119

R117

R118

R116

R119

R117

J110

R120

Inverting Amplifier

J111

R121

J110

R120

J111

R121

J108

Non-In ve r tin g S um m ing Amplif i er

J105

J106

R107

R102

R109

R104

R101

R106

R107

R102

4

+

1/2TS615

_

2

5

R114

R111

R115

_

10

1/2TS615

13

+

11

R113

4

+

1/2TS615

_

2

5

R114

R111

R118

R116

R119

R117

R118

R116

J110

R120

J111

R121

J110

R120

PowerSupply

C102

100nF

J101

J101 +Vcc

+Vcc

J102

J102 GND

GND

J103

-Vcc

C103

100nF

J10412

3

Power down

J112

+Vcc

C105

+Vcc

C101

100uF

C104

100uF

+Vcc

-Vcc

100nF

3

4

+

1/2TS615

_

5

C106

100nF

-Vcc

C107

100nF

+Vcc

_

10

1/2TS615

+

11

11 C108

C108

R122

-Vcc

6

2

1

-Vcc

12

14

-Vcc

-Vcc-Vcc

Expo sed-P ad

13

Differentia l Amplifier

R107

J106

J109

R107

R102

R110

R105

4

+

1/2TS615

_

2

5

R114

R111

R112

R115

10

_

1/2TS615

13

+

11

R113

R118

R116

R119

R117

J110

R120

J111

R121

19/27

TS615

Figure 58 : Component Locations - Top Side

Figure 59 : Component Locations - Bottom Side

Figure 60 : Top Side Board Layout

Figure 61 : Bottom Side Board Layout

20/27

TS615

g

)

NOISE MEASUREMENT Figure 62 : Noise Model

+

eN

+

TS615

_

R2

iN+

R3

N3

iN-

N2

R1

N1

output

HP3577

HP3577 Input noi s e:

Input noi s e: 8nV/√Hz

8nV/√Hz

eN : input voltage noise of the amplifier iNn : negative input current noise of the amplifier iNp : positive input current noise of the amplifier The closed loop gain is :

R

A

g1

V

The six noise sources are :

2

=

V

iNn R2×

54

V

=

V1eN

=

V3iNp R

4

V

61

V

×

2

R

------ -

–=

1

R



------ -

+



R

fb

--------- -+== R

2=

kTR

2

R



------ -

1

+



1

R

2

R



31

×

2 1

-------

+

××



1

R

4

1

kTR

4

3=

kTR

Assuming the thermal noise of a resistance R as:

4kTR F∆

with

∆F the specified bandwidth.

On 1Hz bandwidth the thermal noise is reduced to

4kTR

k is the Boltzmann’s constant equals to

1,374.10-23J/°K. T is the temperature (°K). The output noise eNo is c alculated using the Su-

perposition Theorem. But it is not the sum of all noise sources. The output noise is the square root of the sum of the square of each noise source.

No V12V22V32V42V52V6

+++++ eq1

2

,=

(

2

=

eNo2eN2g2iNn2R

2

R



------ -

+



4

× eq2(

1

R

+

kTR

×

21

× R

++

14

kTR

2

+

iNp

2

R



-------

+



1

R

3

× g2×

2

4

kTR

×

2

3

,

The input noise of the instrumentation must be extracted from the measured noise value. Th e real output noise value of the driver is:

eNo Measured

()

2

instrumentation

– eq3

()

2

,=

()

The input noise is called the Equivalent Input Noise as it is not directly measured but it is evaluated from the meas urement of the output divided by the closed loop gain (eNo/g). After simplification of the fourth and the fifth term of (eq2) we obtain:

2

=

eNo2eN2g2iNn2R

+

… g4kTR

× eq4(),

+

× R

21

×

2

R



------ -

+



1

R

2

+

iNp

2

4

kTR

×

3

× g2×

3

2

Measurement of eN:

We assume a short-circuit on the non-inverting input (R3=0). (eq4) comes:

=

eNo eN2g2iNn2R

+

× eq5(),

×

2

+

g4kTR

2

×

2

In order to easily extract the value of eN, the resistance R2 will be chosen as low as possible. In the other hand, the gain must be large enough. R1=10Ω, R2=910Ω, R3=0, Gai n=92 Equivalent Input Noise: 2.57nV/√Hz

Input Voltage Noise: eN=2.5nV/√Hz Measurement of iNn:

R3=0 and the output noise equation is still the (eq5). This time the gain must be decreased to decrease the thermal noise contribution. R1=100Ω, R2=910Ω, R3=0, Gain=10.1 Equivalent Input Noise: 3.40nV/√Hz

Negative Input Current Noise: iNn =21pA/√Hz Measurement of iNp:

To extract iNp from (eq3), a resist ance R3 is connected to the non-inverting input. The val ue of R3 must be chose n in o rder to keep i ts therm al noise contribution as low as possible against the iNp contribution. R1=100Ω, R2=910Ω, R3=100Ω, Gain=10.1 Equivalent Input Noise: 3.93nV/√Hz

Positive Input Current Noise: iNp=15pA/√Hz

Condit i ons: frequenc y=100kHz, V

Instrumentation: Spectrum Analyzer HP3585A (input noi se of the HP358 5A: 8nV/√Hz)

CC

=±2.5V

21/27

TS615

+

-VCC

+VCC

10µF

+

10nF

TS615

10µF +

10nF

-

+

-VCC

+VCC

10µF

+

10nF

TS615

10µF +

10nF

-

POWER SUPPLY BYPASSIN G

A proper power suppl y bypass ing com es very im portant for optimizing the pe rformanc e in high frequency range. Bypass capacitors should be placed as close as possible to the IC pins to improve high frequency bypassing. A capacitor greater than 1µF is necessary to minimize the distortion. For a better quality bypassing a capacitor of 10nF is added following the same condition of implementation. These bypass capacitors must be incorporated for the negative and the positive supply.

Figure 63 : Circuit for Power Supply Bypassing

SINGLE POWER SUPPLY

The following figure show the case of a 5V single power suppl y configuration

Figure 64 : Circuit for +5V single supply

+5V

10µF

+

IN

+5V

R1

R1 820Ω

820Ω

R2

820Ω

Rin

Rin 1kΩ

1kΩ

+ 1µF

10nF

+

½ TS615

_

G

R

+

G

C

R

910Ω

100µF

fb

OUT

Rs

Rload

necessary to assume a positive output dynamic range between 0V and + V ering the values of V

OH and VOL, the am pl ifier will

supply rails. Consid-

CC

provide an output dynamic from +0.5V to 10.6V on 25Ω load for a 12V supplying, from 0.45V to 3.8V on 10 Ω load for a 5V supplying.

The amplifier must be biased with a mid supply (nominally +V

/2), in order to maintain the DC

CC

component of the signa l at this val ue. Several options are possible to provide this bias supply (such as a virtual ground using an operational amplifier), or a two-resistance di vider which is the cheapest solution. A high resistance value is required to limit the current consumption. On the other hand, the current must be high enough to bias the non-inverting input of t he amplifier. If we consider this bias current (30µA max.) as the 1% o f the current through the resistance divider to keep a stable mid supply, two resistances of 2.2kΩ can be used i n the case of a 12V power supply and two resistances of 820Ω can be used in the case of a 5V power supply.

The input provides a hig h pass filter with a break frequency below 10Hz which is necessary to remove the original 0 volt DC component of the input signal, and to fix it at +V

CC

/2.

CHANNEL SEPARATION - CROSSTALK

The following figure show the crosstalk from an amplifier to a second amplifier. This phenomenon, accented in high frequ encies, is unavoidable and intrinsic of the circuit.

Nevertheless, the PCB layout has also an effect on the crosstalk level. Capacitive coupling between signal wires, distance between critical signal nodes, power supply bypassing, are the most significant points.

Figure 65 : Crosstalk vs. Frequency

AV=+4, Rfb=620Ω, VCC=±6V, Vout=2Vp

-50

-60

-70

-80

-90

The TS615 operates from 12V down to 5V power supplies. This is achieved with a d ual power s up-

ply of ±6V an d ±2.5V or a s ingle power s upply of 12V and 5V referenced to the grou nd. In t he cas e of this asymmet rical supp lying, a new biasing is

22/27

-100

CrossTalk (dB)

-110

-120

-130 10k 100k 1M 10M

Frequency (Hz)

TS615

0 1020304050

-6

-5

-4

-3

-2

-1

0

Vout

Vpdw

Disabled Output

Enabled Output

(Volts)

Time (µs)

POWER DOWN MODE BEHAVIOUR Figure 66 : Equivalent Schematic

V

cc +

3

.

+

Vcc - V

+

V

.

A1

_

cc -

A2

_

cc

+

..

.

pdw

R

1

14

.

12

-Vcc

-Vcc Rpdw

.

..

13

4

5

11

10

2

Ouput 1

Ouput 2

POWER

DOWN

pin6

POWER

DOWN

pin6

Please note that the short circuited output in power down mode is referenced to (-

V

). No problem

CC

appears when used in differential mode. Nevertheless, when used in single ended on a load referenced to GND, the (-

V

) level contributes to a

CC

current consumption through the load. As described on t he Figure 68, t he interest of featuring an output short circuit in power down mode is to keep the best impedance matchi ng between the system and the twisted pair telephone line when the modem is in sleep mode. By this way, the modem can be waked-up with a sign al from the line without any damage of this signal. This concept is particularly intended for the ADSL over voice modems, where the modem in sleep mode, must be waked-up by the phone call.

Figure 67 : Matching in Sleep Mode

Figure 68 : Standby Mode. Time On>Off

Figure 69 : Standby Mode. Time Off>On

1

0

Disabled Output

−1

−2

(Volts)

−3

−4

−5

−6 012345

Time (µs)

Enabled Output

Vout

Vpdw

Figure 70 : Standby Mode. Input/Output Isolation vs. Frequency

AV=+4, Rfb=620Ω, VCC=±6V, Vout=3Vp

Consumption=80µA

12.5Ω

TS615

POWER DOWN

5Ω max.

12.5Ω

The system can be waked-up

from the line

Matching

25Ω

Transformer

1:2

Line (100Ω)

0

-10

-20

-30

-40

-50

-60

-70

-80

Isolation (dB)

-90

-100

-110

-120

-130 10k 100k 1M 10M

Frequency (Hz)

23/27

TS615

CHOICE OF THE FEEDBACK CIRCUIT Table 71 : Closed-Loop Gain - Feedback Compo-

nents

V

CC

±6

±2.5

(V)

Gain

+1 750 +2 680 +4 620 +8 510

-1 680

-2 680

-4 620

-8 510 +1 1.1k +2 1k +4 910 +8 680

-1 1k

-2 1k

-4 910

-8 680

Rfb (Ω)

INVERTING AMPLIFIER BIASING

In this case a resistance is necessary to achieve a good input biasing, as R on (fig.30). This resistance is calculated by assuming the negative and positive input bias current. The aim is to make the compensation of the offset bias current which could affect the input offset voltage and t he output DC component . A ssumin g I b-, Ib+ , R

in, Rfb

and a zero volt output, the resistance R comes: R = R

in // Rfb .

Figure 72 : Compensation of the Input Bias Current

ACTIVE FI LT ER I N G

Figure 73 : Low-Pass Active Filtering. Sallen-Key

C1

1 R2

R

IN

The resistors R

fb and RG give d irectly the gain of

+

C2

RG

_

TS615

910Ω

R

fb

OUT

25Ω

the filter as a classical non-inverting amplification configuration :

R

A

V

g1

fb

--------- -+== R

g

Assuming the following expression as the response of the system:

Vout

T

j

ω

------------------Vin

j

ω

j

ω

---------------------------------------------== 12

------ -

ζ

ω

j

ω

g

c

j

ω()

--------------++ 2

ω

c

2

Ib-

24/27

Rin

Ib+

R

fb

the cutoff frequency is not gain depe ndent and it

Vcc+

TS615

Vcc+

Vcc-

Output

Load

_

+

R

comes:

------------------------------------- -=

ω

c

R1R2C1C2

the damping factor comes:

1

-- -

=

ζ

ω

()

cC1R1C1R2C2R1C1R1

2

1

g–++

The higher the gain the more sensitive the damping factor is. When the gain is hig her than 1 it is preferable to use som e very stable resistors and capacitors values.

In the case of R1=R2:

R

2C2C

------------------------------------=

ζ

2C1C

fb

--------- -–

1

R

g

2

TS615

INCREASING THE LINE LEVEL BY USING AN ACTIVE IMPEDANCE MATCHING

With a passive mat ching, the output signal amplitude of the driver must be twice the amplitude on the load. To go beyond this limitation an active matching impedance can be used. With this technique, it is possible to keep a good impedance matching with an amplitude on the load higher than the half of the output driver amplitude. This concept is shown in figure 74 for a differential line.

Figure 74 : T S615 as a differential line driver with an active impedance matching

µ

100n

Vcc+

1k

Vi

100n

10µ

1k

GND

Vi Vo

1/2 R1

+ _

R2

R3

Vcc/2

R5

Vcc+

Rs1

GND

Vo°

Vcc+

GND

Vo°

Rs2

R4

1

10n

Vo

1:n

Hybrid

&

RL

Transformer

100Ω

2Vi

Vi Vo°–

()

---------

---------------------------and

R1

As Vo

° equals Vo without load, the gain in this

R2

Vi Vo+

()

------------------------, R3

case becomes :

2R2

----------R1

R2

1

–

------- R3

R2

------- -++ R3

G

Vo noload

()

--------------------------------Vi

1

------------------------------------==

The gain, for the loaded system will be (eq1):

2R2

----------R1

R2

1

------- -– R3

R2

------- -++ R3

,==

()

GL

Vo withload

()

-------------------------------------Vi

1

-- -

------------------------------------ e q1

2

As shown in figure76, this system is an ideal generator with a synthesized impedance as the internal impedance of the system. From this, the output voltage becomes:

Vo ViG

()

RoIout

–= eq2

()

,

()

with Ro the synthesized impedance and Iout the output current. On the other hand Vo can be expressed as:

Component Calculation

Let us consider the equivalent c ircuit for a single ended configuration, Figure75.

Figure 75 : Single ended equivalent circuit

+

1/2

+

Vi

R1

R2

_

R3

Vo°

Rs1

-1

Vo

1/2

RL

Let us consider the unloaded system . Assumi ng the currents through R1, R2 and R3

as respectively:

2R2

----------R1

R2

------- -– R3

R2

------- -++ R3

Rs1Iout

----------------------- eq 3 1

R2

------- -– R3

,–=

()



Vi 1

Vo



------------------------------------------------

1

By identification of both equations (eq2) and (eq3), the synthesized impedance is, with Rs1=Rs2=Rs:

Ro

Rs

----------------- e q 4

,=

()

R2

1

------- -– R3

Figure 76 : Equivalent schematic. Ro is the synthesized impedance

Ro

Vi.Gi

Iout

1/2RL

25/27

TS615

Unlike the level Vo° required for a passive impedance, Vo us write Vo

° will be smaller than 2Vo in our case. Let

°=kVo with k the m at ching f actor vary -

ing between 1 and 2. Assum ing that the current through R3 is negligible, it comes the following resistance divider:

Ro

kVoRL

----------------------------- -= RL 2Rs1+

After choosing the k factor, Rs will equal to 1/2RL(k-1). A good impedance matching assume s:

1

Ro

-- -RL eq5

,=

()

2

From (eq4) and (eq5) it becomes:

R2

------- -1 R3

2Rs

----------- eq 6

,–=

()

RL

By fixing an arbitrary value of R2, (eq6) gives:

R3

R2

-------------------- -=

2Rs

1

-----------– RL

Finally, the values of R2 and R3 allow us to extract R1 from (eq1), and it comes:

----------------------------------------------------------- eq 7

R1

21

2R2

R2



–

GL 1–

------- -



R3

R2

–

------- R3

,=

()

with GL the required gain. Figure 77 : Components Calculation for

Impedance Matching Implem entat ion

GL (gain for the loaded system)

R1 2R2/[2(1-R2/R3)GL-1-R2/R3] R2 (=R4) Abritra ry fixed R3 (=R5) R2/(1-Rs/0.5RL)

Rs 0.5RL(k-1)

Load view ed by

each driv er

GL is fixed for the application requirements GL=Vo/Vi=0.5(1+2R2/R1+R2/R3)/(1-R2/R3)

kRL/2

26/27

PACKAGE MECHANICAL DATA

14 PINS - THIN SHRINK SMALL OUT L INE PACKAGE (TSSOP Exposed-Pad)

c

k

0,25 mm

GAUGE PLANE

L

L1

C

PLANE

SEATING

E

E2

89

e

E1

A

A2

A1

TS615

D1

1

PIN 1 IDENTIFICATION

14

D

b

C

aaa

Millimeters Inches

Dimensions

Min. Typ. Max. Min. Typ. Max.

A 1.200 0.047 A1 0.150 0.006 A2 0.800 1.000 1.050 0.031 0.039 0.041

b 0.190 0.300 0.007 0.012

c 0.090 0.200 0.004 0.008

D 4.900 5.000 5.100 0.193 0.197 0.201

D1 3.000 1.18

E 6.200 6.400 6.600 0.244 0.252 0.260 E1 4.300 4.400 4.500 0.169 0.173 0.177 E2 3.000 1.18

e 0.650 0.026

L 0.450 0.600 0.750 0.018 0.024 0.030 L1 1.000 0.039

k0d 8d0d 8d

aaa 0.100 0.004

Information furnished is bel ieved to be accurate and reliable. However, STMicroe lectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No li cense is granted by imp lication or otherwise under a ny patent or patent rig hts of STMicroelectronics. Spec ific at ions mentioned in this publication ar e subject to change without notice. This publication supersedes and replaces all information previously supplied. S TMicroelectronics products are not authorized for use as critica l components in life suppo rt devices or systems without express written approval of STMicroelectronics.

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27/27

SGS Thomson Microelectronics TS615IPWT, TS615 Datasheet

Specifications and Main Features

Frequently Asked Questions

User Manual