SC4503
(A)
1.3MHz Step-Up Switching
Regulator with 1.4A Switch
POWER MANAGEMENT
Description
The SC4503 is a 1.3MHz current-mode step-up switching regulator with an integrated 1.4A power transistor.
Its high switching frequency allows the use of tiny surface-mount external passive components. The SC4503
features a combined shutdown and soft-start pin. The
optional soft-start function eliminates high input current
and output overshoot during start-up. The internal compensation network accommodates a wide range of voltage conversion ratios. The internal switch is rated at 34V
making the device suitable for high voltage applications
such as Boost and SEPIC.
The SC4503 is available in low-profi le 5-lead TSOT-23 and
8-lead 2X2mm MLPD-W packages. The SC4503’s low
shutdown current (< 1μA), high frequency operation and
small size make it suitable for portable applications.
Features
Features
Low Saturation Voltage Switch: 260mV at 1.4A
1.3MHz Constant Switching Frequency
Peak Current-mode Control
Internal Compensation
Programmable Soft-Start
Input Voltage Range From 2.5V to 20V
Output Voltage up to 27V
Uses Small Inductors and Ceramic Capacitors
Low Shutdown Current (< 1μA)
Low Profi le 5-Lead TSOT-23 and 8-Lead 2X2mm
MLPD-W packages
Fully WEEE and RohS compliant
Applications
Local DC-DC Converters
TFT Bias Supplies
XDSL Power Supplies
Medical Equipment
Digital Cameras
Portable Devices
White LED Drivers
Typical Application Circuit
Typical Application Circuit
VIN
5V
C1
1µF
OFF
ON
4
L1
4.7µH
5
IN SW
SC4503
SHDN/SS
GND
2
1
FB
C1: Murata GRM188R61A105K
C2: Murata GRM21BR61C475K
L1: Sumida CDC5D23B-4R7
Figure 1(a). 5V to 12V Boost Converter
May 4, 2007
D1
10BQ015
3
C4
15pF
R1
432k
R2
49.9k
VOUT
12V, 0.5A
C2
4.7µF
95
90
85
80
75
70
Efficiency (%)
65
60
55
50
Efficiency vs Load Current
1.3MHz
V
= 12V
OUT
0.001 0.010 0.100 1.000
Load Current
Figure 1(b). Effi ciency of the 5V to 12V Boost Converter
1
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SC4503
POWER MANAGEMENT
Absolute Maximum Ratings
Absolute Maximum Ratings
Exceeding the specifi cations below may result in permanent damage to the device or device malfunction. Operation outside of the parameters specifi ed in the
Electrical Characteristics section is not recommended.
Parameter Symbol Maximum Units
Supply Voltage V
SW Voltage V
FB Voltages V
SHDN/SS Voltage V
Thermal Resistance Junction to Ambient (TSOT - 23) θ
Thermal Resistance Junction to Ambient (2X2 mm MLPD-W) θ
Maximum Junction Temperature T
Storage Temperature Range T
Lead Temperature (Soldering)10 sec (TSOT - 23) T
Peak IR Refl ow Temperature (2X2mm MLPD-W) T
IN
SW
FB
SHDN
JA
JA
J
STG
LEAD
IR
-0.3 to 20
-0.3 to 34
-0.3 to VIN +0.3
-0.3 to VIN +1
191* °C/W
78* °C/W
150
-65 to +150
260
260
ESD Rating (Human Body Model) ESD 2000 V
*Calculated from package in still air, mounted to 3” x 4.5”, 4 layer FR4 PCB with thermal vias under the exposed pad as per JESD51 standards.
Electrical Characteristics
Unless specifi ed: VIN = V
= 3V, -40°C < TA = TJ < 85°C
SHDN/SS
V
°C
Parameter Conditions Min Typ Max Units
Under-Voltage Lockout Threshold 2.2 2.5
VMaximum Operating Voltage 20
Feedback Voltage 1.225 1.250 1.275
Feedback Line Voltage Regulation 2.5V < VIN < 20V 0.02 %/V
FB Pin Bias Current -25 -50 nA
Switching Frequency 1.15 1.30 1.55 MHz
Minimum Duty Cycle 0
%
Maximum Duty Cycle 86 90
Switch Current Limit 1.4 1.9 2.5 A
Switch Saturation Voltage ISW = 1.4A 260 430 mV
Switch Leakage Current VSW = 5V 0.01 1 µA
VIN Quiescent Supply Current V
= 2V, VFB = 1.5V (not switching) 0.8 1.1 mA
SHDN/SS
VIN Supply Current in Shutdown V
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= 0 0.01 1 µA
SHDN/SS
2
SC4503
POWER MANAGEMENT
Electrical Characteristics (Cont.)
Unless specifi ed: VIN = V
Parameter Conditions Min Typ Max Units
SHDN/SS Switching Threshold VFB = 0V 1.4 V
Shutdown Input High Voltage 2
Shutdown Input Low Voltage 0.4
SHDN/SS Pin Bias Current
Thermal Shutdown Temperature 155
Thermal Shutdown Hysteresis 10
Pin Confi guration - TSOT - 23
= 3V, -40°C < TA = TJ < 85°C
SHDN/SS
V
SHDN/SS
V
= 2V 22 50
SHDN/SS
= 1.8V 20 45
= 0V 0.1
SHDN/SS
Ordering Information
µAV
°C
V
Top View
SW
GND
FB
1
2
3
5
4
IN
SHDN/SS
5-LEAD TSOT-23
Pin Descriptions - TSOT -23
Pin Pin Name Pin Functions
1SW
2 GND Ground. Tie to ground plane.
3FB
4 SHDN/SS
Collector of the internal power transistor. Connect to the boost inductor and the freewheeling
diode. The maximum switching voltage spike at this pin should be limited to 34V.
The inverting input of the error amplifi er. Tie to an external resistive divider to set the output volt-
age.
Shutdown and Soft-start Pin. Pulling this pin below 0.4 shuts down the converter. Applying more
than 2V at this pin enables the SC4503. An external resistor and an external capacitor connected to this pin soft-start the switching regulator. The SC4503 will try to pull the SHDN/SS pin
below its 1.4V switching threshold regardless of the external circuit attached to the pin if VIN
is below the under-voltage lockout threshold. Tie this pin through an optional resistor to IN or
to the output of a controlling logic gate if soft-start is not used. See Applications Information for
more details.
Device
(1,2)
Top Mark Package
SC4503TSKTRT BH00 TSOT-23
SC4503EVB Evaluation Board
Notes:
(1) Available in tape and reel only. A reel contains 3,000 devices.
(2) Available in lead-free package only. Device is WEEE and
RoHS compliant.
5IN
2007 Semtech Corp. www.semtech.com
Power Supply Pin. Bypassed with capacitor close to the pin.
3
POWER MANAGEMENT
SC4503
Pin Confi guration - 2mm X 2mm MLPD
Top View
8
SW
SW
SW
SW
SHDN/SS
SHDN/SS
1
1
2
2
3
3
IN
IN
4
4
8
NC
NC
7
7
GND
GND
6
6
GND
GND
5
5
FB
FB
8-LEAD 2X2mm MLPD-W
Pin Descriptions - 2X2mm MLPD-W
Pin Pin Name Pin Functions
Collector of the internal power transistor. Connect to the boost inductor and the free-
1,2 SW
wheeling diode. The maximum switching voltage spike at this pin should be limited to
34V.
Ordering Information
Device
SC4503WLTRT E00
SC4503_MLPD EVB Evaluation Board
Notes:
(1) Available in tape and reel only. A reel contains 3,000 devices.
(2) Available in lead-free package only. Device is WEEE and
RoHS compliant.
(1,2)
Top Mark Package
2mmX2mm
MLPD-W
3 IN Power Supply Pin. Bypassed with capacitor close to the pin.
Shutdown and Soft-start Pin. Pulling this pin below 0.4 shuts down the converter. Applying more than 2V at this pin enables the SC4503. An external resistor and an external
capacitor connected to this pin soft-start the switching regulator. The SC4503 will try
4 SHDN/SS
to pull the SHDN/SS pin below its 1.4V switching threshold regardless of the external
circuit attached to the pin if VIN is below the under-voltage lockout threshold. Tie this pin
through an optional resistor to IN or to the output of a controlling logic gate if soft-start is
not used. See Applications Information for more details.
5FB
The inverting input of the error amplifi er. Tie to an external resistive divider to set the
output voltage.
6,7 GND Ground. Tie to ground plane.
8 N.C. No Connection.
EDP Solder to the ground plane of the PCB.
2007 Semtech Corp. www.semtech.com
4
SC4503
4
+
+
+
J
POWER MANAGEMENT
Block Diagram
IN
5
SHDN/SS
FB
2
VOLTAGE
REFERENCE
1.25V
Q1
+
EA
-
THERMAL
SHUTDOWN
REF NOT READY
T > 155°°°°C
CLK
-
PWM
SW
1
+
Z1
1V
-
Q2
R
Q
S
ILIM
-
-
Q3
R
SENSE
OSCILLATOR
SLOPE COMP
Figure 2. SC4503 Block Diagram
ΣΣΣΣ
+
ISEN
-
2
GND
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5
POWER MANAGEMENT
Typical Characteristics
SC4503
FB Voltage vs T emperature
1.30
1.25
1.20
FB Voltage (V)
1.15
1.10
-50-25 0 255075100125
Temperature (°C)
VIN Under-voltage Lockout
Threshold vs Temperature
2.6
2.4
2.2
Switching Freque ncy
vs Temperature
1.5
1.4
1.3
1.2
Frequency (MHz)
1.1
1.0
-50-250 255075100125
Temperature (°C)
Switch Current Limit
vs Temperature
2.0
1.8
1.6
2.0
UVLO Threshold (V)
1.8
1.6
-50-250 255075100125
Temperature (°C)
Switch Saturation Voltage
vs Switch Current
400
125°C
300
(mV)
200
CESAT
V
100
0
0.0 0.5 1.0 1.5 2.0
25°C
-40°C
Switch Current (A)
1.4
Current Limit (A)
1.2
V
SHDN /SS
1.0
-50-25 0 255075100125
Temperature (°C)
VIN Quiescent Current
vs Temperature
0.80
0.75
0.70
Current (mA)
IN
V
0.65
VFB = 1.5V
0.60
-50-25 0 255075100125
Temperature (°C)
= 3V
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6
SC4503
POWER MANAGEMENT
Typical Characteristics (Cont.)
Shutdown Pin Current
vs Shutdown Pin Voltage
70
60
A)
µ
µ
µ
µ
50
40
30
20
Shutdown Pin Current (
10
0
0 5 10 15 20
Shutdown Pin Voltage (V)
25°C
85°C
VIN Quiescent Current
vs Shutdown Pin Voltage
1000
VIN = 3V VFB = 1.5V
800
A)
µ
µ
µ
µ
600
125°C
400
Current (
IN
V
200
-40°C
25°C
-40°C
Shutdown Pin Current
vs Shutdown Pin Voltage
50
A)
40
µ
µ
µ
µ
-40°C
30
20
10
Shutdown Pin Current (
0
0.0 0.5 1.0 1.5 2.0 2.5 3.0
Shutdown Pin Voltage (V)
Shutdown Pin
Thresholds vs Temperature
1.5
1.0
0.5
SHDN Thresholds (V)
Shutting Down To IIN < 1µA
85°C
Switching
25°C
0
0.0 0.5 1.0 1.5 2.0
Shutdown Pin Voltage (V)
Switch Current Limit
vs Shutdown Pin Voltage
2.5
D = 50%
2.0
1.5
1.0
Current limit (A)
0.5
0.0
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-40°C
25°C
1.2 1.4 1.6 1.8 2.0
Shutdown Pin Voltage (V)
85°C
7
0.0
-50-250 255075100125
Temperature (°C)
Switch Current Limit
vs Shutdown Pin Voltage
2.5
D = 80%
2.0
1.5
1.0
Current limit (A)
0.5
0.0
1.2 1.4 1.6 1. 8 2.0
-40°C
25°C
Shutdown Pin Voltage (V)
85°C
POWER MANAGEMENT
(
)
(
)
Applications Information
SC4503
Operation
The SC4503 is a 1.3MHz peak current-mode step-up
switching regulator with an integrated 1.4A (minimum)
power transistor. Referring to the block diagram, Figure
2, the clock CLK resets the latch and blanks the power
transistor Q
conduction. Q3 is switched on at the trailing
3
edge of the clock.
Switch current is sensed with an integrated sense resistor.
The sensed current is summed with the slope-compensating ramp and fed into the modulating ramp input of the
PWM comparator. The latch is set and Q3 conduction is
terminated when the modulating ramp intersects the error
amplifi er (EA) output. If the switch current exceeds 1.9A (the
typical current-limit), then the current-limit comparator ILIM
will set the latch and turn off Q3. Due to separate pulsewidth modulating and current limiting paths, cycle-by-cycle
current limiting is not affected by slope compensation.
The current-mode switching regulator is a dual-loop feedback control system. In the inner current loop the EA output
controls the peak inductor current. In the outer loop, the
error amplifi er regulates the output voltage. The double
reactive poles of the output LC fi lter are reduced to a single
real pole by the inner current loop, allowing the internal loop
compensation network to accommodate a wide range of
input and output voltages.
clamped by D1 and Q1, follows the voltage at the
SSSHDN
pin. The input inductor current, which is in turn controlled
by the error amplifi er output, also ramps up gradually.
Soft-starting the SC4503 in this manner eliminates high
input current and output overshoot. Under fault condition
(V
< 2.2V or over-temperature), the soft-start capacitor is
IN
discharged to 1V. When the fault condition disappears, the
converter again undergoes soft-start.
Setting the Output Voltage
An external resistive divider R1 and R2 with its center tap
tied to the FB pin (Figure 3) sets the output voltage.
9
§
287
55
¨
©
Figure 3. R
·
−=
¸
9
¹
VOUT
VOUT
VOUT
VOUT
SC4503
SC4503
SC4503
R1
R1
R1
R1
25nA
25nA
25nA
25nA
R2
R2
R2
R2
- R2 Divider Sets the Output Voltage
1
SC4503
FB3
FB3
FB3
FB3
(1)
SSSHDN
Applying 0.9V at the
ence. The signal “REF NOT READY” does not go low until
V
exceeds its under-voltage lockout threshold (typically
IN
pin enables the voltage refer-
2.2V). Assume that an external resistor is placed between
SSSHDN
the IN and the
reference is enabled when the
0.9V. Before V
reaches 2.2V, “REF NOT READY” is high.
IN
pins during startup. The voltage
SSSHDN
voltage rises to
Q2 turns on and the Zener diode Z1 loosely regulates the
SSSHDN
voltage to 1V (above the reference enabling voltage). The optional external resistor limits the current drawn
during under-voltage lockout.
When VIN exceeds 2.2V, “REF NOT READY” goes low. Q2 turns
SSSHDN
off, releasing
from the
. If an external capacitor is connected
SSSHDN
pin to the ground, the
SSSHDN
voltage
The input bias current of the error amplifi er will introduce
an error of:
•°«•
9
∆
287
9
−=
287
The percentage error of a V
55Q$
9
= 5V converter with R1 =
OUT
(2)
100kΩ and R2 = 301kΩ is
NNQ$
9
∆
287
9
This error is much less than the ratio tolerance resulting
−=
287
•°«•
−=
9
from the use of 1% resistors in the divider string.
will ramp up slowly. The error amplifi er output, which is
2007 Semtech Corp. www.semtech.com
8
SC4503
POWER MANAGEMENT
Applications Information (Cont.)
Duty Cycle
The duty cycle D of a boost converter in continuous-conduction mode (CCM) is:
9
'
=
where V
CESAT
age drop across the rectifying diode.
Maximum Output Current
In a boost switching regulator the inductor is connected
to the input. The inductor DC current is the input current.
When the power switch is turned on, the inductor current
fl ows into the switch. When the power switch is off, the
inductor current fl ows through the rectifying diode to the
output. The output current is the average diode current.
The diode current waveform is trapezoidal with pulse width
(1 – D)T (see Figure 4). The output current available from
Figure 4. Current Waveforms in a Boost ConverterFigure 4. Current Waveforms in a Boost Converter
Figure 4. Current Waveforms in a Boost ConverterFigure 4. Current Waveforms in a Boost Converter
a boost converter therefore depends on the converter operating duty cycle. The power switch current in the SC4503 is
internally limited to at least 1.4A. This is also the maximum
peak inductor or the peak input current. By estimating the
conduction losses in both the switch and the diode, an
expression of the maximum available output current of a
boost converter can be derived:
,1
−
−
9
+
&(6$7
+
99
'287
99
'287
is the switch saturation voltage and VD is volt-
I
I
I
I
IN
IN
IN
IN
Inductor
Inductor
Inductor
Inductor
Current
Current
Current
Current
0
0
0
0
0
0
0
0
ON ONOFF
ON ONOFF
ON ONOFF
ON ONOFF
(1-D)TDT
(1-D)TDT
(1-D)TDT
(1-D)TDT
ON ONOFFON OFF
ON ONOFFON OFF
ON ONOFFON OFF
ON ONOFFON OFF
Switch Current
Switch Current
Switch Current
Switch Current
Diode Current
Diode Current
Diode Current
Diode Current
I
I
I
I
OUT
OUT
OUT
OUT
(3)
where I
It is worth noting that I
ratio
derivation. Equation (4) therefore over-estimates the
maximum output current, however it is a useful fi rst-order
approximation.
Using V
the maximum output current for three VIN and V
nations are tabulated (Table 1).
Maximum Duty-Cycle Limitation
The power transistor in the SC4503 is turned off every
switching period for 80ns. This minimum off time limits the
maximum duty cycle of the regulator. A boost converter with
high
cycle. If the required duty cycle is higher than the attainable maximum, then the converter will operate in dropout.
(Dropout is a condition in which the regulator cannot attain
its set output voltage below current limit.)
Note: dropout can occur when operating at low input voltages (<3V) and with off times approaching 100ns. Shorten
the PCB trace between the power source and the device
input pin, as line drop may be a signifi cant percentage of
the input voltage. A regulator in dropout may appear as
if it is in current limit. The cycle-by-cycle current limit of
the SC4503 is duty-cycle and input voltage invariant and
should be at least 1.4A. If the converter output is below
its set value and switch current limit is not reached (1.4A),
then the converter is likely in dropout.
is the switch current limit.
LIM
is directly proportional to the
9
,1
and that switching losses are neglected in its
9
287
= 0.3V, VD = 0.5V and I
CESAT
VIN (V) V
VIN (V) V
VIN (V) V
VIN (V) V
3.3 12 0.754 0.34
3.3 12 0.754 0.34
3.3 12 0.754 0.34
3.3 12 0.754 0.34
3.3 5 0.423 0.80
3.3 5 0.423 0.80
3.3 5 0.423 0.80
3.3 5 0.423 0.80
5 12 0.615 0.53
5 12 0.615 0.53
5 12 0.615 0.53
5 12 0.615 0.53
Table 1. Calculated Maximum Output Currents
Table 1. Calculated Maximum Output Currents
9
287
ratio requires long switch on time and high duty
9
,1
OUTMAX
LIM
(V) D I
(V) D I
(V) D I
(V) D I
OUT
OUT
OUT
OUT
=1.4A in (3) and (4),
OUT
(A)
(A)
(A)
(A)
OUT
OUT
OUT
OUT
combi-
ª
9,
,
2870$;
2007 Semtech Corp. www.semtech.com
,1/,0
9
287
'
«
¬
−−=
()
−−
9
,1
º
99'9
&(6$7''
»
¼
Example
when boosting from a single Li-ion cell.
(4)
Equation (3) can be re-arranged as:
9
: Determine the highest attainable output voltage
POWER MANAGEMENT
9
(
)
(
)
Applications Information (Cont.)
9
−
'
§
¨
¨
©
&(6$7,1
−
'
=0.5V, V
D
−
OUT
−
99'
&(6$7,1
I/
−=∆
9
'
=0.3V and D=0.86 in (5),
CESAT
•−
. Sensed switch current ramp modulates
·
9
,1,1
¸
¸
+
99
'287
¹
9
=−
CESAT
,
(5)
(6)
(7)
9 −
=
287
Assuming that the voltage of a nearly discharged Li-ion cell
is 2.6V. Using V
9
<
287
Transient headroom requirement further reduces the maximum achievable output voltage to below 16V.
Minimum Controllable On-Time
The operating duty cycle of a boost converter decreases as
VIN approaches V
the pulse width in a current-mode switching regulator. This
current ramp is absent unless the switch is turned on. The
intersection of this ramp with the error amplifi er output
determines the switch on-time. The propagation delay
time required to immediately turn off the switch after it is
turned on is the minimum controllable on time. Measured
minimum on time of the SC4503 is load-dependent and
ranges from 180ns to 220ns at room temperature. The
switch in the SC4503 is either not turned on, or, for at least
this minimum. If the regulator requires a switch on-time
less than this controllable minimum, then it will either skip
cycles or start to jitter.
Inductor Selection
The inductor ripple current ΔIL of a boost converter in continuous-conduction mode is
=∆
,
/
where f is the switching frequency and L is the inductance.
Substituting (3) into (6) and neglecting V
9
,
/
In current-mode control, the slope of the modulating
(sensed switch current) ramp should be steep enough to
I/
SC4503
lessen jittery tendency but not so steep that large fl ux swing
decreases effi ciency. For continuous-conduction mode
operation, inductor ripple current ΔIL between 0.35A and
0.6A is a good compromise. Setting ΔIL = 0.43A, VD = 0.5V
and f = 1.3MHz in (7),
§
9
,1
¨
−
¨
∆
,I
/
©
=
/
where L is in μH.
Equation (7) shows that for a given V
when
=
9
,1
+
choose L based on the nominal input voltage.
The saturation current of the inductor should be 20-30%
higher than the peak current limit (1.9 A). Low-cost powder
iron cores are not suitable for high-frequency switching
power supplies due to their high core losses. Inductors
with ferrite cores should be used.
Discontinuous-Conduction Mode
The output-to-input voltage conversion ratio
continuous-conduction mode is limited by the maximum
duty cycle D
0
<
0
<
:
MAX
=
=
'
−
'
−
0$;
0$;
Higher voltage conversion ratios can be achieved by operating the boost converter in full-time discontinuous-con-
duction mode (DCM). Defi ne
output load resistance. The following inequalities must be
satisfi ed for DCM operation:
/I −
5
0
<
0
and,
9
,
287
287
5
<=
0
·
9
,1
¸
¸
+
99
'287
¹
99
'287
. If VIN varies over a wide range, then
−
−
§
9
¨
−=
¨
=
=
287
©
, ΔIL is the highest
OUT
·
9
,1,1
¸
(8)
¸
9
+
¹
9
287
0 =
9
,1
V
OUT
R = as the equivalent
I
OUT
$
in
(9)
(10)
2007 Semtech Corp. www.semtech.com
10
SC4503
POWER MANAGEMENT
Applications Information (Cont.)
Switch on duty ratio in DCM is given by,
' −=
/I
5
Higher input current ripples and lower output current are
the drawbacks of DCM operation.
Input Capacitor
The input current in a boost converter is the inductor current, which is continuous with low RMS current ripples. A
2.2-4.7µF ceramic input capacitor is adequate for most
applications.
Output Capacitor
Both ceramic and low ESR tantalum capacitors can be
used as output fi ltering capacitors. Multi-layer ceramic
capacitors, due to their extremely low ESR (<5mΩ), are
the best choice. Use ceramic capacitors with stable
temperature and voltage characteristics. One may be
tempted to use Z5U and Y5V ceramic capacitors for output
fi ltering because of their high capacitance density and
small sizes. However these types of capacitors have high
temperature and high voltage coeffi cients. For example,
the capacitance of a Z5U capacitor can drop below 60%
of its room temperature value at –25°C and 90°C. X5R
ceramic capacitors, which have stable temperature and
voltage coeffi cients, are the preferred type.
The diode current waveform in Figure 4 is discontinuous
with high ripple-content. Unlike a buck converter in which
the inductor ripple current
voltage. The output ripple voltage of a boost regulator is
much higher and is determined by the absolute inductor
current. Decreasing the inductor ripple current does not
reduce the output ripple voltage appreciably. The current
flowing in the output filter capacitor is the difference
between the diode current and the output current. This
capacitor current has a RMS value of:
9
287
287
9
,1
,
If a tantalum capacitor is used, then its ripple current rating
in addition to its ESR will need to be considered.
00
∆IL determines the output ripple
−
(11)
(12)
When the switch is turned on, the output capacitor supplies
the load current I
due to charging and discharging of the output capacitor
is therefore:
287
287
&
287
9 =∆
For most applications, a 10-22µF ceramic capacitor is suffi cient for output fi ltering. It is worth noting that the output
ripple voltage due to discharging of a 10µF ceramic capacitor (13) is higher than that due to its ESR.
Rectifying Diode
For high effi ciency, Schottky barrier diodes should be used
as rectifying diodes for the SC4503. These diodes should
have an average forward current rating at least equal to the
output current and a reverse blocking voltage of at least
a few volts higher than the output voltage. For switching
regulators operating at low duty cycles (i.e. low output
voltage to input voltage conversion ratios), it is benefi cial
to use rectifying diodes with somewhat higher average current ratings (thus lower forward voltages). This is because
the diode conduction interval is much longer than that of
the transistor. Converter effi ciency will be improved if the
voltage drop across the diode is lower.
The rectifying diodes should be placed close to the SW
pin of the SC4503 to minimize ringing due to trace inductance. Surface-mount equivalents of 1N5817 and 1N5818,
MBRM120, MBR0520L, ZHCS400, 10BQ015 and equivalent are suitable.
Shutdown and Soft-Start
The shutdown (
driven from a logic gate with VOH>2V, the
functions as an on/off input to the SC4503. When the
shutdown pin is below 2V, it clamps the error amplifi er
output to
Connecting R
9
SS
the voltage rise at the pin during start-up. This forces the
peak inductor current (hence the input current) to follow a
slow ramp, thus achieving soft-start.
(Figure 4). The output ripple voltage
OUT
'7,
) pin is a dual function pin. When
SSSHDN SSSHDN
and reduces the switch current limit.
666+'19666+'1
and CSS to the
SSSHDN SSSHDN
pin (Figure 5) slows
SSSHDN SSSHDN
(13)
pin
2007 Semtech Corp. www.semtech.com
11
POWER MANAGEMENT
Applications Information (Cont.)
The minimum
graph “Switch Current Limit vs. Shutdown Pin Voltage” in
the “Typical Characteristics” shows that the
voltage needs to be at least 2V for the SC4503 to deliver
its rated power. The effect of the
SC4503 is analog between 1.4V and 2V. Within this range
the switch current limit is determined not by ILIM but instead by the PWM signal path (see Figure 2). Moreover it
varies with duty cycle and the shutdown pin voltage.
SSSHDN SSSHDN
voltage for switching is 1.4V. The
SSSHDN SSSHDN
SSSHDN SSSHDN
voltage on the
pin
SC4503
Pulling the
drawing less than 1µA from the input power supply. For
voltages above 2V and below 0.4V, the
regarded as a digital on/off input. Figure 5 shows several
ways of interfacing the control logic to the shutdown pin. In
Figure 5(a) soft-start is not used and the logic gate drives
the shutdown pin through a small ( ≈ 1kΩ ) optional resistor
RSS. RSS limits the current drawn by the SC4503 internal
pin below 0.4V shuts down the SC4503,
SSSHDN SSSHDN
SSSHDN SSSHDN
pin can be
VOH> 2V
VOL< 0.4V
VOL< 0.4V
V
IN
End of Soft-start
V
> 2V
SHDN/SS
R
End of Soft-start
V
SHDN/SS
R
SS
C
SS
R
SS
LIM
> 2V
I
SHDN/SS
IN
SC4503
SHDN/SS
(a)
IN
SC4503
SHDN/SS
1.7V < VOH< 2V
(c)
IN
SC4503
SHDN/SS
C
SS
V
IN
End of Soft-start
V
> 2V
SHDN/SS
V
IN
VOL≈ 0
D
SS
CMDSH-3
V
IN
IN
1N4148
VOH> V
IN
R
SS
C
SS
SC4503
SHDN/SS
(b)
IN
R
I
SHDN/SS
C
SS
SS
SC4503
SHDN/SS
(d)
IN
SC4502
C
SHDN/SS
SS
R
SS
Figure 5.
(e)
Methods of Driving the Shutdown Pin and Soft-starting the SC4503
(a) Directly Driven from a Logic Gate. R
Limits the Gate Output Current during Fault,
LIM
(f)
(b) Soft-start Only,
(c) Driven from a Logic Gate with Soft-start,
(d) Driven from a Logic Gate with Soft-start (1.7V < VOH < 2V),
(e) Driven from an Open-collector NPN Transistor with Soft-start and
(f) Driven from a Logic Gate (whose VOH > VIN) with Soft-start.
2007 Semtech Corp. www.semtech.com
12
SC4503
ω
ω
POWER MANAGEMENT
Applications Information (Cont.)
circuit from the driving logic gate during fault condition.
In Figure 5(f) the shutdown pin is driven from a logic gate
whose VOH is higher than the supply voltage to the SC4503.
The diode clamps the maximum shutdown pin voltage to
one diode voltage above the input power supply.
During soft-start, CSS is charged by the difference between
the RSS current and the shutdown pin current,
steady state, the voltage drop across RSS reduces the shutdown pin voltage according to the following equation:
,599 −=
,599 −=
66(1
666+'1
666+'1
66(1
666+'1
666+'1
In order for the SC4503 to achieve its rated switch current,
9
puts an upper limit on R
voltage applied to R
50µA with
The largest R
must be greater than 2V in steady state. This
666+'19666+'1
= 99
=
666+'1
666+'1
can be found using (14):
SS
9
−
9
−
0,1(1
5
<
5
<
66
66
0,1(1
$
µ
$
µ
for a given enable voltage V
SS
). The maximum specifi ed
SS
99
(see “Electrical Characteristics”).
If the enable signal is less than 2V, then the interfacing
options shown in Figures 5(d) and 5(e) will be preferred. The
methods shown in Figures 5(a) and 5(c) can still be used
however the switch current limit will be reduced. Variations
,
of
and switch current limit with
666+'1,666+'1
and temperature are shown in the “Typical Characteristics”.
Shutdown pin current decreases as temperature increases.
Switch current limit at a given
9
also decreases as
666+'19666+'1
temperature rises. Lower shutdown pin current fl owing
through R
at high temperature results in higher shutdown
SS
pin voltage. However reduction in switch current limit (at
a given
9
) at high temperature is the dominant
666+'19666+'1
effect.
,
,
SSSHDN SSSHDN
pin voltage
666+'1,666+'1
. In
(14)
EN
is
666+'1,666+'1
(=
,
,
287
=
=
FB
FB
FB
FB
1.252V
1.252V
1.252V
1.252V
287
&9
&9
−=ω
−=ω
&5
&5
=
=
Output fi lter pole,
S
S
Compensating zero,
Right half plane (RHP) zero,
V
V
V
V
IN
IN
IN
IN
COMP
COMP
COMP
COMP
R
R
R
R
C
C
C
C
C
C
C
C
C
C
C
C
Figure 6.
Figure 6.
Figure 6.
Figure 6.
R
R
R
R
R
R
R
R
O
O
O
O
POWER
POWER
POWER
POWER
STAGE
STAGE
STAGE
STAGE
-
-
-
-
Gm
Gm
Gm
Gm
+
+
+
+
O
O
O
O
VOLTAGE
VOLTAGE
VOLTAGE
VOLTAGE
REFERENCE
REFERENCE
REFERENCE
REFERENCE
is the equivalent output resistance of the error amplifier
is the equivalent output resistance of the error amplifier
is the equivalent output resistance of the error amplifier
is the equivalent output resistance of the error amplifier
Simplifi ed Equivalent Model of a Boost
Simplifi ed Equivalent Model of a Boost
Simplifi ed Equivalent Model of a Boost
Simplifi ed Equivalent Model of a Boost
Converter
Converter
Converter
Converter
The poles p1, p2 and the RHP zero z2 all increase phase
shift in the loop response. For stable operation, the overall loop gain should cross 0dB with -20dB/decade slope.
Due to the presence of the RHP zero, the 0dB crossover
frequency should not be more than
compensating zero z
provides phase boost beyond p2. In
1
general the converter is more stable with widely spaced
fi lter pole p2 and the RHP zero z2. The RHP zero moves to
low frequency when either the duty-cycle D or the output
current I
increases. It is benefi cial to use small inductors
OUT
and larger output capacitors especially when operating at
9
9
287
287
high
9
9
,1
,1
ratios.
−=−=ω
−=−=ω
5&
5&
287
287
and
&&
&&
()
()
'5
'5
−
−
=ω
=ω
/
/
C4
C4
C4
C4
R1
R1
R1
R1
R2
R2
R2
R2
,
.
I
I
I
I
OUT
OUT
OUT
OUT
V
V
V
V
ESR
ESR
ESR
ESR
R
R
R
R
C2
C2
C2
C2
]
]
. The internal
OUT
OUT
OUT
OUT
Feed-Forward Compensation
Figure 6 shows the equivalent circuit of a boost converter.
Important poles and zeros of the overall loop response
are:
Low frequency integrator pole,
−=ω
S
,
&5
&2
A feed-forward capacitor C
can be determined empirically by observing the induc-
of C
4
tor current and the output voltage during load transient.
Starting with a value between
adjusted until there is no excessive ringing or overshoot in
inductor current and output voltage during load transient.
Sizing the inductor such that its ripple current is about 0.5A
is needed for stability. The value
4
V µ
V µ
and
5
5
5
5
V µ
V µ
, C4 is
also improves phase margin and transient response.
2007 Semtech Corp. www.semtech.com
13
POWER MANAGEMENT
Applications Information (Cont.)
Figures 7(a)-7(c) show the effects of different values of
inductance and feed-forward capacitance on transient responses. In a battery-operated system if C4 is optimized for
the minimum VIN and the maximum load step, the converter
will be stable over the entire input voltage range.
V
OUT
0.5V/div
IL1
0.5A/div
SC4503
Board Layout Considerations
In a step-up switching regulator, the output fi lter capacitor,
the main power switch and the rectifying diode carry pulse
currents with high di/dt. For jitter-free operation, the size of
the loop formed by these components should be minimized.
Since the power switch is integrated inside the SC4503,
grounding the output fi lter capacitor next to the SC4503
ground pin minimizes size of the high di/dt current loop.
The input bypass capacitors should also be placed close to
the input pins. Shortening the trace at the SW node reduces
the parasitic trace inductance. This not only reduces EMI
but also decreases switching voltage spikes.
V
OUT
0.5V/div
0.5A/div
V
OUT
0.5V/div
40µs/div
(a) L1 = 5.6µH and C4 = 2.2pF
I
L1
40µs/div
(b) L1 = 5.6µH and C4 = 3.3pF
Figure 8 shows how various external components are
placed around the SC4503.
The large surrounding ground plane acts as a heat sink
for the device.
VINVOUT
L1D1
SW
JP
R3
R1 C4
GND
R2
C2
C1
U1
FB
C3
SHDN/SS
Figure 8. Suggested PCB Layout for the SC4503.
I
L1
0.5A/div
40µs/div
(c) L1 = 3.3µH and C4 = 2.7pF
Figure 7.
Different inductances and feed-forward capacitances affect the load transient responses of the
3.3V to 12V step-up converter in Figure 10(a).
is switched between 90mA and 280mA.
I
OUT
2007 Semtech Corp. www.semtech.com
14
SC4503
POWER MANAGEMENT
Typical Application Circuits
5V
C1
4.7µF
Figure 9.
Driving Two 6 White LED Strings from 5V. Zener diode D2 protects the converter
from over-voltage damage when both LED strings become open.
R3
54.9k
C3
56nF
5
IN SW
4
SHDN/SS
L1
10µH
SC4503
GND
2
1
FB
D1
ZHCS400
24V
3
+
_
C5
22nF
L1: Murata LQH32C
C1: Murata GRM219R60J475K
D2
MM5Z24VT1
C4
220pF
R4
301k
C2
0.22µF
R1
R2
63.4
63.4
2007 Semtech Corp. www.semtech.com
15
POWER MANAGEMENT
Typical Application Circuits
SC4503
FB
D1
10BQ015
3
C4
2.2pF
R1
866k
R2
100k
VOUT
12V
C2
4.7µF
VIN
3.3V
C1
2.2µF
R3
15k
4
C3
56nF
L1
2.7µH
5
IN SW
SC4503
SHDN/SS
GND
2
1
L1: Coiltronics LD1
C1: Murata GRM188R61A225K
C2: Murata GRM21BR61C475K
Figure 10(a). 3.3V to 12V Boost Converter with Soft-start
Efficiency vs Load Current
95
1.3MHz
90
85
80
75
70
Efficiency (%)
65
60
V
= 12V
55
50
0.001 0.010 0.100 1.000
Load Current (A)
Figure 10(b). Effi ciency vs Load Current
OUT
40µs/div
Upper Trace : Output Voltage, AC Coupled, 0.5V/div
Lower Trace : Input Inductor Current, 0.5A/div
Figure 10(c).
Load Transient Response of the Circuit
in Figure 10(a). I
is switched between
OUT
90mA and 280mA
2007 Semtech Corp. www.semtech.com
16
SC4503
POWER MANAGEMENT
Typical Application Circuits
1
FB
D1
10BQ015
3
C4
10pF
2.6 - 4.2V
1-CELL
LI-ION
OFF
< 0.4V
C1
4.7µF
ON
3.3V
R3
15k
C3
56nF
4
L1
1.5µH
5
IN SW
SC4503
SHDN/SS
GND
2
L1: TDK VLF4012AT
C1: Murata GRM188R60J475K
C2: Murata GRM21BR60J106K
Figure 11(a). Single Li-ion Cell to 5V Boost Converter
R1
187k
R2
60.4k
VOUT
5V
C2
10µF
Figure 11(b).
Efficiency vs Load Current
95
90
85
VIN = 4.2V
80
75
70
Efficiency (%)
65
60
55
50
0.001 0.010 0.100 1.000
Effi ciency of the Li-ion Cell to 5V
Boost Converter
VIN = 3.6V VIN = 2.6V
V
OUT
1.3MHz
Load Current (A)
= 5V
VIN= 2.6V
40µs/div
Upper Trace : Output Voltage, AC Coupled, 0.2V/div
Lower Trace : Inductor Current, 0.5A/div
Figure 11(c).
Load Transient Response. I
is switched
OUT
between 0.1A and 0.5A
2007 Semtech Corp. www.semtech.com
Upper Trace : Output Voltage, AC Coupled, 0.2V/div
Lower Trace : Inductor Current, 0.5A/div
Figure 11(d).
Load Transient Response. I
between 0.15A and 0.9A
17
40µs/div
VIN= 4.2V
is switched
OUT
POWER MANAGEMENT
Typical Application Circuits
SC4503
2.6 - 4.2V
1-CELL
LI-ION
C1
1µF
R3
8.06k
C3
0.22µF
5
IN SW
SC4503
4
SHDN/SS
L1
3.3µH
GND
C5
2.2µF
1
3
FB
2
D1
10BQ015
L2
3.3µH
L1 and L2: Coiltronics DRQ73-3R3
C1: Murata GRM188R61A105K
C2: Murata GRM21BR60J106K
C5: Murata GRM188R61A225K
Figure 12(a). Single Li-ion Cell to 3.3V SEPIC Converter.
C4
15pF
R1
412k
R2
249k
VOUT
3.3V, 0.45A
C2
10µF
Efficiency vs Load Current
85
V
= 3.3V
OUT
80
75
70
65
60
55
50
Efficiency (%)
45
40
35
30
0.001 0.010 0.100 1.000
Load Current (A)
VIN = 2.6V
VIN = 3.6V
VIN = 4.2V
Figure 12(b). Effi ciency vs Load Current
40µs/div
Upper Trace : Output Voltage, AC Coupled, 0.2V/div
Lower Trace : Input Inductor Current, 0.2A/div
Figure 12(c).
Load Transient Response of the Circuit
in Figure 12(a). I
VIN= 3.6V
is switched between
OUT
100mA and 500mA
2007 Semtech Corp. www.semtech.com
18
SC4503
POWER MANAGEMENT
Typical Application Circuits
3.3V
3.3V
ONOFF
< 0.4V
RUN
C1
4.7µF
R3
17.8k
5
IN SW
SC4503
4
SHDN/SS
C3
56nF
L1
4.7µH
GND
D2
1
FB
2
D3
C5
0.1µF
10BQ015
3
C9
0.1µF
D6
D1
D7
C6
0.1µF
D4
C4
12pF
OUT3
-8.5V (10mA)
C10
1µF
C7
0.1µF
R1
309k
R2
49.9k
D5
OUT2
26V (10mA)
C8
1µF
OUT1
9V (0.3A)
C2
4.7µF X 2
D2 - D7 : BAT54S
L1 : Sumida CDC5D23B-4R7M
C2: Murata GRM21BR61C475K
C1: Murata GRM188R61A105K
Figure 13(a). Triple-Output TFT Power Supply with Soft-Start
CH4
CH1
CH2
CH3
400µs/div
CH1 : OUT1 Voltage, 5V/div
CH2 : OUT2 Voltage, 20V/div
CH3 : OUT3 Voltage, 5V/div
CH4 : RUN Voltage, 5V/div
40µs/div
Upper Trace : Output Voltage, AC Coupled, 0.5V/div
Lower Trace : Inductor Current, 0.5A/div
Figure 13(b).
2007 Semtech Corp. www.semtech.com
TFT Power Supply Start-up Transient as
the RUN Voltage is Stepped from 0 to
3.3V
19
Figure 13(c).
Load Transient Response. I
OUT1
is switched between 50mA and
350mA
POWER MANAGEMENT
EVB Schematic
SC4503
12VOUT
R1
0R
C4
15pF
R2
432K
49.9K
C2
N.P.
R4
0R
C3
10uF
D1
SS13
8
N.C.
7
GND
6
GND
SC4503_MLPD
D1
SS13
L1
4.7uH
U1
1
SW
2
SW
3
VIN
SHDN/SSFB
45
C5
100nF
L1
4.7uH
1
SW
VIN
5
C1
10uF
R3
47K
5VIN
OFF/ON
JP1R5
5VIN12VOUT
R1
0R
C4
15pF
R2
432K
R5
49.9K
R4
0R
C2
N.P.
C3
10uF
2
GND
34
FB SHDN
U1
SC4503
R3
47K
C5
100n
C1
10uF
JP1
OFF/ON
2007 Semtech Corp. www.semtech.com
20
SC4503
POWER MANAGEMENT
Outline Drawing - TSOT-23
SEATING
PLANE
NOTES:
CONTROLLING DIMENSI ONS ARE IN MILLIMETERS (ANGLES IN DEGREES).
1.
DATUMS AND TO BE DETERMINED AT DATUM PLANE
2.
DIMENSIONS "E1" AND "D" DO NOT INCLUDE MOLD FLASH, PROTRUSIONS
3.
OR GATE BURRS.
REFERENCE JEDEC STD MO-193, VARIATION AB.
4.
A
e1
D
2X E/2
2X N/2 TIPS
N
E
E1
12
ccc
C
e
B
C
SIDE VIEW
-B-
D
A1
bxN
bbb C A-B D
A2
A
SEE DETAIL
aaa
C
-A- -H-
DIM
A
A1
A2
b
c
D
E1
E
e
e1
L
L1
N
01
aaa
bbb
ccc
GAGE
PLANE
0.25
A
MIN
.000
.028
.012
.003
.060
.012
0°
---
INCHES
.110 BSC
.037 BSC
H
DIMENSIONS
NOM
-
-
-
.114
.063
.018
(.024)
5
-
.004
.008
.010
MILLIMETERS
MAX MIN NOM MAX
.039
.004 0.00
.035
0.70
.020 0.30
.008
0.08
2.80.110
.118
.067 1.50
0.30
8° 0°
(L1)
DETAIL
A
-
2.80 BSC
0.95 BSC
1.90 BSC.075 BSC
L
2.90
1.60
0.45.024
(0.60)
5
0.10
0.20
0.25
-
1.00
-
0.10
-
0.90
-
0.50
-
0.20
3.00
1.70
0.60
8°
c
01
Land Pattern - TSOT-23
DIMENSIONS
X
DIM
DIM
C
C
G
21
G
P
P
X
X
Y
Y
Z
Z
(C)
G
Z
Y
P
NOTES:
1.
THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY.
CONSULT YOUR MANUFACTURING GROUP TO ENSURE YOUR
COMPANY'S MANUFACTURING GUIDELINES ARE MET.
2007 Semtech Corp. www.semtech.com
DIMENSIONS
INCHES
INCHES
(.087)
.031
.037
.024
.055
.141
MILLIMETERS
MILLIMETERS
(2.20)
0.80
0.95
0.60
1.40
3.60
POWER MANAGEMENT
Outline Drawing - 8 Lead 2X2mm MLPD-W
SC4503
A
PIN 1
INDICATOR
(LASER MARK)
aaa
C
A1
LxN
E/2
e
NOTES:
1.
CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES).
COPLANARITY APPLIES TO THE EXPOSED PAD AS WELL AS THE TERMINALS.
2.
Land Pattern - 8 Lead 2X2mm MLPD-W
D
D1
1
2
N
D/2
B
E
A
SEATING
PLANE
A2
C
E1
bxN
bbb C A B
e/2
DIM
A1
A2
D1
E1
aaa
bbb
INCHES
MIN
.028
A
.000
b
.007
D
.075
.059 .063 .067
E
.075
.031 .035 .039
e
.020 BSC
L
.008
N
DIMENSIONS
NOM
MAX
.031
.030
.002
.001
(.008)
.010
.083
.079
.079
.083
.012
.016
8
.003
.003
MILLIMETERS
MAX
NOM
MIN
0.70
0.00
0.18.012
1.90
1.50 1.60 1.70
1.90
0.80 0.90 1.00
0.75
0.02
(0.20)
0.25
2.00
2.00
0.50 BSC
0.30
8
0.08
0.08
0.80
0.05
0.30
2.10
2.10
0.400.20
Contact Information
Semtech Corporation
Power Management Products Division
200 Flynn Road, Camarillo, CA 93012
Phone: (805) 498-2111 Fax: (805) 498-3804
www.semtech.com
2007 Semtech Corp. www.semtech.com
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